A Phase Shifted Semi-Bridgeless Boost Power Factor Corrected Converter for Plug in Hybrid Electric

Vehicle Battery Chargers

Fariborz Musavi Department of Research, Engineering

Delta-q Technologies Corp. Burnaby, BC, Canada fmusavi@delta-q.com

1 Wilson Eberle and 2 William G. Dunford Dept. of Electrical and Computer Engineering

University of British Columbia | 1 Okanagan | 2 Vancouver 1 Kelowna, BC, Canada | 2 Vancouver, BC, Canada

1 wilson.eberle@ubc.ca | 2 wgd@ece.ubc.ca

Abstract—In this paper, a phase shifted semi-bridgeless boost power factor corrected converter is proposed for plug in hybrid electric vehicle battery chargers. The converter features high efficiency at light loads and low lines, which is critical to minimize the charger size, charging time and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level I residential charging applications.

A detailed converter description and steady state operation analysis of this converter is presented. Experimental results of a prototype boost converter, converting universal AC input voltage to 400 V DC at 3.4 kW are given and the results are compared to an interleaved boost converter to verify the proof of concept, and analytical work reported. The results show a power factor greater than 0.99 from 750 W to 3.4 kW, THD less than 5% from half load to full load and a peak efficiency of 98.6 % at 240 V input and 1000 W load.

I. INTRODUCTION A plug in hybrid electric vehicle (PHEV) is a hybrid

vehicle with a storage system that can be recharged by connecting a plug to an external electric power source. The charging AC outlet inevitably needs an on-board AC-DC charger with power factor correction [1]. An on-board 3.4 kW charger could charge a depleted battery pack in PHEVs to 95% charge in about four hours from a 240 V supply [2].

A variety of circuit topologies and control methods have been developed for PHEV battery chargers. The two-stage approach with cascaded PFC AC-DC and DC-DC converters is the common architecture of choice for PHEV battery chargers, where the power rating is relatively high, and lithium-ion batteries are used as the main energy storage system [3]. The single-stage approach is generally only suitable for lead acid batteries due to large low frequency ripple in the output current.

In the two-stage architecture, the PFC stage rectifies the input AC voltage and transfers it into a regulated intermediate DC link bus. At the same time, power factor correction is achieved [4]. The boost circuit-based PFC topology operated in CCM is employed in this study as the main candidate for front end single-phase solutions for AC-DC power factor corrected converters used in PHEV battery chargers.

A. Conventional Boost Converter The conventional boost topology shown in Fig.1 uses a

dedicated diode bridge to rectify the AC input voltage to DC, which is then followed by the boost section. In this topology, the output capacitor ripple current is very high [5] and is the difference between diode current and the dc output current. Furthermore, as the power level increases, the diode bridge losses significantly degrade the efficiency, so dealing with the heat dissipation in a limited area becomes problematic. Due to these constraints, this topology is good for a low to medium power range up to approximately 1kW. For power levels >1kW, typically, designers parallel semiconductors in order to deliver greater output power. The inductor volume also becomes a problematic design issue at high power because of permeability drops at higher load and heat associated with core and copper losses.

Figure 1. Conventional PFC boost topology

This work has been sponsored and supported by Delta-q Technologies Corporation.

B. Bridgeless Boost Converter The bridgeless configuration topology shown in Fig.2

avoids the need for the rectifier input bridge yet maintains the classic boost topology [6-13]. It is an attractive solution for applications >1kW, where power density and efficiency are important. The bridgeless boost converter solves the problem of heat management in the input rectifier diode bridge, but it introduces increased EMI [14, 15]. Another disadvantage of this topology is the floating input line with respect to the PFC stage ground, which makes it impossible to sense the input voltage without a low frequency transformer or an optical coupler. Also in order to sense the input current, complex circuitry is needed to sense the current in the MOSFET and diode paths separately, since the current path does not share the same ground during each half-line cycle [8, 16].

Figure 2. Bridgeless PFC boost topology

C. Interleaved Boost Cconverter The interleaved boost converter shown in Fig.3 is simply

two boost converters in parallel operating 180° out of phase [20-22]. The input current is the sum of the two inductor currents. Because the inductors ripple currents are out of phase, they tend to cancel each other and reduce the input ripple current caused by the boost switching action. The interleaved boost converter has the advantage of paralleled semiconductors. Furthermore, by switching 180° out of phase, it doubles the effective switching frequency and introduces smaller input current ripple, so the input EMI filter can be smaller [23-25]. This converter also has reduced output capacitor high frequency ripple, but it still has the problem of heat management for the input diode bridge rectifiers.

Figure 3. Interleaved PFC boost topology

In the following section, a new phase shifted semi-bridgeless boost PFC converter is proposed in order to improve overall efficiency of the AC-DC PFC converter,

while maintaining all the advantages of the existing solutions.

II. PHASE SHIFTED SEMI-BRIDGELESS BOOST TOPOLOGY The phase shifted semi-bridgeless topology shown in

Fig.4 is proposed as a solution to address the problems outlined in section I for the conventional boost, bridgeless boost and interleaved boost topologies. This topology features high efficiency at light loads and low lines, which is critical to minimize the charger size, charging time and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level I residential charging applications in North America where the typical supply is limited to 120V and 1.44kVA.

The proposed topology introduces two more slow diodes (Da and Db) to the bridgeless configuration to link the ground of the PFC to the input line. However, the current does not always return through these diodes, so their associated conduction losses are low. This occurs since the inductors exhibit low impedance at the line frequency, a large portion of the current flows through the FET intrinsic body diodes. Also the gating signals for FETs are 180° out of phase.

A detailed converter description and steady-state operation analysis is given in the following section.

Figure 4. Phase shifted semi-bridgeless PFC boost topology

III. OPERATING PRINCIPLE AND STEADY-STATE ANALYSIS

To analyze the circuit operation, the input line cycle has been separated into the positive and negative half-cycles as explained in sub-sections A and B that follow. In addition, the detailed circuit operation depends on the duty cycle. Positive half-cycle operation analysis is provided for D > 0.5 in sub-section C and D < 0.5 in sub-section D.

A. Positive Half-Cycle Operation Referring to Fig. 4, during the positive half-cycle, when

the AC input voltage is positive, Q1 turns on and current flows through L1 and Q1 and continues through Q2 and then L2, returning to the line while storing energy in L1 and L2. When Q1 turns off, energy stored in L1 and L2 is released as current flows through D1, through the load and returns through the body diode of Q2/partially through Db back to the input.

B. Negative Half-Cycle Operation Referring to Fig. 4, during the negative half-cycle, when

the AC input voltage is negative, Q2 turns on and current flows through L2 and Q2 and continues through Q1 and then L1, returning to the line while storing energy in L2 and L1. When Q2 turns off, energy stored in L2 and L1 is released as current flows through D2, through the load and returns split between the body diode of Q1 and Da back to the input.

C. Detailed Positive Half-Cycle Operation and Analysis for D > 0.5 The detailed operation of the proposed converter depends

on the duty cycle. During any half-cycle, the converter duty cycle is either greater than 0.5 (when the input voltage is smaller than half of output voltage) or smaller than 0.5 (when the input voltage is greater than half of output voltage). The three unique operating interval circuits of the proposed converter are provided in Fig. 5 to Fig. 7 for duty cycles larger than 0.5 during the positive half-cycle.

Figure 5. Interval 1and 3: Q1 and Q2 are ON

Figure 6. Interval 2: Q1 ON, body diode of Q2 conducting

Figure 7. Interval 4: Q1 OFF and Q2 ON

Waveforms of the proposed converter during positive half-cycle operation with D>0.5 are shown in Fig. 8. The intervals of operation are explained as follows. In addition,

the ripple current components are derived, enabling calculation of the input ripple current, which provides design guidance to meet the required input current ripple standard.

Figure 8. Phase shifted semi-bridgeless boost converter steady-state Waveforms at D > 0.5

Interval 1 [t0-t1]: At t0, Q1/ Q2 are on, as shown in Fig.5. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The energy stored in Co provides energy to the load. The ripple currents in Q1 and Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by:

∆I L L v D T (1)

Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown in Fig.6. During this interval, the current in series inductances L1 and L2 continues to increase linearly and store the energy in these inductors. The energy stored in Co provides the load energy. The ripple currents in Q1 and body diode of Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by:

∆I L L v 1 D T (2)

Interval 3 [t2-t3]: At t2, Q1/Q2 are on again, and interval 1 is repeated, as shown in Fig. 5. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The ripple currents in Q1 and Q2 are the same as the ripple current in series inductances L1 and L2, as shown in equation (1).

Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown I Fig. 7. During this interval, the energy stored in L1 and L2 is released to the output through L1, D1, Q2 and L2. The ripple currents in D1 and Q2 are the same as the ripple currents in L1 and L2:

∆ 1 (3)

Figure 9. Interval 1and 3: Q1 and Q2 are OFF, body diode of Q2 conducting

Figure 10. Interval 2: Q1 ON, body diode of Q2 conducting

Figure 11. Interval 4: Q1 OFF and Q2 ON

D. Detailed Positive Half-Cycle Operation and Analysis for D < 0.5

The three unique operating interval circuits of the proposed converter are given in Fig. 9 to Fig. 11 for duty cycles smaller than 0.5 during the positive half-cycle. The waveforms of the proposed converter during these conditions

are shown in Fig. 12. The intervals of operation are explained as follows.

Figure 12. Phase shifted semi-bridgeless boost converter steady-state waveforms at D < 0.5

Interval 1 [t0-t1]: At t0, Q1/ Q2 are off, as shown in Fig.9. During this interval, the energy stored in L1 and L2 are released to the output through L1, D1, body diode of Q2 and L2. The ripple currents in D1 and body diode of Q2 are the same as the ripple currents in L1 and L2:

∆I L L D T (4)

Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown in Fig.10. During this interval, the current in series inductances L1 and L2 continues to increase linearly and store the energy in these inductors. The energy stored in Co provides energy to the load. The ripple currents in Q1 and the body diode of Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by:

∆I L L v DT (5)

Interval 3 [t2-t3]: At t2, Q1/Q2 are off again, and interval 1 is repeated, as shown in Fig. 9. During this interval, the current in series inductances L1 and L2 increases linearly

and stores the energy in these inductors. The ripple currents in D1 and body diode of Q2 are the same as the ripple current in series inductances L1 and L2, as shown in equation (1).

Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown I Fig. 11. During this interval, the energy stored in L1 and L2 is released to the output through L1, D1, Q2 and L2. The ripple currents in D1 and Q2 are the same as the ripple currents in L1 and L2:

∆ (6)

The operation of converter during the negative input voltage half-cycle is similar to the operation of converter during the positive input voltage half-cycle.

IV. LOSS EVALUATION The estimated loss distribution of the semiconductors is

provided in Fig. 13 at 70 kHz switching frequency, 240V input and 3300W load for benchmark conventional boost and interleaved boost converters and the proposed phase shifted semi-bridgeless boost converter. The currents in regular diodes Da and Db were assumed to be split with the current going through intrinsic body diodes for phase shifted semi-bridgeless topology. The regular diodes in input bridge rectifiers have the largest share of losses among the topologies with the input bridge rectifier. The phase shifted semi-bridgeless topology nearly eliminates this large loss component (~30W). However, the tradeoff is that the FET losses are higher and the intrinsic body diodes of FETs conduct, producing new losses (~8W). The fast diodes in the conventional and interleaved PFC have slightly lower power losses, since the boost RMS current is higher in these topologies.

Figure 13. Comparison of the estimated loss distribution in the semiconductors at 70kHz switching frequency, 240V input, 3300W load at 400V

Overall the FETs are under slightly more stress in phase shifted semi-bridgeless topology, but the total loss for the

proposed phase shifted semi-bridgeless boost are 17% lower than the benchmark conventional boost and 7% lower than the interleaved boost . Since the benchmark converter bridge rectifier losses are large, it is expected that phase shifted semi-bridgeless boost converter should have the lowest losses among the topologies investigated. Additionally, it is noted that the losses in the input bridge rectifiers are 63% of total losses in the conventional PFC converter and 71% of total losses in the interleaved PFC converter. Therefore, eliminating the input bridge in PFC converters is justified despite that the introduction of new losses.

V. EXPERIMENTAL RESULTS

Prototypes of a phase shifted bridgeless boost converter and an interleaved boost converter were built to verify the proof-of-concept and analytical work presented in this paper and to benchmark the proposed converter. The devices used in experimental prototypes are provided in Table 1.

TABLE I. DEVICES/COMPONENTS USED IN EXPERIMENTAL PROTOTYPES

Topology Components Used in Prototype Unit Head

Device Part # / Value # of Devices

Phas

e Sh

ifted

Se

mi-b

ridge

less

PF

C c

onve

rter Regular Diode 25ETS08S 2

Fast Diode IDB06S60C 2

MOSFET IPB60R099CP 2

Inductors 400 μH 2

Inte

rleav

ed P

FC

conv

erte

r

Regular Diode 25ETS08S 4

Fast Diode IDB06S60C 2

MOSFET IPB60R099CP 2

Inductors 400 μH 2

Pictures of the proposed phase shifted bridgeless boost prototype are provided in Fig. 14. It consists of a control board, a capacitor bank of 820 μF and an IMS power board attached to a heatsink with the PFC inductors.

Figure 14. Top: control board, Bottom: power board

30.0

W

6.9W 10

.8W

0.0W

47.7

W

30.0

W

6.9W

5.4W

0.0W

42.3

W

4.1

9.8

21.6

4.1

39.4

0

10

20

30

40

50

60

Reg

ular

D

iode

s

Fast

D

iode

s

FETs

Intri

nsic

B

ody

Dio

des

Tota

l Lo

sses

Pow

er L

osse

s (W

)

Semiconductor Losses

Conventional Boost

Interleaved Boost

Phase Shifted Semi-Bridgeless Boost

The experimental efficiency of the phase shifted bridgeless boost converter and benchmark interleaved boost converter is provided in Fig. 15 for 240V input and Fig. 17 for 120V input at 70 kHz switching frequency and 400 V output. Loss reduction curves as a function of output power are provided in Fig. 16 and Fig. 18 for 240V and 120V input, respectively.

Figure 15. Efficiency as a function of output power at Vin = 240V, Vo=400V and 70kHz switching frequency

Figure 16. Loss reduction as a function of output power at Vin = 240V, Vo=400V and 70kHz switching frequency

Figure 17. Efficiency as a function of output power at Vin = 120V, Vo=400V and 70kHz switching frequency

Figure 18. Loss reduction as a function of output power at Vin = 120V, Vo=400V and 70kHz switching frequency

From the results, it is noted that proposed semi-bridgeless PFC converter achieves a peak efficiency of 98.6% at 1 kW output power. Additionally, the light load efficiency of the proposed converter is significantly better than that of the interleaved PFC due to the absence of input bridge rectifier. However, as the load increases, the efficiency drops due to additional heat dissipation in the intrinsic body diodes of the FETs.

Figure 19. THD as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz switching frequency

Figure 20. Power Factor as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz switching frequency

94

95

96

97

98

99

100

0

500

1000

1500

2000

2500

3000

3500

Effic

ienc

y (%

)

Output Power (W)

Interleaved PFC Converter

Phase Shifted Semi-Bridgeless PFC Converter

0

10

20

30

40

50

60

70

0

500

1000

1500

2000

2500

3000

3500

Loss

Red

ucti

on (%

)

Output Power (W)

Loss Reduction for PFC Converters at Vin = 240 V

91

92

93

94

95

96

97

98

0

200

400

600

800

1000

1200

1400

1600

1800

Effic

ienc

y (%

)

Output Power (W)

Interleaved PFC Converter

Phase Shifted Semi-Bridgeless PFC Converter

0

10

20

30

40

50

60

70

0

200

400

600

800

1000

1200

1400

1600

1800

Loss

Red

ucti

on (%

)

Output Power (W)

Loss Reduction for PFC Converters at Vin = 120 V

051015202530354045

0

500

1000

1500

2000

2500

3000

3500

THD

(%)

Output Power (W)

Vin=240

Vin=120

0.840.860.880.9

0.920.940.960.98

11.02

0

500

1000

1500

2000

2500

3000

3500

Pow

er fa

ctor

Output Power (W)

Vin=240

Vin=120

Figure 21. Harmonics orders at Vin = 120 V and 240V, compared against EN61000-3-2 standard.

In order to verify the quality of the input current, the input current THD is shown in Fig.19. The power factor and harmonic orders are given and compared with EN 61000-3-2 standard in Fig.20 and 21. It is noted that mains current THD is less than 5% from 50% load to full load and it is compliant to IEC 6100-3-2 (Fig. 19 and Fig. 21). The converter power factor is shown over entire load range for 120 and 240V input in Fig. 20. The power factor is greater than 0.99 from 50% load to full load.

Experimental waveforms from the proposed converter prototype are provided in Fig. 22 through Fig. 26. The input current, input voltage and output voltage are given in Fig. 22. As it can be seen, the input current is in phase with the input voltage and has a sinusoidal shape. Additionally, there is a low frequency ripple on output voltage, which is inversely proportional to the value of PFC bus output capacitors.

In Fig. 23, the inductor current is provided in addition to the above mentioned waveforms from Fig. 22. It is noted that during the positive half-cycle, the inductor current is the same as input current. However, during the negative half-cycle, the input current is partially flowing through slow diodes, Da and Db.

Figure 22. Inut current, input voltage and output voltage. Ch1= Vo 100V/div. Ch2= Vin 100V/div. Ch4= Iin 10A/div.

Figure 23. Inut current, inducotr current, input voltage and output voltage. Ch1= Vo 100V/div. Ch2= Vin 100V/div. Ch3= IL1 10A/div. Ch4= Iin

10A/div.

Figure 24. Inductor current, input current and sensed FET current. Ch1= Sensed IQ1 2V/div. Ch3= IL1 / IDb 10A/div. Ch4= Iin 10A/div

Figure 25. Gating signal, Inductor and sensed FET current for D < 0.5 Ch1= Vg 10V/div. Ch2= IQ1 2V/div. Ch3= IL1 10A/div

In Fig.24 the inductor current, input current and current sensed in the FET through a current transformer are given.

0

0.5

1

1.5

2

2.53 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39

Am

plit

ude

(A)

Harmonics Order

EN 61000-3-2 Class D Limits (A)

Amplitude (A) Vin = 120 V

Amplitude (A) Vin = 240 V

Input Voltage

Output Voltage

Input Current

Input Voltage

Output Voltage

Input Current

Inductor Current

Input Current Inductor

Current

Sensed FET Current

Inductor Current

Gating Signal

Sensed FET Current

The gating signals, sensed FET current and the inductor current are provided for duty cycles less than 0.5, Fig 25, and greater than 0.5, Fig. 26. These waveforms match the theoretical models.

Figure 26. Gating signal, Inductor and sensed FET current for D > 0.5 Ch1= Vg 10V/div. Ch2= IQ1 2V/div. Ch3= IL1 10A/div

VI. CONCLUSIONS

A new high performance phase shifted semi-bridgeless AC-DC Boost converter topology has been presented in this paper for the front-end AC-DC converter in PHEV battery chargers. The proposed converter features high efficiency at light loads and low lines, which is critical to minimize the charger size, charging time and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level I residential charging applications in North America where the typical supply is limited to 120V and 1.44kVA.

An analysis and performance characteristics are presented. A breadboard converter circuit has been built to verify the proof-of-concept. The theoretical waveforms were compared with the results taken from prototype unit. Additionally, key experimental waveforms were provided and input current harmonics at each harmonic order were compared more explicitly with the IEC 6100-3-2 standard limits.

Experimental results demonstrate that the mains current THD is smaller than 5% from 50% load to full load and the converter is compliant with the IEC 6100-3-2 standard. The converter power factor was also provided for full power range at 120 and 240V input. The power factor is greater than 0.99 from 50% load to full load. The proposed converter achieves a peak efficiency of 98.6 % at 240 V input and 1 kW output power.

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[15] Baur, T.; Reddig, M.; Schlenk, M.; "Line-conducted EMI-behaviour of a High Efficient PFC-stage without input rectification," Infineon Technology – Application Note, 2006.

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Inductor Current

Gating Signal

Sensed FET Current