594 IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND …wangcheng.mit.edu/sites/default/files/documents/TST2013.pdf · 2018. 6. 5. · Real time imaging capability is en-abled by using

594 IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY, VOL. 3, NO. 5, SEPTEMBER 2013

Real-Time ImagingWith a 140 GHz Inverse SyntheticAperture Radar

Binbin Cheng, Ge Jiang, Cheng Wang, Chen Yang, Yingwu Cai, Qi Chen, Xiang Huang, Genghua Zeng,Jun Jiang, Xianjin Deng, and Jian Zhang

Abstract—A summary of a 140 GHz inverse synthetic apertureradar (ISAR) is presented. High resolution and real time imagingare two key points in the design. Using the frequency-modulatedcontinuous-wave (FMCW) radar technique with a 5 GHz band-width, 3 cm range resolution is achieved with assistance of softwarecalibration compensating for nonlinearities of the LFMwaveform.In cross-range region, synthetic aperture technique is adopted toimprove imaging resolution. Real time imaging capability is en-abled by using a hardware structure constructed with

in signal processing unit. The final system was usedto get ISAR image of a rotating target at 5 m. The 2-D (range andcross-range) imaging resolution is 3 cm 3 cm with a real timeimage frame rate of 5 Hz.

Index Terms—Real time imaging, terahertz (THz) radar, ISAR.

I. INTRODUCTION

I N ORDER TO make efficient classification and identifica-tion for radar targets, high detecting resolution is needed in

the detecting system. One method is to increase the carrier fre-quency to get larger sufficient bandwidth [1]. In sub-centimeterwaves, Ka-band systems were preferred because the technologyis available and the atmospheric propagation loss is lower espe-cially in adverse weather conditions [2]. But to get centimeter-scale or even higher resolution, the extension from Ka- to W orterahertz (THz) band is needed. With the extension of existingtechnologies and the development of semiconductor techniquesat higher millimeter wave bands, the 140 GHz transmissionwindow region may gain more importance in the future becauseof the considerable progress in THz technology, especially forshort and medium range detections [3]. As a fully harmless elec-tromagnetic wave, THz is also of high interest for security ap-plications because it can penetrate dielectric or non-conductingmaterials like plastics, ceramics, paper, cardboard, wood, clothand so on, but reflected by conducting surfaces like metallic

Manuscript received April 09, 2013; revised May 10, 2013; accepted May30, 2013. Date of publication July 09, 2013; date of current version September18, 2013. This work was supported by Terahertz Research Program, TerahertzResearch Center, China Academy of Engineering Physics (IEE-CAEP).B. Cheng, C. Wang, Y. Cai, G. Zeng, J. Jiang, X. Deng, and J. Zhang are

with the Institute of Electronic Engineering, China Academy of EngineeringPhysics, Mianyang 621900, China, and also with Terahertz Research Center,China Academy of Engineering Physics, Mianyang 621900, China (e-mail:[email protected]).G. Jiang, C. Yang, Q. Chen, and X. Huang are with Institute of Electronic En-

gineering, China Academy of Engineering Physics, Mianyang 621900, China.Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TTHZ.2013.2268317

materials [4]. These characteristics make it possible to detectperson-borne concealed threats even in a standoff geometry [5].As a candidate technology for high resolution imaging, THz

radar has drawn more and more attentions in recent years. Thereare two main methods to get an image of the targets for THzradar. The first method is beam scanning [6]–[8], which is touse an aperture focusing the beam to a small spot size to get asingle-pixel and rotate the reflector in the horizontal and verticaldirections to scan over the target region. According to the 1-Drange profile of every spot and the position of this point, the3-D image of the target can be reconstructed. The representa-tive system of this type is an ultra wide band 675 GHz radardeveloped by NASA Jet Propulsion Laboratory (JPL) whichwas designed for standoff personnel screening [9]. The sourceof this system was designed and built using multiplier chainand get a 25 m operation range. The imaging resolution is 0.7cm (range) 1 cm (cross-range) with a 1 Hz frame rate. Thesecondmethod is inverse synthetic aperture radar (ISAR) whoseimaging depends on the rotation of the targets [10], [11]. Com-pared with the first one, the image resolution of ISAR is in-dependent with range and the optical path of which is sim-pler. Fraunhofer IAF has developed a 220 GHz ISAR experi-mental radar COBRA [12] using all solid-state semiconductortechnology. Data were sampled over a full turn of 360 andthe resolution is about 2 cm. Massachusetts Lowell Universitybuilt a THz inverse synthetic aperture radar (ISAR) with 2.4THz quantum cascade laser transmitter [13]. By controlling thetarget rotating around a vertical axis and a horizontal axis, ISARgot azimuth and elevation image with sub-millimeter resolution.But this system needs an optically pumped molecular laser as areference and local oscillator.Recently, Terahertz Research Center in China Academy of

Engineering Physics (CAEP) has developed a 140 GHz ISARthat is capable of real time imaging [14]. In this paper, wepresent an overview of CAEP’s THz imaging radar technology.The radar is currently a portable laboratory prototype systemoperating in a frequency-modulated continuous-wave (FMCW)mode with a bandwidth of 5 GHz and the center frequency isabout 139.9 GHz. With a peak output power below 0.5 mW, theradar is far below the relevant accepted RF or laser safety healthhazard thresholds. The radar can generate five ISAR imagesper second, in real time, of target at a standoff range of 5 m.The 3 dB beam width of the transmitting antenna in this THzISAR system is about 7 , which causes a the correspondingbeam excursion at 5 m standoff to about 60 cm in the horizontaland vertical directions. The scattering points on the rotatingtargets covered by this beam can be identified as long as the

2156-342X © 2013 IEEE

CHENG et al.: REAL-TIME IMAGING WITH A 140 GHz INVERSE SAR 595

Fig. 1. Basic principle for ISAR.

distance between them exceeds the radar’s 3 cm scale rangeand cross-range resolution.The THz radar has been built largely from self-research com-

ponents, and integrating them into a functioning system posesa substantial challenge. Here we will describe some key de-sign approaches taken for the radar’s detection methodology,key components design, electronics architecture, and real timesignal processing unit, which have enabled 5 frames/s imagingspeed. And at last the images of several targets such as he-licopter model and corner reflectors made by the THz ISARsystem are demonstrated. The experiment results prove the 3cm scale resolution of this THz radar.

II. ISAR DETECTION

A. Basic Concepts of ISAR

Typical ISAR emits wide band waveform to get high resolu-tion in range profile, while the cross-range, or so called Dopplerrange, attained by target’s rotational motion. Fig. 1 shows theprojection of a simple situation to - plane that coherent radarilluminates a target rotating around -axis with an angular ve-locity .As in Fig. 1, assume that a scattering point located initially

at has a polar coordinate , the relation-ship between distance from to radar and time is

(1)

where is the distance from radar to the rotation center A and. The approximation can be made for (1) to get the

following:

(2)

From (2), we can get the Doppler frequency of the reflectedwaves at

(3)

Normally, the data for ISAR processing will be acquired ina short term near . Equations (2) and (3) can be approxi-mated as (4) and (5), respectively

(4)

(5)

Functions (4) and (5) indicate that the equidistance planesand equi-Doppler planes are perpendicular and parallel to thedirection of propagation from radar toward the target, whichmake the basic concepts of Range-Doppler (R-D) imaging forISAR.

B. FMCW Radar and the Range Resolution

FMCW radar is a useful method to get high signal-to-noiseratio (SNR) for low maximum power radiation source inshort pulse duration. This is just the condition that THz radarconfronted with for standoff detection. Though large progresshas been made recently in solid state THz source, the peakoutput power of continuous radiation in most THz region is nomore than several milliwatts. In the 140 GHz ISAR, a solidstate Schottky-diode-based Subharmonic Mixer (SHM) wasemployed for modulation, and then power amplified to produce0.4 mW peak output power.The basic principles of FMCW radar can be seen in [20].

Here we just discuss the range resolution of the 140 GHz ISARaccording to mathematical description of FMCW detection.The complex exponential format of transmitted wave ampli-

tude versus time of FMCW signal can be represented as (6)

(6)

where is the chirp’s starting angular frequency and is thechip rate. The chirp signal sweep linearly over a bandwidth intime with a rate of . After reflection from a targetat range , the received signal will be a time delayof with amplitude distortion.

(7)

In the receiver mixer, this signal is multiplied with the LOsignal similar to the transmitted signal but with an offset startingangular frequency, . The output is filtered to get narrowband IF signal with description of

(8)

Thus, for point-like target, the dechirped IF signal is singletone with a frequency of , which meansthat apart from an additive constant , the detected inter-mediate frequency is proportional to the range to target with aproportionality factor of . The finite LO offset frequency

is for convenience of simpler IF impedance matchingin the receiver mixer, lower noise contributions, and down-stream quadrate I/Q down conversion prior to digitization [9].After digitization, the IF signals is analyzed by Fourier trans-

form, whose spectral resolution is the inverse of the total mea-


Fig. 2. Signal dechirping processing. (a) Time–frequency representation of theemit signal. (b) Time–frequency representation of the echo of target with threescattering centers at different range (dotted lines) compared with the emission(solid line). (c) Time–frequency representation of IF signal after dechirpingmixer, three simple-frequency signals (dotted lines) indicating three scatteringcenters.

surement time, in this design, with formula .So we get

(9)

Equation (9) means that the range resolution of an idealFMCW radar depends only on the radar’s bandwidth and isindependent of SNR and .For the 140 GHz ISAR imaging radar, 5 GHz bandwidth ands yield an IF shift of about 2 MHz at 6 m distance.

The theoretical range resolution, which can be calculated by (9),is 3 cm.

C. Range-Doppler Algorithm and Cross-Range Resolution

In the receiver, stretch processing is employed using sameSubharmonic Mixer as in the transmitter following a low noiseamplifier. The wide band FMCW echo is dechirped to get anarrow band intermediate frequency (IF) signal. The dechirptechnique simplifies the analog-to-digital conversion and signalprocessing unit. Fig. 2 shows the dechirp processing for a targetwith three scattering points at different range. Targets at dif-ferent ranges will act as single tones with corresponding fre-quencies after dechirping.The digitized IF signal is processed using the R-D algorithm

in the digital signal processor. A simplified R-D processing isshown in Fig. 3. Suppose that samples were acquired in asingle echo pulse, and for one frame, pulses were required.These samples make an matrix in total. In range profile,-point inverse discrete Fourier transform (IDFT) was made

for all M rows to get One-dimensional range profile matrix.After motion compensation, M-point discrete Fourier transform(DFT) was made in cross-range profile and the 2-D image forthe target is obtained.The cross-range resolution can be calculated by (10) [20]

(10)

where corresponds to the wavelength for the center fre-quency, . is the total viewing angle while the target rotating.In the 140 GHz imaging radar, the cross-range resolution is

selected to be equal to range resolution at 3 cm for convenience.The center frequency is 139.9 GHz. So according to (10), thetotal viewing angle should be about 2.1 .

Fig. 3. Block diagrams of R-D image processing for ISAR.

Fig. 4. Block diagram of 140 GHz ISAR.

III. 140 GHZ RADAR ELECTRONICS

A. RF Architecture Overview

The basic architecture of the 140 GHz ISAR is shown inFig. 4 for simplified block diagram, with the hardware dividedinto seven modules labeled by the dashed boxes from (A)–(F).The LFM base band signal with a span from 3 GHz to 8 GHzgenerated by AWG 7122B was divided into two paths (A). Oneis up-converted and power amplified through the Tx chain (C).And the other one is used for stretch processing in IFmodule (F).Module (B) acts as the local oscillator for subharmonic mixers(SHMs) in both Tx and Rx chains (D). The main component of(B) is two V-band multipliers driving by Ka-band sourceunit operating at 33.6 GHz and 33.593 GHz, respectively. The137.4–142.4 GHz transmit beam signal is propagated by thetransmitting antenna in (E), which is a 24 mm 18 mm hornwith 26 dB gain and 7 beam width (3 dB). The reflected signalis collected by a receiving antenna that has same characteristicswith the transmitting one. Through a WR6 standard rectangularwaveguide interface, the signal is low noise amplified and downconverted by (D) with a same SHM. The IF signals is stretchedwith a mixer in (F) to get a narrow base band. A 200 MSPSanalog-to-digital converter (ADC) is used for sampling to con-venient subsequent signal processing.


Fig. 5. Structure of 140 GHz sub-harmonic mixer.

B. 140 GHz Critical RF Components

In the 140 GHz ISAR imaging system based on super-het-erodyne structure, mixers, filters, and LOs are the critical com-ponents in determining the performance of THz transmitter andreceiver. In this part, the design methods and testing results ofthe three critical components are described.140 GHz Subharmonic Mixer: The 140 GHz subharmonic

mixer, used for frequency up-converting in Tx chain anddown-converting in Rx chain, is designed based on low para-sitic component Schottky diode UMS DBES105a. Suspendedmicrostrip structure is employed for the SHM designed withhigh frequency structure simulator (HFSS) advanced designsystem (ADS) co-simulation method.The architecture of 140 GHz sub-harmonic mixer is pre-

sented in Fig. 5. Classical standard rectangular waveguide Eplane split-block and hybrid integrated architecture are ap-plied. The RF and LO input waveguide are WR6 and WR15.Mixer passive circuit is based on shielded microstrip, which isfabricated on 127 m thickness quartz substrate. The discreteanti-parallel Schottky barrier diodes are flipped-chip mountedon to circuit with silver-epoxy. The RF and LO single arecoupled to the diodes by wideband waveguide E plane probe.The RF rejection low pass filter is designed to rejection RFsingle’s leakage to LO and IF port. The DC grounding of diodesis also achieved by long gold wire connection. A wideband LOIF diplexer which consists of LO probe, LO rejection low passfilter and long gold wire connection is designed to introduce IFsignal. IF port is connected with 2.92 mm K type connector.Fused silica chip based suspended microstrip and the Schottkydiode are boded to the oxygen-free copper cavity by conductivesilver paste using micropackage technology.The prototype of this mixer architecture comes from a

220 GHz tripler design [15]. The main advantage is that itcould provide the wider mixer frequency response comparedwith mixer presented in [16], which is restricted by probe re-sponse. The mixer covers nearly the entire waveguide operationband.Schottky diode modeling and impedance matching are two

key points in the SHM design. The 3-D EM diode model isestablished under the research on physical structure of diodesemiconductor.

Fig. 6. Simulation/measurement results of SSB conversion loss for the 140GHz sub-harmonic mixer.

Back-to-back method [17] was adopted in the 140 GHzsubharmonic mixer’s conversion loss simulation and mea-surements. The results are shown in Fig. 6. Simulation resultsindicate that the SSB conversion loss of the mixer is 6.3 dB in140 GHz with 7 dBm, 65 GHz LO pumping; 1 dB conversionloss bandwidth is 14 GHz and DSB noise temperature withinbandwidth is below 400 K. This SHM is also used in 140 GHzcommunication system [18].140 GHz BPF: The structure of 140 GHz waveguide H

ladder filter based on WR6 standard rectangular waveguide( mm, mm) consists of 7 half wavelengthresonators and 8 coupling structures. The Chebyshev filtermatrix synthesize method used in current design comes from[19].The 140 GHz band pass filter (BPF) was fabricated as split

block by micrometer precision high speed CNC milling ma-chines with tolerance 2.5 m based on micro- electromechan-ical systems (MEMS) technology. Blockmaterial is copper withsurface plated with nickel–gold to avoid rusting. Fig. 7(a) showsthe picture of the final filter block.The filter is first simulated by mode matching method with

mode number set to 30. Generally, mode matching method ismore precision than finite element simulation tools like HFSS(high frequency structure simulator), which is restricted by thegirding density. And then the filter S -parameter has been mea-sured from 110 GHz to 170 GHz by Farran FEV-06-TR-0003D band vector network analyzer frequency extender.The difference between measured and simulated data is

showed in Fig. 7(b). Seven return loss resonating points couldbe observed clearly. Measured data indicates the filter centerfrequency is 139.2 GHz. The 3 dB bandwidth is 11.6 GHz,relatively 8.3%. The minimum insertion loss is 1.0 dB. Thepass band return loss is lower than 16.4 dB. The stop bandrejection is dB at 130 GHz. The center frequency is 0.8GHz lower than simulated data, which is about 0.6% leavingfrom 140 GHz. Bandwidth is 0.2 GHz or less, about 98.9% ofdesign target. Return loss is 1.7 dB higher than simulation. Themeasured data coincide well with simulation results.The difference between measured and simulated data is

caused by many reasons. First, the metal loss and surface


Fig. 7. Fabricated 140 GHz BDF with 10 GHz bandwidth (left) and the simulation/measurement results of parameter (right).

Fig. 8. Block diagrams for V-band multiplier of two different ways. (a) Amplified multiplier with power combining. (b) Multiply amplified power combining.

roughness affect the transmission loss. Second, the fabrication(milling and plating) error is added to primary design. Third, thesimulation error cannot be ignored. Finally, the measurementsystem has error about 0.5 dB.V Band Multiplier: The V band multiplier is used as local

oscillator to drive 140 GHz sub-harmonic mixer in both trans-mitter and receiver. Four-way of multiplier power combiningmethod [Fig. 8(a)] was adopted instead of multiply amplifiedpower combining method [Fig. 8(b)] to achieve higher outputpower. Measured output power of the V-band frequency sourceand the corresponding power combining efficiency are shownin Fig. 9.In the 4-way structure, component is an individual mul-

tiplier chip UMS CHU3277-98F. The detailed information canbe seen in the chip datasheet. Key performance parameters forthis chip were tested, and the results indicate that the operatingfrequency range is from 66 to 74 GHz with an average outputpower of 15 dBm (maximum 18 dBm).Through the multiple frequency amplification, THz multi-

plier chain can obtain high stability, low phase noise THz signalfrom the low frequency microwave signal. In order to improvethe power of the V-band frequency source, 4-way multiplierpower combining was developed. Combined analysis of electro-

magnetic field theory and 3-D electromagnetic simulation soft-ware HFSS and ADS, the design performance and efficiency ofthe source were analyzed. The experimental results show thatoutput power is above 20 dBm in range of 66–75 GHz, withmaximum power of 24 dBm at 68 GHz, and the synthetic effi-ciency is above 85%.The 1-way power is measured by PM4 power meter. For con-

venience of the measurements, a microstrip-waveguide transi-tion is designed for the signal chip. The tested results inFig. 9 are powers at the waveguide output instead of chip output,with no consideration of insertion loss in transition. The actualoutput power of the chip in the four-way structure is higherthen the dashed black curve in Fig. 9.

C. Front-End Transceiver Performances Test

The block diagram and integrated module of the front-endtransceiver can be seen in Fig. 10. In the transmitter chain, aKa-band source is octamonic amplified to drive the 140 GHzSHM as local oscillator. The up-converted signal is then am-plified and filtered by 140 GHz MEMS BDF (BPF) to suppressimage frequency. The upper sideband is amplified to about 0.4mW and propagated through the antenna. A 140 GHz isolator isplaced between the amplifier and the BPF to decrease standing


Fig. 9. Measurement results for the V band multiplier.

Fig. 10. Block diagram (top) and integrated module of the front-end transceiver (bottom). In (b), the numbered components are Horn, LNA, Isalator,BPF, SHM, and V-band multiplier.

wave. The reflected signal from the target feeds into the receiverthough a rectangle-horn antenna. The first stage of the receiveris a low noise amplifier to make the noise factor of the receivechain as low as possible. The amplified signal is down-con-verted by an SHM with a same local oscillator as the transmitchain to get an intermediate frequency. The D-band low noiseamplifiers in this transceiver can be obtained commercially. Theminimum gain is about 12 dB in the whole bandwidth from 110GHz to 170 GHz with input power below dBm.

The testing results of transmit and receive chain is showing inFig. 11(a) and (b) respectively. In Fig. 11(a), the transmitter wasdriven by various intermediate frequencies in the pass band withan interval of 0.5 GHz. The saturation output of the transmitteris about 2 dBm with a 0 dBm input IF signal. The whole gainof the transmit chain is about 4–6 dB in linear region for weaksignals. Fig. 11(b) is the gain curve of the receiver in the passband with an interval of 0.5 GHz. The maximum gain is about4 dB and in the frequency band (137.4–142.4 GHz) we used in


Fig. 11. Testing results for front-end transceiver. (a) Output power of the transmitter in the pass-band driving by different IF signal. The blue curve is the saturationoutput with a 0 dBm input IF signal, while the cyan is the gained output for 10 dBm IF signal. (b) Total gain of the receiver chain for weak signal at differentfrequencies.

Fig. 12. Hardware architecture of the data acquisition and processing module in 140 GHz ISAR for real time imaging.

the ISAR radar, the gain is around 3.5 dB with a fluctuation of1 dB.

IV. DATA ACQUISITION AND SIGNAL PROCESSING FORREAL TIME IMAGING

A. Hardware Architecture

The imaging system adopts the real-time high-performancecomputing (RT-HPC) platform which is constituted by GPUand FPGA. The architecture of the RT-HPC platform is shownin Fig. 12. Using virtual instrument technology (NI-VISA) tocontrol the network ports, AWG7122B generates the radar baseband signal. NI FlexRIO FPGA module generates the systemtrigger, then converts the target echo signal to digital data andcompletes the data pre-distortion. The graphics workstation asa remote controller receives the pre-distortion data through NIMXI technology. Finally, the GPU (Tesla 2070) achieves realtime imaging.

B. Signal Process Algorithm

The radar target scattering properties have a great impact withthe electromagnetic wave. The ISAR imaging model for 140GHz wave in this paper satisfies the following condition.

1) Using Born first order approximation, the target backscat-tering can be expressed by the scattering points distributedover the target surface.

2) The target is usually located in the radar far-field, where theelectromagnetic waves can be considered as plane wave.

Satisfying the conditions above, the available ISAR imagingmainly include three categories:1) Fourier transform-based imaging methods;2) super-resolution imaging methods based on modern signalspectrum estimation;

3) non-stationary signal processing-based imaging methods.In the 140 GHz ISAR, the range-Doppler algorithm was used

to get higher imaging speed as mentioned in previous section.L-class Wigner–Ville distribution (LWVD) was adopted to im-prove the cross-range resolution. The detailed processing can beseen in [21].In this application, motion compensation is not needed be-

cause the target position is fixed and the only motion is rotating.But for operational ISAR imaging, the motion compensationprocess is blind because motion parameters of the target are un-known to radar. This makes motion compensation one of themost challenging tasks in ISAR imaging research to obtain afocused image of the target.


C. Migration Through Resolution Cell Correction

For 140 GHz wave imaging, however, scattered migrationthrough resolution cell (MTRC) will take place and the Dopplerinformation will be varying greatly, which will give rise toimage defocusing both in the range and azimuth domain if theconventional Range-Doppler (RD) algorithm. Therefore ISARimaging of 140 GHz wave has become a bottleneck problem.In addition to the RD algorithm, the available imaging al-

gorithms for 140 GHz wave can use the Keystone algorithm,polar format algorithm (PFA) and complex inverse Radon trans-form (CIRT) algorithm. The Keystone algorithm mainly con-cerns about the linear MTRC correction by timescale transformin the wave number domain and applies to moving targets witha small rotation angle. This algorithm no longer holds for therange curve. The PFA interpolates the sector support region ofthe wave number spectrum into a rectangular one and then im-plements two-dimensional inverse Fourier transform (2-DIFT)for image formation according to the central slice theorem. TheCIRT algorithm transforms curves in the range-slow time do-main into peaks with different locations in the image domain,which has the widest observation angle in the three algorithms.In the 140 GHz ISAR, an algorithm called Keystone trans-

formation is used to dispose the time domain signal by dechirpprocessing, and it eliminates the migration through resolutioncell (MTRC) in the target imaging.

V. NONLINEARITY COMPENSATION

With a band width of 5 GHz, the theoretical range resolutionof the 140 GHz imaging radar is 3 cm, but this bandwidth-lim-ited value is only achieved if there is no unwanted modulationin the chirp signal. But actually the measurements results indi-cate that the gains are not flat across the radar bandwidth in bothtransmit and receive chain, which displayed as a modulation inthe chirp waveform. These unwantedmodulations will broad thespectrum peaks and reduce range resolution. To achieve theoret-ical range resolution, the nonlinearity of the radar system shouldbe compensating for.In 140 GHz radar, the dominant source of nonlinearity are the

D-band power amplifiers and the SHM in both transmitting andreceiving chains, which have conversion efficiencies that are notflat across the radar bandwidth, as well as frequency-dependentphase delays. The wideband LFM signal generator is calibratedto get an extremely linear base band input signal for the SHM.The nonlinearity in both transmitted signal and receive chain

will impart distortion to IF signal as well by the SHM downconverting. This distortion can be modeled as perturbations inamplitude and phase . These perturbations will importextra modulations and make the signal for a single point targetno longer a perfect sinusoid. The spectrum is broadened and therange resolution will be degraded.To compensate for the degradation, first a calibration signal is

acquired using a bright target at a range close to the radar’soperating range, and the resulting IF signal is dig-itally stored. All subsequent radar measurements are then di-vided by this calibration waveform prior to spectral analysis.After calibration, a higher range resolution performance is

achieving. As an example, Fig. 13(a), shows the raw, uncom-pensated IF spectral power of a point-like target at about 5 m

Fig. 13. Comparisons of target detection before and after calibration. (a) Anuncompensated raw range spectrum of a point-like target is severely broadenedby chirp waveform nonlinearities with a resolution of about 6 cm. (b) Dividingthe complex IF signal by a calibration waveform focuses the range spectrumand achieves bandwidth-limited 3 cm resolution. (c), (d) are the inverse syn-thetic aperture image the point-like target without and with calibration. Therange profile is compressed by nonlinearity calibration but it has no effect withcross-range profile. (e), (f) are the image of a letter “C” constructed by severalcorner reflectors.

Fig. 14. Photograph of 140 GHz radar imaging scenes.

using the 140 GHz radar. The wide peak near the center of thespectrum is the target, but it is severely broadened by the am-plitude and phase modulation with a resolution of about 6 cm.In contrast, Fig. 13(b) shows the same measurement where the


Fig. 15. A model airplane made by corner reflector and its THz ISAR image.

Fig. 16. A model helicopter covered by tinfoil with a size of about 0.4 m 0.6 m and its THz ISAR image.

signal is first divided by previously-acquired calibration wave-form prior to calculating the spectrum. The broadened peak isnow much narrower with a resolution of about 3 cm.Fig. 13(c) and (d) is the contrast ISAR image for a point-like

target before and after calibration respectively. The range profileis compressed by nonlinearity calibration but it has no effectwith cross-range profile. Fig. 13(e) and (f) is the image of aletter “C” constructed by several corner reflectors before andafter calibration respectively, from which we can see that thecalibration is efficient for scattering points focusing.Higher dynamic range possible and better spectral focusing

is achieved by using nonlinearity compensation. This pre-cal-ibration is not so convenient because the will bedifferent versus environment temperatures, background and soon. A real-time calibration method is under going.

VI. ROTATING TARGETS IMAGING

Two 140 GHz horns are used in both transmitter and receiverto get a large sight view. The 3 dB beam width of the horn isabout 7 deg with 25 dBi gain. The diameter of the high-lightedregion at 6 m is about 0.7 m, which is enough to cover the wholetargets.Fig. 14 shows the schematic diagram of imaging system. The

THz transmitting and receiving chain lie parallel on the deskabout 5 m away from the rotor. The target is fixed on the rotor

and rotating according to the control command from the signalprocessing system.While imaging, the target is rotating around -axis with an

angular velocity of about 10 deg per second. The transmitteremitted 100 s width pulse trains with PRI of 8 kHz. Forone frame, about 1600 pulses were acquired, which made a2000 1600 original data matrix. The data is compensated inFPGA processed in GPU to get a real-time image.The whole data acquisition and processing unit is controlled

by a CPU running NI LabView. The processing is multi-threading with more streamlined coding to reduce computingtime. A 3 cm 3 cm resolution image is achieved while thetarget rotating over 2 degrees in 200 ms, yielding a imagingframe rate of 5 Hz.Figs. 15–17 are the imaging results of the 140 GHz ISAR for

different targets. In the top of Fig. 18, several corner reflectorsare fixed on a foam board to make word “C”, “A”, “E”, “P”,respectively. The distance between centers of two neighborlylocated corner reflectors is about 5 cm and the foam plane has alean of about 45 relative to the vertical plane. The range differ-ence between two neighbor scattering points projected in hori-zontal plane is about 3.5 cm. The four words are imaged withthe 140 GHz ISAR, respectively, and the results showing in thebottom of Fig. 18. The scattering points can be identified defi-nite in the ISAR image, which implies that the range resolutionof this THz radar is about 3 cm. In cross-range profile, the sameresolution can be realized by adjusting rotation angle.


Fig. 17. Corner reflector made “CAEP” (top) and the imaging results (bottom) with the 140 GHz ISAR. The distance between centers of two neighborly locatedcorner reflectors is about 5 cm and the “word plane” has a lean of about 45 acording to the vertical plane.

Fig. 18. Photographs of the 140 GHz Cassegrain antenna and the integrationto radar transceiver.

VII. CONCLUSION AND DISCUSSION

CAEP’s 140 GHz ISAR is effective as a tool for performinghigh resolution imaging in two dimensions (range and cross-range) in the laboratory. But realizing an useful THz imagingISAR operating in a long range, with the potential for even-tual use in the detection field, has required substantial engi-neering, development, and integration of diverse subsystems in-cluding RF electronics, THz components, and digital signal pro-cessing. The most important advantage of ISAR imaging is thatthe imaging resolution is only concerned with the bandwidth ofthe radar but has nothing to do with the target distance. But nowthe output power of the 140 GHz transmitter is about 0.4 mW.Using a 25 dBi gain horn antenna, the operating range is only

about 20 m for target with an RCS of 0.1 m . Progresses on highoutput power sources and larger gain antenna need to be maderecently. In THz source aspect, the 140 GHz multiplier chainmade by CAEP can get an output power of 15 dBm without am-plifier, and this can be used to drive a folded wave-guide trav-eling-wave tube (FWG-TWT) to get a peak power of more than1W. The FWG-TWT is now under designing. Also a Cassegrainantenna with a 0.36 m diameter main reflector has been ma-chined and tested. Testing results imply that the gain is better

than 51 dBi in the frequency region of 130–145 GHz with a 3dB beam width of about 0.4 0.4 . Fig. 18 shows the photo-graph of the 140 GHz Cassegrain antenna. Both the FWG-TWTand the Cassegrain antenna will be integrated to the 140 GHzISAR and the operating range can get kilometer level.

REFERENCES

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Binbin Cheng was born in Hubei, China, onNovember 27, 1980. He received the B.S. degreeand Ph.D. degree in engineering physics fromTsinghua University, Beijing, China, in 2004 and2009, respectively.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 2009. He is now areseacher in Terahertz Research Center of CAEP.His current research involves mmW/Terahertz radarand imaging system.

Ge Jiang was born in Sichuan, China on April 12,1982. He received the B.S. in electrical engineeringand automation, M.S. degree in electronic engi-neering from University of Electronica of Scienceand Technology of China, Chengdu, China, in 2004,and 2007 respectively, and is currently workingtoward the Ph.D. degree in radiophysics at ChinaAcademy of Engineering Physics (CAEP).He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 2007. His research

interests lie in the areas of radar and tomographic imaging theory with applica-tion to mmW/Terahertz imaging.

Cheng Wang was born in Suining, China on March8, 1987. He received the B.S. degree in engineeringphysics from Tsinghua University, Beijing, China, in2008. He received the M.S. degree in radiophysicsfrom China Academy of Engineering Physics(CAEP), Mianyang, China, in 2011.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 2011. His currentresearch involves mmW/Terahertz communicationsystem, solid-state mixers and multipliers based on

Schottky diodes, Terahertz passive filters and circuits.

Chen Yang was born in Chongqing, China onAugust 8, 1983. He received the B.S. degree inelectronic and information engineering from CentralSouth University (CSU), Changsha, China, in 2005.He received the M.S. degree in circuit system fromUniversity of Electronic Science of China (UESTC),Chengdu, China, in 2008.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China in 2008. His current re-search involves signal processing of mmW/Terahertz

radar system.

Yingwu Cai was born in Sichuan, China, onSeptember 6, 1971. He received the B.S. degree inautomation from Chongqing University, Chongqing,China, in 1993. He received the M.S. degree insignal and information processing from Universityof Electronic Science of China (UESTC), Chengdu,China, in 1998.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China in 1993. His currentresearch involves mmW/Terahertz radar/communi-

cation system, signal processing in radar.

Qi Chen was born in Chongqing, China, onNovember 3, 1981. He received the B.S. in remotesensing techniques and instrument, and the M.S.degree in electromagnetic and microwave techniquesfrom Xidian University, Xian, China, in 2003 and2007, respectively, and is currently working towardthe Ph.D. degree in radiophysics at China Academyof Engineering Physics (CAEP).He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 2007. His research

interests include mmW/Terahertz antennas, photo crystal and metamaterials.

Xiang Huang was born in Guangxi, China, on June21, 1985. He received the B.S. degree in electronicengineering from Tsinghua University, Beijing,China, in 2007. He received the M.S. degree incommunication and information system from ChinaAcademy of Engineering Physics, Mianyang, China,in 2010.He has worked as an engineer since 2010 in

Institute of Electronic Engineering (IEE), ChinaAcademy of Engineering Physics (CAEP), Mi-anyang, China, where he is engaged in research on

Micro-system and mmW/Teraertz instruments.


Genghua Zeng was born in Jiangxi, China, onSeptember 27, 1982. He received the B.S. degree inelectronic engineering and information science fromUniversity of Science And Technology of China,Hefei, China, in 2004. He received the M.S. degreein radiophysics from China Academy of EngineeringPhysics (CAEP), Mianyang, China, in 2007.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China in 2007. His currentresearch involves Micro-system and mmW/Teraertz

technology.

Jun Jiang was born in Chongqing, China, on April7, 1987. He received the B.S. degree in engineeringphysics from Tsinghua University, Beijing, Chinain 2011, and is currently working toward the M.S.degree in radiophysics at China Academy of Engi-neering Physics (CAEP).His current research involves mmW/Teraertz

instruments, including solid-state mixers and multi-pliers based on Schottky diodes.

Xianjin Deng was born in Sichuan, China, on June11, 1973. He received the B.S. degree in electronicengineering from Xidian University, Xian, China, in1998, and the M.S. degree in electronic science andtechniques fromUniversity of Electronic Science andTechnology of China, Chengdu, China, in 2006.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 1998. His currentresearch involves mmW/Terahertz communicationsystem, mmW solid-state power combining, and

microwave active circuits.

Jian Zhang was born in Sichuan, China, onNovember 27, 1968. He received the B.S. degree inelectronic techniques from National University ofDefense Technology, Changsha, China, in 1989, andM.S. degree in communication engineering fromChina Academy of Engineering Physics (CAEP),Mianyang, China, in 1994, and the Ph.D. degree inelectrical engineering from Chongqing University,Chongqing, China, in 1998.He joined the Institute of Electronic Engineering

(IEE), China Academy of Engineering Physics(CAEP), Mianyang, China, in 1989. His current research involves electronicsystem, wireless communication, terahertz science and technology.

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