A Five-Phase Brushless DC-Machine Direct Drive Systeminside.mines.edu/~msimoes/documents/pap20.pdf · lap winding construction. ... A Five-Phase Brushless DC-Machine Direct Drive

A Five-Phase Brushless DC-Machine Direct Drive System

EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004 15

Introduction

The majority of electrical drive systems are three-phase systems.Recently some quasi-four-phase systems employing neutral legalso have been used for harmonic optimization and fault-tolerantdrives. Three-phase drive systems have been widely used for years because of the availability of such machines, their inverters,modeling and control. However, polyphase schemes have beenused in the past in drive systems where an induction machine withasymmetric windings has three-phase sets advanced by 30 degreesfor twelve-step industrial applications. Such multiphase drives arelikely to be limited to specialized applications where high perfor-mance and reliability are required (such as EV, HEV, aerospace,ship propulsion and high power applications) and when costrequirements are not so oppressive when compared to the overallenvironment.

The recent literature indicates several advantages for using a mul-tiphase multi-pole electrical machine in hub-wheel systems –high-torque low-speed motors can directly drive systems, avoi-ding mechanical losses incurred by the clutch, reduction and differential gear during power transmission from the motor to thewheels. This work presents the design, analysis, simulation,modeling and control implementation of a high-torque, low-speed, multiphase, permanent magnet, brushless dc-machine. Thepaper focuses on issues regarding the high-level modeling,comprised of a transient model, in conjunction with correspond-ing experimental evaluation. Analyses were made to put togetherthe modeling efforts with the expected behavior in order to haverealistic simulation results verified by the experimental setup;comprehensive experimental results corroborate the work.

Five-phase brusheless dc-machine

A high number of phases yield a smaller magnetic yoke anddecreased volume and weight. However, the number of poles isrestricted physically to the size of the permanent magnets and therotor diameter. Multiple phase arrangement for electricalmachines minimizes torque ripple, increases power density andimproves fault-tolerance in respect to open-circuit legs. The criti-cism against a higher number of phases is because of its morecomplex inverter control scheme and higher cost.

Control complexity is easily managed with the new generation ofDSP controllers targeted to high performance motion systems. A

DSP microcontroller offers the advantages of a single chip systemcombined with the power of a high performance DSP core and isactually the ideal device to implement the complex control lawsrequired for high performance drives where a full set of functionsrequired for ac drive control systems include transfer functions,filter algorithms, and some special ac motor control functions.Loop compensator and matrix vector multiplication that isrequired for state space control and generation can be implementedwithin some hundreds of microseconds.

The second important issue about the high cost of multiphasedrives can be properly addressed with the support of modelingmulti-machine multi-converter systems (MMS) [1]. The MMSformalism is used to build a representation of a system where thepower structure can have several couplings (electrical, magneticand mechanical); there are four conversion structures, namelymono-structures, multi-structures, upstream and downstream.

Within such a MMS analysis and modeling approach severalapplications have been described. A five-leg inverter has beenshown to be able to supply two three-phase induction motors; suchpower structure allows reduced global cost and weight. Otherstructures allow a back-to-back connection of a generator (or athree-phase grid) with a three-phase machine with just one five-phase inverter [2]. The torque control of electric vehicles with separated wheel drives has been recently addressed under theMMS approach [3, 4] and a polyphase cartesian vector approachto control polyphase machines demonstrated that a single inverterwith (2N + 1) phases can independently control (N) ac-machinesconnected in series with appropriate phase swapping [5]. Suchrecent MMS initiatives have been driving forces for furtherresearch and development of multiphase machines as the one presented in this paper.

Analysis and modeling

A radial flux-based motor was designed to be applied as an in-wheel, high-torque, and low-speed motor direct drive system [6].The machine is composed of a rotor with 275 mm of externaldiameter. Magnets are bonded on the internal surface making upthe twelve pole structure [6, 7], where five phases were accom-modated within 60 slots (12 coils for each phase) in a double-layerlap winding construction. With a built-in shaft optical sensor, themotor phase windings are excited sequentially, the optical positionsystem addresses a lookup table, a phase is turned on at the same


M. Godoy Simões, Colorado School of Mines, Golden, Colorado, USAP. Vieira Jr., Federal University of Pará Belém, Brazil

Abstract

The paper describes the design, analysis, simulation, modeling and control implementation of a high-torque, low-speed,multiphase, permanent magnet, brushless dc machine. The main focus is on issues regarding the high-level modeling,comprised of a transient model, in conjunction with corresponding experimental evaluation. The general assumption ofideal rectangular current waveforms for brushless-dc machines is not encountered in practice; the existing distortionscan be modeled by incorporating mutual inductance and armature reaction in order to avoid erroneous control strategydevelopment. Analyses are made to put together modeling efforts with the expected behavior so as to build a model of theexpected behavior so realistic simulation results can be verified. Coherent and consistent results were observed by com-paring simulation and experimentation. A digital signal processing (DSP) system control was developed to implement thestrategies that corroborate the work.

time that one coil leaves the polar section. The permanent magnetflux produces a trapezoidal back-EMF and the currents must becommanded to be ideally in phase with the back-EMF voltages.Table 1 presents the nominal parameters for the machine underconsideration. A Motorola 56824 embedded board is integratedinto the feedback control system as indicated in Fig. 1.

The DSP board was specially designed with a dual-port memorywith shared addresses with the PC host, easing the developmentand high-speed communication needs. The torque control loop isdescribed later. Detailed simulation studies have been performedinitially in order to fully develop the control strategy. The conceptof commutated linear equivalent circuits was applied to themachine, i.e., every 72 electrical degrees two equivalent circuitsvalid for forced and freewheeling conditions were devised. Figs. 2(a) and (b) show the first driving stage where transistors Q1, Q3,Q2 and Q4 are on, impressing excitation for phases A, B, D andE, while there is still current flowing on the machine coils due tothe last driving stage as indicated by the free-wheeling path. Theequations (1) to (7) represent this condition. Such an approach isapplied subsequently to all the phases. Therefore, the modelingwas extended as indicated by the equivalent circuits per stage inFig. 3. A mechanism of switching all the equations, saving initialconditions for next circuit and retrieving the currents from allthose difference equations was implemented in Simulink/Matlab.Even though such a modeling approach has been used in the literature [8-11], other important issues were found for accuratemachine mathematical modeling: the mutual inductance betweenphases and the armature reaction because of the distortion thatoccurs in existing brushless dc-machines [6, 7].

The mutual inductance can be considered by observing how theair-gap flux is composed by all five contributions and building upthe inductance matrix (through experimental parameter identifica-tion of self and mutual inductance parameters). Equation (8)shows that currents have been identified as state variables.Equation (8) also shows that the inductance matrix needs to beinverted and there is one phase with null current every time, i.e.the five-phase system is reduced to a fourth-order system becauseevery 36 electrical degrees there are only four phases conducting,

M. Godoy Simões, P. Vieira Jr.

EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 200416

Fig. 1: Five-phase brushless machine control block diagram.

Table I: Motor ParametersExternal diameter: 275 mmAxial length: 130 mmPower rating: 3.2 HPPoles: 12Phases: 5Nominal voltage: 140 VNominal current: 7.5 ARated speed: 750 rpmRated torque: 30 Nm

Fig. 2: Linear commutated equivalent circuits, (a) Forcedexcitation, (b) Free-wheeling path

a)

b)


while a fifth phase is kept off. For dynamic simulation the induc-tance matrix must be numerically inverted each simulation step-size and a Cholesky decomposition helped the matrix to beexpressed analytically permitting the dynamic simulation. Thesame fourth-order system is equivalent every 36 electricaldegrees, as long as the equations have their variables redefined inaccordance with the flowing currents and the four integrator ini-tial conditions introduced from the previous stage, i.e., each stagehas its own current matrix, back-EMF matrix, mutual matrix, tran-sistor and diode-drop matrix.

As described in [6] and [7], the incorporation of armature reactionin the simulation studies is absolutely necessary in order to have amore realistic system response. From the energy-modeling point ofview, the induced voltage due to the changing electromagnetic

stored energy (back-EMF) delivers mechanical energy. Thus, thetorque for the five-phase BPM machine has to account for the air-gap flux distortion by the armature reaction as shown in Equation(9). The actual experimental evaluation of air-gap back-EMF wasused for torque calculation. Equation (9) needs two lookup tables, atrapezoidal normalized function (Γω) due to the permanent magnetinduced motional voltage, and a triangular normalized function (Λc)for the armature reaction voltage. Therefore, two experimental testsfor back-EMF were required – open-circuit back-EMF in the wholespeed operating range and full-loaded shaft machine in order toobtain Γω and Λc. The experimental lookup tables for obtain Γω andΛc can easily incorporate the effect of rotor position shift in respectto the actual value, a problem that arises in real world. The drivemodel is given with the following equations:


Fig. 3: Equivalent circuits for each commutation state


(1)

(2)

(3)

(4)

(5)

(6)

(7)

where:

– self inductance of phase a, b, c, d and e respectively.

– mutual inductance between phase a and b

– mutual inductance between phase a and c

– mutual inductance between phase a and d

L M L M Mado

dao= ( ) = = ( ) =cos cos144 216 2

L M L M Maco

cao= ( ) = = ( ) =cos cos218 144 2

L M L M Mo oab ba= ( ) = = ( ) =cos cos288 72 1

L L L L L L Maa bb cc dd ee ls= = = = = +

′ = + + + +

′ = + + + +

′ = + + + +

′

e e Li

tL

i

tL

i

tL

i

t

e e Li

tL

i

tL

i

tL

i

t

e e Li

tL

i

tL

i

tL

i

t

e

a a abb

acc

add

aee

b b baa

bcc

bdd

bee

c c caa

cbb

cdd

cee

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

dd d daa

bdb

dcd

dee

e e eaa

ebb

ecc

edd

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

d

= + + + +

′ = + + + +

e Li

tL

i

tL

i

tL

i

t

e e Li

tL

i

tL

i

tL

i

t

d

de

a ei

t Le Ri= −

1 1

3

d

dd

a di

t Le Ri= −

1 1

3

d

d a ai

t Le Ria = − −

1 2

3

d

d e a b d e s qi

t LRi e e e e V Ve = − + ′ + ′ + ′ − ′ − +( )1

44 3 2 4

d

dd

d a b d e s qi

t LRi e e e e V V= − + ′ + ′ − ′ + ′ − +( )1

44 3 2 4

d

d b a b d e s qi

t LRi e e e e V Vb = − + ′ − ′ + ′ + ′ + −( )1

44 3 2 4

d

d a a b d e s qi

t LRi e e e e V Va = − − ′ + ′ + ′ + ′ + −( )1

44 3 2 4

– mutual inductance between phase a and e

– mutual inductance between phase b and c

– mutual inductance between phase b and d

– mutual inductance between phase b and e

– mutual inductance between phase a and d

– mutual inductance between phase e and c

– mutual inductance between phase d and e

ea, eb, ec, ed and ie

– back-EMF developed by phases a, b, c, d and e respectively.

ia, ib, ic, id and ie

– current in phases a, b, c, d and e respectively.

Lls – leakage inductanceM – air-gap inductanceR – stator resistanceVs – inverter voltage supplyVq – diode and transistor voltage drop

(8)

where :

– is the current vector for each phase for stage II

i

i

i

i

abde

a

b

d

e

[ ] =

d

d abde abde

abde abde s q

tI I

M

LM

LR I E V V

[ ] = [ ] − ⋅[ ]

⋅

− [ ]⋅[ ] +

−−

−−

⋅[ ] + [ ]

−

4

1

44

3 1 1 1

1 3 1 1

1 1 3 1

1 1 1 3

1

L M L M Mdeo

edo= ( ) = = ( ) =cos cos288 72 1

L M L M Mceo

eco= ( ) = = ( ) =cos cos216 144 2

L M L M Mcdo

dco= ( ) = = ( ) =cos cos288 72 1

L M L M Mbeo

beo= ( ) = = ( ) =cos cos144 216 2

L M L M Mbdo

dbo= ( ) = = ( ) =cos cos216 144 2

L M L M Mbco

cbo= ( ) = = ( ) =cos cos288 72 1

L M L M Maeo

eao= ( ) = = ( ) =cos cos72 288 1



– is the back-EMF vector for each phase for stage I

– is the inductance matrix for state I.

– is the resistance matrix

– is the identity matrix

[Vs Vq] – is the voltage matrix for transistors and diodes

The torque equation is

(9)Tt

K t tn

K i t tn

i te

v

a n c

n

n 0

=( )

( ) −

+

( ) −

( )

=∑1 5

2

52

4

ω

ω ωΓ

Λ

π

π

I[ ] =

1 0 0 0

0 1 0 0

0 0 1 0

0 0 0 1

R

R

R

R

R

[ ] =

−−

−−

4 0 0 0

0 4 0 0

0 0 4 0

0 0 0 4

M

M M M M M M M M

M M M M M M M M

M M M M M M M M

M M

abde[ ] =

+( ) − +( ) −( ) − +( )− +( ) +( ) −( ) −( )

−( ) −( ) +( ) − +( )− +( )

2 3 2 2 2

2 2 2 2 3

2 3 2 2 2

2

1 2 1 2 1 2 1 2

1 2 1 2 1 2 1 2

1 2 1 2 1 2 1 2

1 2 MM M M M M M1 2 1 2 1 22 3 2 2−( ) − +( ) +( )

E

e

e

e

e

abde

a

b

d

e

[ ] =


Fig. 4: System modeling approach

a)

b)

c)

Fig. 5: Simulation study of displacement effect of positionencoder in electrical degrees for fixed PWM duty-cycle; (a) 3 electrical degrees displacement, (b) 12 electrical degreesdisplacement, (c) 18 electrical degrees displacement.


where :

Te – electromagnetic torque (Nm)ω – electrical speed (rad/s)n – phase contributing with torqueKv – velocity constantΓω – contribution of flux from armature current for the

armature reactionKa – torque constantIn – current at phase nΛc – contribution of flux from permanent magnet for the

armature reaction

The equations (1)–(9) were implemented in Matlab/Simulink; thedifferential equations are algebraically commanded by the rotorangle so as to reinitialize the initial conditions and redefine vari-ables and system matrices in accordance to the absolute angleposition of the rotor. The model needs the machine electricalparameters and the back-EMF experimental characterization aspresented before.

Fig. 4 shows the complete system model where initially an openloop operation was evaluated in order to observe steady stateresponse and allow the understanding of the full operating rangeof such drive. The effect of rotor position displacement was studiedin order to validate the inclusion of armature reaction. Figs. 5 (a)and (b) depict the simulated waveforms for two different con-ditions of displacement showing that for 3 degrees of displace-ment the current can barely be established with a very sluggishresponse, while for 27 degrees the current will establish muchfaster at the expense of strong distortion. An open loop validationof the drive system was performed in order to substantiate thestudy of the variables’ range for a closed loop control. Figs. 6 (a),(b), (c) and (d) show phase current, phase voltage, torque andspeed for open loop operation, imposing a variation in the duty-cycle for all the five phases in the PWM modulator. There is atorque and speed oscillation due to duty-cycle change. The torqueresponse is not optimized since it is an open loop operation, butoverall the system is well behaved with this open loop command.A closed loop system was designed in accordance with the blockdiagram of Fig. 4, and the proportional-integral controller wasfine-tuned by imposing several transient conditions. Fig. 7 shows thesystem response for a torque reference step command from 5 Nmto 20 Nm. The system response is so fast that only two electricalcycles are required for establishing the operation, as indicated bythe current and voltage waveforms. The simulation does not consider any current limitations but in the real application suchhigh current peak will be avoided through protection circuitry. Theoutput torque has an overshoot as indicated in Fig. 7 (c) but shaftand load friction impose a damped speed response, as indicated byFig. 7 (d).

Control system implementation

A custom made Motorola 56824-based DSP board was designedand implemented for this project. The board is represented in Fig. 8.It is connected to a personal computer with the well-documentedISA bus. The description of such a board is outside the scope ofthis paper and is available in [12]. CodeWarrior user-friendly toolswere used for software development and debugging [13] [14]. Adual-access 64K memory on the DSP board, addressed as loca-tions 0 × 030000 through 0 × 3FFFF, provide the backgroundcommunication. Although the PC can read and write other mem-ory areas on the board, it incurs more overhead because the DSPchip must be “held” during PC accesses. The variables or arraysthat are being passed must be defined as global variables. Thoselocations can hold pointers to variables or arrays of pointers;therefore, they are used to communicate between the ProcessorBoards and the PC.


a)

b)

c)

d)

Fig. 6: Open loop behavior. The transient is because of theprescribed PWM duty-cycle variation, (a) Terminal current,(b) Terminal voltage, (c) Machine torque, (d) Machine speed.

The control software that runs in the DSP-PC Board is a multi-tasking, real-time program as indicated by Fig. 9. Therefore, it isnecessary to trigger the interruptions by software in such a waythat lower level interruptions are able to be interrupted by higherlevel ones with automatic context save/restoring of variables. Themost critical time assignment task is the operation of the PWMtimers and the ADC converters. The eight channel ADC conver-sion is triggered at the end of Task 1 and the complete conversionof the four channels takes 32 ms, which is inside the 100 ms setupfor the PWM. Two timers activate the PWM, one decrementingfrom an initial value and another one setting up it again to re-count. For the Motorola DSP56824 with the duty-cycle resolutionranges from 100ns to 13.1msec.

A semaphore handler coordinates Task 1 for acquisition and PWMsetup plus Task 2 running at 1 msec for torque control loop whichcontains a first order IIR filter for torque estimation plus a discretePI controller. The timing is roomy enough for background com-


munications within the control loop structure indicated in Fig. 10.The control loop structure has four main routines: (1) input/outputof analog data, (2) input/output of digital data (PWM five phaseoutput included), (3) signal filtering, and (4) torque control.

The algorithm procedure acquires and filters the five-phase cur-rents, estimates the average value proportional to the instanta-neous torque, estimates torque by using the experimental torqueparameter, computes error with the reference, processes a propor-tional-integral difference equation, and programs the PWM ini-tialization with the PI result bounded to internal scaling.

Before closing the loop, an experimental evaluation, depicted inFig. 11, of the relationship of duty-cycle with respect to the aver-age current was performed so as to have a clear indication of thewide machine range. Fig. 11 is a family of curves taken fromexperimental evaluation of the open loop response of the armaturecurrent in respect to the PWM duty-cycle imposed to the five


a)

b)

Fig. 7: Closed loop behavior. An outer torque loop commands the machine operation (a) Terminal current, (b) Terminal voltage,(c) Machine torque, (d) Machine speed.

c)

d)

Fig. 8: DSP-PC system Fig. 9: Tasks coordination chart for the DSP control code


phase converter for several operating speeds. Such family ofcurves helped to understand the expected range of operation of thedrive system in closed loop control.

A mechanical optimization of the absolute encoder position wascarefully conducted in the laboratory in order to optimize align-ment. Figs. 12 (a), (b), (c) and (d) show the response for amechanical shift of the absolute encoder set at 5 degrees, 10degrees, 15 degrees and 20 degrees. A heuristic procedure wasused in this mechanical adjustment by looking at the best currentrise and fall time response, bounded current spikes, and personalevaluation of vibrating noise. The experimental displacement wasalso used to validate the simulation studies where the misalign-ment was simulated with lookup tables, as previously described.A perfect comparison of the simulated strategy (Fig. 5) with the

real implementation (Fig. 12) was not possible because that wouldrequire much more elaborated instrumentation that is only avai-lable in Fine Mechanics laboratories. However, the overall simu-lated behavior was very close to the observed electrical response,validating our simulation strategy of the optical encoder mecha-nical adjustment.

A torque controller was implemented in the DSP system in accor-dance with Fig. 10; the PI parameters were fine tuned to Kp = 1and Ki = 0.01. Fig. 13 (a) shows a step torque response from 5 Nm to 15 Nm, where the real mechanical torque is presentedwith the reference value, showing a very fast response. Figs. 13(b) and (c) show the steady-state current and voltage waveformsbefore and after the step torque command. The five-phasemachine was connected to a dc-machine with shunt resistances to


Fig. 11: Open loop experimental evaluation of duty-cycle relation withcurrent

Fig. 10: Torque control implementation structure

Fig. 12: Experimental evaluation of currentwaveform due to the optical disc mechanicaldisplacement; (a) 5 electrical degrees displacement,(b) 10 electrical degrees displacement,(c) 15 electrical degrees displacement,(d) 20 electrical degrees displacement

a)

b)

c)

d)


absorb incoming power and the speed was kept within the range650 rpm to 750 rpm. The steady-state voltage and current wave-form indicated in Figs. 13 (b) and (c) confirmed the expectedduty-cycle and average current evaluation in open-loop as indicatedpreviously by Fig. 11.

Conclusion

The paper showed that the design and evaluation of a polyphasebrushless dc-machine direct drive system suitable for high perfor-mance and reliability is critical for applications such as EV, HEV,aerospace and the requirements are not cost oppressive. This workpresented the design, analysis and issues regarding the high-levelmodeling comprised of a transient model in conjunction with theircorresponding experimental evaluation. Analysis was made to puttogether the modeling efforts with the expected behavior so as tohave realistic simulation results verified by the experimentalsetup. The dynamic modeling permitted the control strategydesign and validation with a DSP-based torque loop control andcomprehensive experimental results validated the work.

References

[1] A. Bouscayrol, B. Davat, B. B. De Fornel, B. Francois, J.P.Hautier, J.P. F. Meibody-Tabar, M. Pietrzak-David: Multi-machinemulti-converter system for drives: analysis of coupling by a globalmodeling, Conf. Rec. of IEEE IAS Annual Meeting, Oct. 8–12,2000, Rome, Italy, vol. 3, pp. 1474–1481.

[2] P. Delarue, A. Bouscayrol and B. Francois: Control implementa-tion of a five-leg voltage-source-inverter supplying two three-phase induction machines, IEEE International Electric Machinesand Drives Conference, Villeneuve d'Ascq, June 1-4, 2003, France,vol. 3, pp. 1909–1915.

[3] B. Arnet and M. Jufer: Torque control on electric vehicles with sep-arate wheel drives, EPE '97, Trondheim, Norway, September 1997,pp. 659–664.

[4] B. Hredzak and P.S.M. Chin: Design of a novel multi-drive systemwith reduced torque pulsations for an electric vehicle, PowerEngineering Society Winter Meeting, Jan. 23–27, 2000, Singapore,vol. 1, pp. 208–212.

[5] S. Gataric: A polyphase cartesian vector approach to control ofpolyphase ac machines, Conf. Rec. of IEEE IAS Annual Meeting,Oct. 8–12, 2000, Rome, Italy, vol. 3, pp. 1648–1654.

[6] M. Godoy Simões and Petronio Vieira Jr.: A high torque low-speedmultiphase brushless machine – A Perspective application for elec-tric vehicles, IEEE Trans. on Industrial Electronics, October 2002,vol. 49, no. 5 pp. 1154–1164.

[7] M. Godoy Simões and P. Vieira Jr.: Model development and designof a wheel-motor drive system, Proceedings of EPE – PEMC,September 5–7, 2000, Kosice, Slovak Republic, pp. 74–79.

[8] P. Pillay and R. Krishnan: Modeling of permanent magnet motordrives, IEEE Trans. on Industrial Electronics, November/December 1988, vol. 35, no. 4, pp. 537–541.

[9] P. D. Evans and D. Brown: Simulation of brushless dc drives, IEEProceedings – B, vol. 137, issue 6, November 1990, pp. 299–308.

[10] T. Kenjo and S. Nagamori. Permanent-Magnet and Brushless DCMotors, Oxford Science Publications, 1985.

[11] T. S. Low, K. J. Tseng, T. H. Lee, K. W. Lim, K. S. Lock: Strategyfor the instantaneous torque control of permanent-magnet brush-less dc drives, IEE Proceedings – B, vol. 137, issue 6, November1990, pp. 355–363.

[12] M. Godoy Simões and S. Szafir: A DSP computer platform formechatronics teaching and research, ASEE Computers inEducation Journal, Oct./Dec. 2003, vol. XIII, no. 4, pp. 15–25.

[13] Motorola, DSP 56L811 User’s Manual, Motorola Inc., 1996

[14] Motorola, Motorola DSP 56800 Family Manual, Motorola Inc.,1996

The Authors

M. Godoy Simões, Ph.D., earned the B.Sc. andhis M.Sc. degrees from the University of SãoPaulo, Brazil, and his Ph.D. degree from theUniversity of Tennessee, in 1985, 1990 and 1995,respectively. In 1998, he also received a D.Sc.degree (Livre-Docência) from the University ofSão Paulo. Dr. Simões joined the faculty of theUniversity of São Paulo from 1989 to 2000 andColorado School of Mines in April 2000. He hasbeen working to establish research and educationactivities in the development of intelligent con-trol for high-power electronics applications in

renewable and distributed energy systems. Dr. Simões is currently ser-ving as IEEE Power Electronics Society Intersociety chairman. He isassociate editor of Energy Conversion as well as editor of IntelligentSystems of IEEE Transactions on Aerospace and Electronic Systems.He is also associate editor of Power Electronics in Drives of IEEETransactions on Power Electronics . He has been actively involved in theSteering and Organization Committee of the IEEE 2005 InternationalFuture Energy Challenge. Dr. Simões is IEEE Senior-Member, EPE,IEE and Cigré Member. He was the recipient of a National ScienceFoundation (NSF) Faculty Early Career Development (CAREER) in2002. It is the NSF’s most prestigious award for new faculty members,recognizing activities of teacher-scholars who are considered most likely to become the academic leaders of the 21st century.


Fig. 13: Torque control (a) step response, (b) terminal voltageand current for low torque level, (c) terminal voltage and current for high torque level

a)

b)

c)


Petronio Vieira Jr. received his B.Sc degree(1985) in Electrical Engineering from FederalUniversity of Pará (UFPA), his M.Sc. degree(1996) in Power Electronics from FederalUniversity of Santa Catarina (UFSC) and hisPh.D. degree (2000) in Mechatronics fromUniversity of São Paulo. He was a visitingresearch fellow at Colorado School of Mines,USA, in 2000. Dr. Vieira is member of BrazilianPower Electronics Society (SOBRAEP), theBrazilian Society of Maintenance (ABRAMAN)and member of the Institute of Electrical and

Electronics Engineers. (IEEE). He worked for Centrais Elétricas doNorte do Brasil S.A from 1985 to 1987 in the maintenance of syn-chronous 300MW generators. He has been a faculty member at theComputer and Electrical Engineering Department (DEEC) of FederalUniversity of Pará since 1987. Dr. Vieira has been involved in teachingundergraduate courses in energy conversion and power electronics, andgraduate courses in electronic drives where he coordinates and supervises the research of efficiency issues of electrical drives. He is thehead of the Power Electronics Laboratory of DEEC. His main research interests are control, instrumentation and electronic drives for industrialapplications and systems.


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Deadlines

EPE Journal is published quarterly. Submitted papers undergo areviewing process of about 6 months. Publication dates are as fol-lows:

Vol. 13 no 1: out 15/02/2003, special issue on Matrix Converters

Vol. 13 no 2: out 15/05/2003, normal issue with peer reviewedpapers

Vol. 13 no 3: out 15/08/2003, special issue on Sensorless Drives

Vol. 13 no 4: out 30/11/2002, normal issue with peer reviewedpapers

EPE 2005, 11 – 14 September in Dresden, Germany

http://www.epe2005.comhttp://www.epe-association.org

Call for papers on p. 5 of this issue.Don’t miss the event!

Documents

A Five-Phase Brushless DC-Machine Direct Drive Systeminside.mines.edu/~msimoes/documents/pap20.pdf · lap winding construction. ... A Five-Phase Brushless DC-Machine Direct Drive