A Microcomputer-based Induction Motor Drive_1999

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874 IEEE Transactions on Energy Conversion, Vol. 14, No. 4, December 1999A Microcomputer-based Induction Motor DriveSystem Using Current and Torque ControlChih-Yi Huang Tien-Chi Chen*, IEEE Member Ching-Lien Huang, IEEE Member

Institute of Electrical EngineeringCheng K ung UniversityTainan, Taiw an, R.O.C.Abstract-This paper proposes an induction motor drive withcurrent and torque control. The current control based on the

current error with the current controller yields hi signal. Thetorque control based on the torque error with the torque controlleryields a h, signal. According to th e h, signal, the h, signal andthe appropriate voltage vector is selected by using a look-up tableto control the induction motor drive to obtain a rapid speedresponse. The torque controller, current controller, and d - qframc transform are constructed by the hardware which reduce therunning time of the microcomputer to obtain a high performancedrive. Computer sim ulations and experimental rcsults demonstratethat the proposed method can obtain a high performance inductioninotor drive. Meanwhile, employing the advantages of the addedzero voltage vector to reduce the inverter switching frequencygreatly increasing the efficiency of the inverter.Keywords: current and torque control, voltage vector, look-uptable, switching frequencyI Introduction

To deal with the advances of the power electronics andmicroprocessors [I], the induction motor used in variable speeddrive and position servo control have become more and moreattractive. In the past, dc motors were extensively used in variablespeed, e.g. industrial robots and numerically controlled machinery[8], because their flux and torque can be easily controlled.However, dc motors have certain disadvantages owing to theexistence of the commutator. To overcome these obstacles, theinduction motor has simple and rugged structure, lowmaintainability, and economy etc. Therefore, the induction motorcan be controlled like dc motor by using the space vector field-oriented control approach [2-131. Torque generation is based onthe interaction between the flux and current, in which theinduction motor can be operated at high performance like dcmotor.In the reccnt years, advanced control strategies for PWMinverter-fed induction motor drive based on space vector controlhave' become m ore and m ore popular. To achieve a quickPE-023-EC-0.12-1998 A paper recommended and approved by theIEEE Electric Machinety Committee of the IEEE Power EngineeringSociety far publication in the IEEE Transactions on EnergyConversion. Manuscript submitted J anua ry 6, 1998; made availablefar printing December 17, 1998.

*Institute of Engineering Sc ienceCheng Kung UniversityTainan, Taiwan, R.O.C.response, the conventional adaptive PWM control method ispreferred. Three-independent current controllers track the currentcommand that directly influences the drive performance [4, 5 , io I I ,131. This method is easy to implement, quick response, currentregular, and insensitive to motor parameter variation. However, someproblems arise regarding th e three-independent current controllers: itis difTicult to avoid an increase of the inverter switching frequency,torque ripple, and harmonic losse s [8-131.This paper pres ents a microcomputer-based control with indirectrotor flux orientation and with current a nd torque vector control. Inthe proposed control scheme, coordinate transformation and statorvoltage sensing are unnecessary to reduce the hardware structure.The torque generation is based on the interaction between the statorcurrent and rotor flux command. The torque and current compone ntscan be controlled by selecting the optimum voltage vector switchingtable to restrict the errors of the torque and current within thehysteresis bands, which can reduce the torque and current ripple thatcan indirectly regulate the voltage and rotor flux. Therefore,advantages oft he added zero voltage vector can reduce the switchingfrequency so as to decrease the switching loss. The proposed controlscheme shows a rather simple structure, easy to implement for theinduction motor drive system.2 Configuration of the cont rol system

Figure 1 depicts the proposed control sche me block diagram2.1 Indirect field-oriented control

Part 1 of Fig. 1 presents the indirect field-oriented control of aninduction motor. The electrical dynamics of an induction motor inthe synchronously rotating reference frame (d and q c axis) can beexpressed as

L .r,T P

wherevis , isC, jk4 ,R * , R ,

d e and q' axis stator voltages,d o and axis stator currents,

d o and q' axis rotor fluxes,stator and rotor resistances.

0885-8969/99/$10.00 Q 1998 IEEE


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875RX.3..................................

I .............................................................

WFig. I The proposed control sc hemes block diagram.

Ls, L , , L,r, leakage inductance,p differential operator,0 electrical angular velocity,0 rotor angular velocity,Os we a slip angular fxquency .The generating torque ofthe induction motor is

stator, rotor and mutual inductances,

4

C - - ( 4 ~ , , C - 4 ~ , i ~ )p L (2)4 4The two rotor equations corresponding to the third and fourth rowof equation I ) can be rewritten as

- R, - I b, - o,4: + (-R, + P ) = 0 4)L , L,For field-oriented control [4], the insta ntaneous speed of the rotortlux vector is selected to revolve at the synchronous speed, and thed-axis is aligned. Then, the q*- axi s component of the rotor fluxvanishes and the rotor flux is entirely in the de -axis, i.e.,and = p@ = o ( 5 )

( 6 )Substituting equations ( 5 ) and ( 6 ) nto equations (3). (4) and (2)

b = 4, =constantwhere 4, denotes the rotor flux.yields * R,L, .


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876necessary to take equations (7), (8) and (9) into consideration. Thetorque component of stator current command i;? is computed byusing equation (9). The torque command Te is derived from thespeed controller. The flux component of stator current commandjj,: for the desired rotor flux is computed by using equation 8).For normal operation, as in a dc machine, the current commandii,: remains constant and the torque is varied by the i:,:component. The slip angular velocity command m:., is determinedby using equation (7).

2.2 Generation of the s ta to r cu rren t command)

P a n 2 of Fig. I indicates the generation of the stator currentcommand. The stator current command i: can be expressed as

The phase angle 8' between and the fixed as-axis can beexpressed as

8 = l ( m , + o:, dl+ =0: 6; (11)where 0. = m,-l( j /, j can acce lerate the transient response of thesystem, w, denotes rotor speed which is measured from theencoder.

Three phases stator currents of induction motor ins i andi can be measured by the current sensor and is converted to the d-and q- components in slationary reference frame, iis and i:s as

(12)s '10s = G

The synthesis of the actual stator currents is can be expressed asis = /m (14)2.3 The developed torque

The indirect vector control method de pends o n the generation ofunit vector signals from the rotor flux command 4:. The d andq* axis stator fluxes in the stationary reference, and & canbe computed by the rotor flux command 4: asand

< 4; coso' 15)4 = sin 0 (16)15 ) and 16), the developed torque can berom I Z ) , 13),expressed as

For the microprocessor implementation, equation ( I 1 ) can use thefollowing simple equation to estimate the angular position8 ( k + 1) at (he next sampling instant

for k = l , 2 , 3 , ...... 18)where T is the controller's sampling time. The stator flux in 15)and (16) and the developed torque in (17) can be expressed in thefollowing forms

o * ( k t l ) = o ' ( k ) + o , T

iv(k+ I) = 4 cos(O'(k + 1)) 19)q $ * ( k + l ) = :sin(6 (k+l)) 20)

and3PL,~ : ( k + = -( ,'(k+ I ) i i , ( k + 1) - ; , ( k ~ ) i ; , ( k + )) (21)4 L

1.4 Voltage vectors and PWM controlIn the paper, the voltage source inverter (VSI) is employed for,induction motor drive. Figure 2reveals the diagram of the VSI.Define the spa ce-voltage vector of output phase voltages of the VSIin the stationary frame as

I i iv.

v,

I I I

Fig. 2 The diagram of the VSI

where v vb,, ,, are the phase voltages of the VSI. Accordingto Fig. 2, the VSI opera tes on only two states of each phase at everyinstant. The IGBT controlled the phase voltages V , ,Vbs andV , 22) is expressed as

in which vd is the convelter output voltage's dc value. These threebits 3 ,s ,,ye ) respectively in a digital way denote the states ofthe IGBT of the V SI; I indicates the upper-IGBT to be on and thelower-IGBT off; 0 indicates the upper-1GBT to be off and thelower-IGBT on. For instance v,(l,o,o) means Q ~ , Q ~ , Q ~n andQ,, Q,,Q, off. Equation (23) depicts that the VSI has space-voltagevectors of 2 = 8 state s which are given by

23v, = v,,(O,O,O)= 0, v , = v , ( O , O , l )= - v de ' (+ ' ' )

23v, = vs(l,l,O)= - ~ ~ ~ e ' ' ~ ' ,i = v , ( O , O , O ) = 0Equation (24) indicates that these space-voltage vectors in the

stationaty d" - q' frame, where v , 2 ..., 6 are nonzero-voltage vectors, have the same value vd, ut a phase 2 ~ 1 3different from one another. v o and V 7 are the sam e vector; theiroutput voltage is zero, the so-called zero-voltage vector.


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2 . 5 Torque and c urren t con tro lKefcrring to part 3 of Fig. I , the speed controller generate torque

command c* s compared to the develop ed torque in equation( 2 ) that yields the torque error, c' - T , When the torque error isinputted, the torque controller yields the h, signal. The torquecontroller is a three-level hysteresis compar ator whose output signalh, is determined by the torque error that is defined as

h, = 1, for e* A Th ,= - I , for c ' - c S - A Th , =O , for c*-c=O (25)

where AT denotes the preset torque band. In equation (25) ,11, = 0 represents r,= r,*,. e., when the developed torque r,increases and reaches the torque command r ,t is better to usethe zero voltage vectors v , (O,O,O) or v,(1,1,1) to decrease

T . . Signal h, = 1 implies that r, ST^ - A T , i. e., when thedeveloped torque r, decreases and reaches the lower hysteresislimit ? ; ' -A T. it is better to use the non-zero voltage vectors toincrease T , , By using this technique, the developed torque cantrack the torque command and the torque error is limited withinA T .

The stator current command, i , in (10) is compared to theactual stator currents is n 14) that yields the current error signal,;: - i s he signal is inputted to the current controller and yieldsthe b, signal. The current controller is a two-level hysteresiscomparator whose output signal h, is determined by the torqueerror that is dcfined as

(26)where A i denotes the preset current band. In equation (26),hi = 0 implies i , k ,; + & { 2 , i. e., when the actual statorcurrent i increases and reaches the upper hysteresislimit i, 2 ; + ~ i i 2 ,t is better to select the appropriate non-zerovoltage vector to decrease is. On the other hand, hi = 1represents j , 5 : - ~ i i 2 ,. e., when the actual stator current i,decreases and reaches the lower hysteresis limit i 2 : - A j 1 2 , it isbetter to use the anothcr appropriate non-zero voltage vector toincrease i By using this technique, the developed torque cantrack the torque command and the torque error is limited withinA i .

Selecting the appropriate non-zero voltage vector relies not onlyon the current error i: - , 7 , ut also on the phase angle e ' in (18).The phase angle 8 can be divided into six areas in a circle asfollowing

O L 2--, for N = 1 , 2 , 3 , 4 , 5 , 6 (27)According to (25), (26) and (27), the switching table of voltagevector is listed in Table I . The switching table is implemented by a

h , = l , f o r i ; - i , , k A i / 2h, = 0, for i - i , 5 -A i l2

( N - l ) z N T3 3

877ROM as shown in Fig. 1. The torque error re* T, and the currenterror i; - , are digitized by the three-level hysteresis comparatorand the two-level hysteresis comparator, respectively. A 2-bitsignal h, a I-bit signal h, , and a 3-bit signale: are developed.The hardware of the two-level hysteresis cor

r v18..

Fig. 3 The hardware of the proposed control schemethe three-level hysteresis comparator and a ROM is shown in Fig.3 . The torque error T**-T, and the current error i -i, aredigitized by three-level hysteresis comparator and, A 2-bit signal 00,Oland 10 denote h, = 0 , = 1 and h, = -1, respectively. AI-bit signal 0 and I denote hi = 0 and h, = 1 respectively. A3-bit signal 000, 001, 010, 011, 100 and 101 denotee, , b , 6; 6; 6 and 0; . respectively. For instance, in thc0,. area, if the developed torque T, decreases and reaches thelower limit of the three-level hysteresis comparator, i. e.,r, s c' - AT9 and the actual stator current i, increases and

reaches the upper limit of the two-level hysteresis comparator, i.e., i 2 ; + ~ i 1 2 ,hen h,= 1 and h, =O . In this case, the vectorv 6 ( l , l , 0 ) i s cho sen , t h eo u tp u t o fR O M i s S , = I , S , = ] , S , =O ,

then Q,, Q nd Q,,Qa, Qs f the VSI are turned on and off,respectively.

Table 1 The voltage vector switching table for the proposedcontroller.


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8783 Computer simulations

This section presents a computer simulation for the proposedcontrol scheme. To demonstrate the performance of the proposedcontrol scheme, some simulations of the proposed control schemeare compared to those of the conventional adaptive PWM controlmethod. In general, in the conventional adaptive PWM control offield-oriented control induction motor drive, the current commandsin the synchronously rotating frame, denoted by i and iJ8 mustbe transformed into three phase domain to yield the referencecurrent commands, denoted by j : s , j;, and i : * , using thecoordinate translator and sinlcos generator. The actual statorcurrents of the induction motor, in, i a s ,,, are generated from thePWM inverter by comparing the current commands with threeindependent two-level hysteresis comparators. Under th e simulation,a dc link voltage of IOOV all fed the proposed control schemesinverler and the conventional adaptive PWM control method. Thedead band of the current controller in both the proposed controlscheme and the conventional adaptive PWM control scheme are 0.3A . The proportional and integral gains of speed controller, K , andK in these control scheme are 2.5 and 0.8, respectively. The

parameters o fth e test induction motor ar e& = 1.1n, R = 1.3 Cl,L8 =0,14S H, L ,=O. l45 H, Lm 0 . 1 3 6 H, ~ = 0 . 0 00 6 8K ~ . and number of poles = 2. Figures 4 to 6 summarize the

cornputer simulation results.For a speed command of 600 rpm, the speed response of theproposed control scheme and the conventional adaptive PWMcontrol method are shown in Figs. 4(a) and 4(b), respectively. Figs.4(a) and 4(b) indicate that the speed response of the proposedcontrol scheme is faster than that o ft he conventional scheme. Figs.5(a) and S(b) reveal simulation results of line current o f one phaseand developed torque for the proposed control scheme and theconventional adaptive PWM control method, respectively. Onecan see that the current and the torque ripple of the proposed c ontrolscheme are smaller than those of the conventional adaptive PWMcontrol method. It can be said that the proposed control scheme canachieve current and torque control.Figs. 6(a) and 6(b) reveal the voltage vector distribution andthe switching number of the proposed control scheme and theconventional adaptive PWM control method, respec tively. One cansee that the zero voltage vector is distributed more uniformly in theproposed control scheme. Moreover, to reduce the inverterswitching frequency, zero voltage vectors are considered to add.The proposed control scheme can achieve the switching numberwith approximately 30 more reduction than the conventionaladaptive PWM control me thod.4 Experimental results

4s

The proposed approach in part 1 and 2 of Fig. 1 and part 3 isimplemented by a 80586 personal computer and the circuit shownin Fig. 3, respectively. Fig. 3depicts the motor speed encoderwhich generates the pulse train of 1000 pulseslrev, is convertedinto voltage signal by using the F N 9400 to reflect the motorspeed. Comparators (UI-U4) are applied to construct the proposedtorque controller and a comparator (U5) is employed t o construct acurrent controller. T he output bits of the torque controller, currenthysterisis controller and angular position control bits are createddigital words, which access the address of the ROM (2716) toselect the appropriate voltage vector according to the voltagevector switching table. Operation amplifiers (U6-U7) areemployed to construct a three-phase line current into d- and g

component, denoted by i and i in the stationary referenceframe, respectively. The induction motors parametersa r e R s = l . l n , R , = 1 . 3 C l , L , =O .1 45 H, L,=O.145 H,Lm=0.136 H, J-0.00068 ~ g . m * ,ated power = 1 HP andnumber of poles = 2. The dead band of the current controllers ofthe proposed control scheme is set to 0.3 A. The proportional andintegral gains of the proposed control schemes speed controllerare set to K , =2.5 and K , ~0.8.

Some experimental results of the proposed control scheme aregiven as follows. Figs. 7(a) and (b) reveal the speed response for aspeed command of 600rpm of the proposed control sche me and thcconventional adaptive PWM control method, respectively. One canfind that the proposed control scheme is faster than that of theconventional scheme. The experimental result is similar to thesimulation result in Fig. 4.

q,,

00 0.2 0.4TIME [sec](a)

II0U 0.2 0.4TIME [ s e c ](b)Fig. 4 Speed response for a command of 60 0 rpm. (a) The proposed

control scheme. (b) Conventional adaptive PWM control method.


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879

0 100 200TIME [ms] 0 100 200TIME [ms]

TIME [ms](a)

3 10~3wz -10A I I

0 100 200TIME [ms]i:I__.

P I00 100 200TIME [ms]

b)Pig. 5 Line current and developed torque of the induction motor.(a) The proposed control scheme. (b) Conventional adaptivePWM control method.Fig.

0 100 200TIME [ms](a)

00 100 200TIME [ms]

? 100

0 100 200TIME [ms]b)6 The voltage vector distribution and the switching number. (a)The proposed conu ol scheme. (b) Conventional adaptive PWMcontrol method.


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ss0

Time

. . . . , .. .. . . . .. . .

: : : jTime(b)Fig. 7 Experimental result of speed response. (a) The proposedcontrol scheme. (b) Conventional adaptive PWM controlmethod. (speed scale=200 rp dd iv , time sca le= 100 mddiv).

5 ConclusionsA microprocessor-based VSI-fed in duction motor drive withtorque control loop, current control loop, and speed control loop ispresented herein. With the torq ue control loop and current controlloop, the induction motor can obtain a rapid speed response. Withthe PWM control, the stator current is nearly a pure sinusoidalwave with low harmonics.Simulation results indicate that speed response, the ripple ofcurrent and torque, PWM voltage, and voltage vector distribution

of the proposed control scheme are better than those of theconventional adaptive PWM control m ethod. Moreover, to reducethe inverter switching frequency, it employs the advantages ofadding the zero voltage vector. According to the experimentalresults. the proposed control schemes current controller, torquecontroller, and d - q frame are constructed by the hardwarewhere the purpose is to reduce the running time of themicroprocessor. Moreover, coordinate transformation and voltagesensing arc not required to reduce the hardware structure for theproposed control sche me. The simulation and experim ental resultsindicate that the proposed control scheme has a highly dynamicservo performance.6 References[ I ] B. K. Dose, Recent advances in power electronics, IEEETrans. Power Elec., vol. 7, no. I p. 2-16, 1992.[21 Liu, C. C. Hwu, a nd Y .F. Feng, Modeling and implementationof a microprocessor-based CSI-fed indu ction motor drive usingfield-oriented control, IEEE Trans. Ind. Appi., vol. 25, no. 4,pp, 558-597, 1989.131 J. Holtz, and E. Bube, Field-oriented asynchronous pulse-width modulation for high-performance ac machine driveoperating at low switching frequency, IEEE Trans. In d. Appl.,vol. 27, no. 3, pp. 574-581, 1991.141 A . J. Pollmann, Software pulsewidth modulation for p p

control of ac drives, IEEE Ind. Appi., vol. IA-22, no. 4, pp,691-696, 1986.[ 5 ] D. M. Brod, and D. W. Novotny, Current control of VSI-PWM inverter, IEEE Trans. Ind. Appl., vol. IA-21, no. 4, pp,562-570, 1985.[6 ] L. H. Hoang a nd L. A Dessaint, An adaptive curre nt controlscheme for PWM synchronous motor drives: analysis andsimulation, IEEE Trans. Power Elec. vol. 4, no. 4, 1989.[7] T. C. Chen, and J. S. Chen, A microprocessor-based inductionmotor drives with a novel PWM control, Journal of ControlSystems and Technology, vol. 3, no. I , pp. 71-78, 1995.[8] T. G. Habetler, and D. M. Divan, Control strategies for directtorque control using discrete pulse modulation, IEEE Trans.Ind. Appl., vol. 27, no. 5, pp. 893-901, 1991.[9] I. Takashi, and T. Noguchi, A new quick-res ponse and high-efficiency control strategy of an induction motor, IEEE Trans.Ind. Appl., vol. 1.4-22, no. 5, 1986.[IO] M. P. Kazmierkowski, and A . B. Kasprowicz, Improve directtorque and flux vector control of PWM inverter-fed inductionmotor drives, IEEE Ind. Elec. vol. 42, no. 4, 1995.[ I I] A. Nabae, S . Ogasawara, and H. Akagi, A novel controlscheme for current-controlled PWM inverters, IEEE Trans.Ind. Appl., vol. 1.4-22, no. 4, 1986.[I21 M. P. Kazmierkowski, and W. Sulkowski, A novel vectorcontrol scheme for transistor PWM inverter-fed inductionmotor drive, IEEE T rans. Ind. Elec., voi. 38, no. 1, 1991,[I31 T. Y . Chang, and C. T. Pan, A practical vector controlalgorithm for f l -based induction motor drives using a new

spac e vector current controller, IEEE Trans. Ind. Elec., vol. 41,no. 1, 1994.Tien-Chi Chen received the B.S.E.E. degree from NationalTaiwan Institute of Technology, Taipei, Taiwan, in 1983, theM.S.E.E. degree from the National Taiwan University, Taipei,Taiwan, in 1985, and the Ph.D. degree in electrical engineeringfrom the National Cheng Kung University, Tainan, Taiwan, in1989. He has been with Department of Engineering Science atNational Cheng Kung University. His areas of research are controlof motor drive, neural network and fuzzy control system, andpower e lectronics. Dr. Chen is a member of the IEEE.Chih-Yi. Hu ang received the B.S.E.E. degree from the NationalTaipei Institute of Technology, Taipei, Taiwan, in 1980, the M.S.degree from Institute of Engineering Science at National ChengKung University, Tainan, Taiwan, in 1993. Since 1994, he attendsthe Ph.D. program in the Institute of Electrical Engineering atNational Cheng Kung University. His major interests are control ofser vo drive and power electronics.Ching-Lien Huang received the B.S.E.E. degree from NationalCheng Kung University, Tainan, Taiwan. In 1957, the M.S.E.E.degree from Osaka University, Japan, in 1973. He has beenworking in Department of Electrical Engineering at NationalCheng Kung University since 1964. His areas of reach are controlof motor drive and high voltage engineering.

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A Microcomputer-based Induction Motor Drive_1999