A New Portable 449-MHz Spaced Antenna Wind Proﬁler Radarecee.colorado.edu/microwave/docs/publications/2012/... · LINDSETHet al.:NEW PORTABLE 449MHzSPACED ANTENNA WIND PROFILER

3544 IEEE TRANSACTIONS ON GEOSCIENCE AND REMOTE SENSING, VOL. 50, NO. 9, SEPTEMBER 2012

A New Portable 449-MHz SpacedAntenna Wind Profiler Radar

Brad Lindseth, Senior Member, IEEE, William O. J. Brown, Jim Jordan, Daniel Law, Terry Hock, Member, IEEE,Stephen A. Cohn, and Zoya Popovic, Fellow, IEEE

Abstract—This paper presents the design of a 449-MHz radarfor wind profiling, with a focus on modularity and solid-statetransmitter design. It is one of the first wind profiler radarsto use low-cost Laterally Diffused Metal Oxide Semiconductor(LDMOS) power amplifiers (PAs) combined with spaced antennas.The system is portable and designed for 2–3 month deployments.The transmitter PA consists of three 1-kW peak power moduleswhich feed 54 antenna elements arranged in a hexagonal array,scalable directly to 126 elements. The PA is operated in pulsedmode with a 10% duty cycle at 54% power added efficiency. Theantenna array is designed to have low sidelobes, confirmed bymeasurements. The radar was operated in Boulder, Colorado andSalt Lake City, Utah. Atmospheric wind vertical and horizontalcomponents at altitudes between 200 m and 4 km were calculatedfrom the collected atmospheric return signals.

Index Terms—High power amplifier, push-pull amplifier, windprofiler radar.

I. INTRODUCTION

THE SPACED antenna (SA) wind profiling method [1],[2] is a way to determine the horizontal velocity of the

wind without steering the antenna beam as in the Doppler beamsteering (DBS) method [3]. Advantages of this method includea simpler RF network for the antenna feed and improved timeresolution of the wind velocities. Existing wind profiler systemsat the National Center for Atmospheric Research (NCAR)operate at 915 MHz. Other wind profiling systems such as theNational Profiling Network [4], the MU radar [5], the OQNet[6], the Lindenberg radar [7], the Gadanki MST radar [8] andother commercially built systems [9] use 50, 449, and 482, and915-MHz frequencies. Further review of the current systems,networks, and techniques can be found in [10] and [11]. Higherfrequencies, e.g., 915 MHz are more sensitive to Rayleigh scat-

Manuscript received July 16, 2011; revised November 15, 2011; acceptedDecember 23, 2011. Date of publication March 8, 2012; date of current versionAugust 22, 2012. The National Center for Atmospheric Research is sponsoredby the National Science Foundation.

B. Lindseth is with the National Center for Atmospheric Research, Boulder,CO 80307-3000 USA, and also with the University of Colorado at Boulder,Boulder, CO 80309 USA (e-mail: [email protected]).

W. O. J. Brown, T. Hock, and S. A. Cohn are with the NationalCenter for Atmospheric Research, Boulder, CO 80307-3000 USA (e-mail:[email protected]; [email protected]; [email protected]).

J. Jordan and D. Law are with the National Oceanic and AtmosphericAdministration, Boulder, CO 80305-3337 USA (e-mail: [email protected];[email protected]).

Z. Popovic is with the University of Colorado at Boulder, Boulder, CO 80309USA (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TGRS.2012.2184837

ter from precipitation, while lower frequencies, e.g., 50 MHzare more sensitive to clear-air echoes from temperature andhumidity fluctuations [12].

While vertical wind velocities are found from measuredDoppler shift, the horizontal winds are computed using the SAmethod. Historically the SA technique has been used with largedipole antennas at HF wavelengths to study the ionosphere [13].More recently, the SA technique has been developed to measureatmospheric winds with a simpler antenna topology, whileadding some processing and receiver complexity. It has beensuccessfully used, e.g., at Jicamarca for wind measurementsat 50 MHz using dipole arrays [14], and also at the Adelaideradar [15], and MU radar [16]. This method determines thevelocity by computing the cross correlation between three ormore different receivers using a method called full correlationanalysis (FCA) [17]. As the wind and turbulence move overthe receivers, the velocity can be computed from the time lagmeasured between adjacent receivers. Another disadvantage ofSA is that the signal-to-noise ratio is typically lower than withDBS (empirical observations suggests SA has a 10 dB lowerSNR), which uses the first moment. FCA uses higher ordermoments.

The SA technique offers improved time resolution over DBSbecause the beam is pointed in one direction, while DBSsystems require averaging over at least three different beamdirections. In DBS, the sampling volumes are widely separated(up to 500 m apart at 1-km range), so to satisfy continuityamong the sampling volumes, long time averages (> 10 min)are used. The continuous vertical beam used for SA can alsoallow measurement of boundary layer fluxes of momentum,sensible heat, and latent heat [2]. Another technique that can beused with a SA system is postbeam steering, where the phaseof the SA receiver data is altered in postprocessing so that thereceived beam is steered.

This new radar system is similar to the NCAR multipleantenna profiler radar (MAPR) which is a deployable 915-MHzSA radar [2]. Because of high antenna sidelobe levels, theMAPR antenna requires a ground clutter fence, which consistsof metal panels placed around the perimeter of the antennaand is difficult to deploy because of its large size (about3 m × 3 m square) and mass (about 100 kg). The 449-MHzradar presented here uses an antenna design with lower sidelobelevels that eliminate the need for a clutter fence. While 50-MHzsystems are most sensitive to clear air echoes from temperatureand humidity fluctuations, the size of the antenna arrays at thisfrequency is quite large and not as suitable for portable systems.

0196-2892/$31.00 © 2012 IEEE

LINDSETH et al.: NEW PORTABLE 449 MHz SPACED ANTENNA WIND PROFILER RADAR 3545

Fig. 1. (a) Photograph of the 449-MHz radar wind profiler at Salt Lake City,Utah in November 2010. The receivers are located under the antenna panels.The transmitter and data system are located inside the trailer. (b) Block diagramof the 449 MHz radar wind profiler.

The 449-MHz wind frequency is a good compromise betweenantenna size and clear air wind sensitivity and is chosen for thesystem described in this work and shown in Fig. 1(a).

The requirements for wind profilers are different than thosefor other radars, e.g., precipitation radars [18], because windprofilers receive very little of the transmitted signal. The desiredtarget is Bragg scatter from turbulence [19], which occurs fromirregularities in the index of refraction produced by temperatureand humidity fluctuations. The radar is designed to measureboundary layer winds from an altitude of 100 m to 5 km.

A general block diagram of the 449-MHz wind profiler isshown in Fig. 1(b), while the basic radar parameters are givenin Table I. A pulse is generated by a D/A converter withinthe computer system. The system uses a pulse with a 4-bitcomplementary code [20] and an interpulse period of 50 μs.This pulse is amplified by the transmitter and transmitted on allthree antennas. Each antenna in the block diagram representsa 18-element array, as described in Section II. Each of thethree antennas receives the radar return signal separately. Thesignal from each of the three receivers, described in detailin Section III, is processed separately to compute horizontalwinds, as discussed in Section V. The LDMOS power amplifier(PA) design and characterization is presented in Section IV.

II. MODULAR ANTENNA ARRAY DESIGN

The antenna used in this system is a modular design usingthree 18-element linearly polarized circular patch antennahexagonal arrays (54 elements total). Circular patch antennasin hexagonal arrays have been considered by simulations earlier[21]. This type of antenna array is chosen for the wind profiler

TABLE IRADAR PARAMETERS

due to its low sidelobe levels which minimize ground clutterand allow operation without a clutter fence. The basic unit arrayis the 18-element hexagonal array, shown in Fig. 2(a). This ar-ray can be used as a subarray and lends itself easily to modularand scalable arrays. Fig. 2(b) shows a simulated antenna patternfor the 18-element hexagonal array, using Ansys high frequencystructural simulator [22] for the single element and a standardarray pattern calculation for the array.

The circular patch antennas are probe-fed for linear polar-ization [23] and implemented using single-sided copper FR4circuit board material separated from the ground plane by a13-mm thick Nida-Core H8PP honeycomb material whichhas a relative permittivity of εr = 1.12. The ground plane isa 3-mm thick aluminum sheet chosen for light weight andmechanical stability. Fig. 3(a) shows the construction of oneof the six-element parallelograms that make up the hexago-nal array. The patch feed points are 5.8 cm offset from thecenter for a good match to 50-Ω SubMiniature version A(SMA) connectors. After adding a 25-mm thick polystyreneflat radome above the patches, the return loss of the antennawas measured using a vector network analyzer calibratedto the SMA connector. The match is better than −15 dBat the 449-MHz design frequency, as shown in Fig. 3(b).

Due to the size of the array and the large wavelength, itwas not possible to measure the antenna pattern in an anechoicchamber. To confirm the important low sidelobe levels, mea-surements were made at an outdoor antenna range as shownin Fig. 4(a). The 18-element array was driven by a calibrated18-way splitter network. The sidelobe levels were measuredat horizon (0◦), 12.6◦ and 20◦, limited by the height of thetelephone pole used for the probe antenna in the far field of thearray. The measurement data are shown in Fig. 4(b), confirmingthe low sidelobe levels at the horizon. The measured −30 dBlevel is sufficient to eliminate the need for a ground clutterfence.

After measurements confirming low sidelobe levels in the18-element array, two additional 18-element arrays were


Fig. 2. (a) Eighteen-element hexagonal circular patch antenna array for449 MHz. (b) Antenna pattern simulation. Scale is in dB with reference to mainbeam. Edge of circle is horizon, center is zenith. Sidelobes at horizon are morethan 35 dB less than main beam.

built to complete the three-receiver system with a total of54 elements. The configuration and simulated antenna patternsfor the 54-element array factor are shown in Fig. 5(a) and (b).The size of the arrays was chosen to allow easy transportbetween field projects. After construction of the three arraysand their splitter networks and feed cables, phase and amplitudewere measured at each element to confirm proper phasing of theantenna. This test was accomplished by using a Vector NetworkAnalyzer (VNA) and a near-field test patch positioned aboveeach element in a repeatable manner. Amplitude data from oneof the hexagons are shown in Fig. 6.

III. RECEIVER DESIGN AND IMPLEMENTATION

The receive section is shown in Fig. 7. The receive sectionconsists of a limiter to protect the low noise amplifier (LNA),a 5-MHz filter, and a mixer/IF amplifier stage to downconvertthe 449 MHz to 60 MHz. The 5-MHz filter blocks out RFI andalso keeps out of band noise from aliasing into band during themixing process. The IF amplifier drives the 30-m cable backto the data system and A/D converter. The gain of the receivesection is designed so that the receiver noise will be amplifiedenough that the two least significant bits of the A/D converterwill always be active.

Fig. 3. (a) Photograph of a six-element subarray from Fig. 2(a), showing thepatch probe feed position. A 25-mm thick polystyrene layer is epoxied on topof the panel to protect against moisture. (b) Measured and simulated |S11| ofone of the circular patch antennas.

Fig. 4. (a) Procedure for measuring 18-element antenna array sidelobes.Antenna is rotated about its axis for each elevation angle. (b) Single hexagon,18-element array measured antenna sidelobe levels (receiver polarization ori-ented horizontally). Main beam is pointing at zenith.


Fig. 5. (a) Fifty-four-element hexagonal circular patch antenna array for449 MHz with connections to the transmitter and receivers as shown. The centerto center spacing is D = 231 cm. (b) Antenna pattern simulation. Scale is indB with reference to main beam. Edge of circle is horizon, center is zenith.Sidelobes at horizon are more than 35 dB less than main beam.

Fig. 6. Amplitude check of each element of an 18-element array using a VNA.A theoretical 54-way split is −17 dB.

A receiver noise temperature of 250 K is measured afterthe LNA, this includes sky noise, LNA noise figure, and cablelosses. To calculate required system gain, first the receiver noise

Fig. 7. Block diagram illustrating one of the three receiver channels.

power is calculated using the front end filter bandwidth of5 MHz and the receiver noise temperature of 250 K

PNoise = 10 logkTB

.001= −107.6 dBm.

Because winds and atmosphere have radar returns in the−140 to −150 dBm range, we need additional processing tosee below −107 dBm. The SNR after coherent integration ofthe radar signal is given by

SNR = N · SNRSinglePulse

where N ∼= 128 is the number of coherent integrations. For thisvalue, the coherent averaging gain is about 21 dB. The mini-mum detectible signal-to-noise ratio due to spectral averagingwas determined empirically by [24], [25]

SNRS.Avg. = 10 log25√

NSP − 2.3125 + 170NFFT

(NFFT )(NSP )[dB]

where NFFT = 256 is the number of points in the FFT andNSP = 17 is the number of spectral averages. For these values,the spectral averaging gain is about 16.5 dB. This gives a totalprocessing gain of 37.5 dB, resulting in a minimal detectablesignal of −(107 + 37.5) = −144.5 dBm.

Full scale input for the Pentek 7642 A/D converter is+10 dBm. The SNR in the LTC2255 datasheet is given as71 dBFS at 60 MHz [26]. Thus, the required input power to beabove the A/D noise is +10 dBm − 71 dB = −61 dBm. Thisresults in a required IF gain of at least 107.6 dBm − 61 dBm =46.6 dB.

During the prototype phase, a number of sensitivity testswere conducted. This test involves checking each receiver forsensitivity to a −150 dBm test signal. This test is importantbecause it verifies the performance of the system from theantenna terminal through the receiver to the signal processingsoftware. All of the receivers had a detection threshold near−150 dBm. The receive signals are processed with SA softwarebased on the NCAR Maprdisplay package [27] and furtherprocessed for horizontal winds using Briggs’ FCA method [17].

IV. TRANSMITTER ARCHITECTURE

The transmit section of the system is shown in Fig. 8. Itconsists of a mixer to convert the 60-MHz transmit pulse fromthe D/A converter to 449 MHz. A 10-W driver and 80-Wamplifier stage drive the final amplifier. The core of the transmitsection is the three 1-kW peak PAs. These three amplifiers arecombined using a reactive combiner, the output is transmittedthrough a 30-m heliax cable to a three-way splitter locatedunderneath the outdoor antenna. The output is split to the three


Fig. 8. Diagram illustrating the transmit section of the radar. The 1-kW high power amplifiers are driven by 80-W and 10-W stages.

Fig. 9. Photograph of 80-W amplifier based on a Freescale MRF5S9070Ntransistor. The transistor is mounted underneath a clamp in the center of thecircuit.

hexagonal antennas and then split 18 ways using phase matchedcables to each circular patch antenna.

The MRF5S9070N is a low cost (∼$35) LDMOS transistorcapable of 80-W CW. To design a high efficiency amplifierusing this transistor, Class-E amplifier theory was used [28],[29]. The transistor is modeled as a switch with an outputcapacitance. For class-E operation, a network is added to theoutput of the transistor that forms a low-pass filter so that onlya sinusoidal waveform is seen across the output load. Using theClass-E theory derived in [30], the ideal output impedance toachieve a sinusoidal waveform is given by

Z =0.0446

Csfej49.05

◦[Ω]

Cs can be calculated from the given S-parameters for a transis-tor. In this case, the manufacturer provided a value for Cs in thedatasheet, 34 pF. Using this capacitance value, the calculatedvalue of the output impedance is Z = 2.8 + 3.3j Ω. Because ofthe high output capacitance of this device, a low output circuitimpedance is required.

The substrate used for the 80-W amplifier is Rogers 4350B.The output network was designed for the target impedance,and then a shunt capacitor was used on the output to tune theamplifier for best power added efficiency (PAE) and outputpower, Fig. 9. A PAE of 68% was measured with an outputpower of about 49 dBm, Fig. 10. Additional tests were con-ducted to confirm the phase stability of multiple amplifiers overtemperature. The measured phase variation was a maximum of8 ◦ of phase over the −15C to 40C temperature range. Becauseof the low cost, this transistor was initially considered for usein an active array design with an amplifier located behind each

Fig. 10. Measured 80-W LDMOS Pout, gain, and PAE versus input power.The operating point is Pin = 30 dBm, Pout = 49.1 dBm CW, PAE = 68%,gain = 19 dB.

antenna. As higher power final stages were considered, thisamplifier became a low-cost driver amplifier.

Improvements in LDMOS device technology have enabledkW-level amplifiers. The Freescale MRF6VP41KHR6 [31] wasevaluated for use as a 1-kW peak power pulsed 449-MHz highPA. It is packaged in a push-pull configuration, so that two de-vices are easily combined. A manufacturer test circuit was mod-ified for best gain, efficiency, and output power at 449 MHz.The goal for this application is to have a 1 kW, 10% duty cycle,449-MHz pulse amplifier.

The 1-kW peak pulse amplifier uses lumped element com-ponents. The overall schematic including bias tees is shownin Fig. 11(a). The amplifier layout was fabricated on Rogers4350 substrate. Some of the benefits of a push-pull amplifierare a doubling of the input and output impedances and reducedeven harmonics. Coaxial baluns made of 25 Ω coax are usedto transform the unbalanced 50 Ω input and output impedancesinto lower impedances that are easier to match to the transistor.The input return loss of the balun at 449 MHz is −12 dB(VSWR 1.7:1). The insertion loss of both balun networksmeasured back-to-back is 0.1 dB. The amplifier is tuned for bestgain, PAE, and output power at 449 MHz by changing the valueand position of the capacitors in the input and output matchingnetworks; however, the efficiency at 1-kW output is limitedby the 110-V maximum rating for Vds. The output matchingnetwork is shown in Fig. 11(b). The input matching networkhas a similar topology.

A photograph of the amplifier with a light-weight aluminumheat sink is shown in Fig. 12. A copper insert was installedbelow the transistor to allow more heat transfer between thetransistor and the aluminum heatsink. A transistor clamp madeof Teflon allows for easy test and replacement of the transistorif needed.


Fig. 11. (a) Schematic of the 1-kW (peak) LDMOS pulse amplifier. Coaxialbaluns drive the push-pull transistor pair and combine the output. (b) Outputmatching network of the 1-kW LDMOS amplifier using low impedance mi-crostrip transmission lines.

Fig. 12. Photograph of the 1-kW (peak) pulse amplifier module based on theFreescale MRF6VP41KHR6 LDMOS transistor in push-pull configuration.

Output power, efficiency, and gain of the amplifier as afunction of input power are shown in Fig. 13. The best ampli-fier operating point is Pin (peak) = 41.5 dBm, Pout (peak) =60.1 dBm, PAE = 53.8%, gain = 18.5 dB. Since the final PAstage has 18.5 dB gain, the overall efficiency of the amplifierchain is dominated by that stage. The PAE at 1 kW output islimited by the 110 V maximum rating for Vds. Fig. 14 showsthe amplifier performance versus frequency. The amplifier per-formance is frequency dependent because of the narrowbandnature of the input and output matching networks. Becausethe only modulation of the radar pulse is phase shifts, thebandwidth needed for the amplifier is less than 5 MHz.

Fig. 13. One-kilowatt LDMOS amplifier performance versus input power.Pulsed operation, 10% duty cycle. Best amplifier operating point is Pin(peak) = 41.5 dBm, Pout (peak) = 60.1 dBm, PAE = 53.8%, gain =18.5 dB.

Fig. 14. Measured 1-kW LDMOS amplifier performance versus frequency.Pulsed operation, 10% duty cycle. Amplifier has output power above 59 dBmand efficiency above 44% from 440 to 455 MHz. Input power was reduced to40.38 dBm for this sweep.

Another amplifier requirement is to produce low noise be-tween pulses. To accomplish this simply, the transistor is biasedbelow cutoff with Vgs = 0.9 V. This bias voltage turns thetransistor off between pulses and allows for sufficient gainduring the pulses.

V. SYSTEM MEASUREMENTS

Final tests involved the whole system including both thetransmitter and receiver. The radar was tested for compliancewith ITU requirements [32] and the United States requirements[33]. Pulse shaping is used to limit the bandwidth of thetransmit pulse to the requirement of a −20 dB bandwidth of2 MHz. There are a number of frequency-dependent compo-nents that aid in filtering the 449-MHz pulse bandwidth to the−20 and −40 dB levels and its harmonics down to the −60 dBlevels (e.g., the circulators and splitters). All other requirementssuch as side-lobe suppression, frequency tolerance, and peakEIRP are also satisfied.

Another significant test is a blanker delay test. In a pulsedradar, the blanker switches off the receive signal to the A/Dconverter during the transmit pulse. Connecting the A/D con-verter to the receiver during the radar pulse will saturate theA/D converter. This saturation takes a longer time to recoverfrom than the time it takes to switch the blanker. For this test,


Fig. 15. Measurement of the lowest range gate SNR while varying the A/Dconverter blanker turn-off time. The SNR of the lowest range gates is affectedby the time that the blanker deactivates.

Fig. 16. Boulder, Colorado wind profiler data on 23 October 2010 from449 MHz system using three combined 1-kW amplifiers. The plots showaltitude versus time with the color code indicating SNR for the top plot andvelocity in m/s for the bottom two plots. Precipitation is indicated by strongSNR and negative (downward) vertical velocities.

the system is run normally as a radar during a period with goodatmospheric SNR in the range below 1 km and little or no RFI.Every 2 min, the blanker turn-off time is adjusted using theprofiler control software and a 0-dB SNR level is collected foreach blanker turn-off time. Fig. 15 shows the data from this test.Note that 0-dB SNR was a level chosen for this test. The systemis able to compute winds below 0-dB SNR using additionalaveraging. The goal of this test is to find the optimum blankerturn-off time. As shown in Fig. 15, the SNR at the lower rangegates can be significantly improved with the optimum blankerturn-off time.

A. Wind Data From Boulder, Colorado

As a confirmation of the performance of the new radar, thenew wind profiler radar was first operated during prototypetests at the NCAR Foothills Laboratory in Boulder, CO. Fig. 16shows data from October 23rd, 2010. Precipitation can be seenin the data at 9 UT and between 15 and 18 UT as indicated bythe downward vertical velocities.

Fig. 17. The 449-MHz spaced antenna wind profiler data at West Jordan, Utahon 9 January 2011. (a) Three channel raw signal level versus altitude and timein dB. (b) SNR (dB) and horizontal winds (m/s).

The top plot in Fig. 16 shows signal-to-noise ratio (SNR).Note that SNR is decreased below 1 km because of groundclutter and antenna ringing. SNR is higher during the precip-itation events at 9 UT and 15-18 UT. The middle plot showsthe vertical velocity. The vertical velocity is computed directlyfrom the Doppler shift of the return signal. The bottom plotshows horizontal wind barb data. These horizontal winds are


Fig. 18. Data from commercial 915-MHz Doppler beam steering wind pro-filer located at the same site. SNR (top) and winds (bottom) during the sametime period.

computed by cross-correlations between the receivers using theSA method. The wind direction is indicated by position of thebarb (e.g., a barb pointed downwards with the tail straight up isa wind from the North). The wind velocity is indicated by thenumber of lines at the tail of the barb and also by the color code.

B. Wind Data From West Jordan, Utah

After initial prototype tests in Boulder, the radar was trans-ported and deployed as part of the Persistent Cold Air PoolStudy (PCAPS) in West Jordan, Utah, south of Salt Lake City.A commercially available 915-MHz wind profiler was alsolocated at this site to allow comparison of the data between thetwo profilers. A nearby site contained a SOnic Detection AndRanging (SODAR) wind profiler and radiosonde launch site foradditional comparisons. Data for all systems at PCAPS werecollected from 15 November 2010 until 15 February 2011.

Figs. 17 and 18 show a comparison of data from both windprofilers. The SNR and wind data are quite similar. One issuethat can be seen in the raw signal data is the presence ofRFI. Because wind profiler radars typically have low SNR,RFI is a common problem [8]. RFI is seen in the data as aconstant signal return from all range gates. Because of theposition of the antennas, each antenna sees a different level ofRFI. As an example, RFI is seen in Fig. 17, Signal Channel2 from 12–21 UT. The FCA processing algorithm is normallyable to detect winds in the presence of RFI. Because snow isthe only expected precipitation during this project, a modifiedButterworth filter rejects all vertical Doppler velocities greaterthan 5 m/s. This filter can be modified for other precipitation.

RFI can still affect the SNR as seen at 06 UT. The RFI sourcesare usually communication and pager sources at frequenciessuch as 450 and 451 MHz.

A precipitation event on 9 January 2011 is seen in the datain Figs. 17 and 18 between 05 and 10 UT. The precipitationis indicated by the high SNR signals. A comparison of the449-MHz and 915-MHz profiler data shows that the 449-MHzradar is able to sense winds at a higher altitude than the915-MHz profiler. Higher height coverage can be attributed tothe higher transmit power (500-W peak for the 915-MHz sys-tem versus 3000-W peak for the 449-MHz system). Increasedsensitivity can also be explained by the wavelength dependenceof the effective aperture, which is directly related to SNR. Theeffective aperture of the 449-MHz system was calculated usingan antenna gain simulation to be 10.5 m2, and for the 915-MHzsystem, it is 3.4 m2. The difference in aperture provides about5 dB of difference in SNR. These calculations assume no lossesfrom the antenna feed networks and 100% antenna efficiency.The clear air backscatter cross section is only weakly dependenton wavelength (λ−1/3), so this provides about 1 dB of SNRdifference between 449 and 915 MHz. Total SNR improvementis 13 dB. Given that the SA method has about a 10-dB decreasein SNR, then there is a 3-dB net improvement, and this isconsistent with the data. The data also show that the 449-MHzsystem is able to sense winds down to the 500-m level, while the915-MHz system can sense down to the 200–300 m level. Thereason for this difference is that short transmit pulses were notyet implemented in the profiler control software; this is plannedin future work.

VI. CONCLUSION

A new 449-MHz radar wind profiler has been demonstrated.This system will continue to be deployed in support of NCARscientific field projects. A transmit section and three receiverswere designed and tested. With low noise figure and adequategain, these receivers were tested to detect return signals withpower levels down to −150 dBm.

An 80-W LDMOS amplifier with PAE of 65% and gain of13 dB was demonstrated at 449 MHz and used as a driveramplifier. Three 1-kW amplifiers based on LDMOS transistortechnology were successfully combined to operate as a highpower transmitter for this application. The 1-kW amplifiershave a PAE of 53.8% and a gain of 18.5 dB for operation upto 10% duty cycle.

Future plans include the integration of a PA with phase/amplitude adjustment behind each antenna element. This willrequire evaluation of individual 1-kW amplifiers for phase sta-bility versus temperature. Higher power versions of this systemwith larger 7 and 19 hexagon antennas (126 and 342 elements)are also planned for the future.

ACKNOWLEDGMENT

The authors wish to thank Warner Ecklund, Charlie Martinat NCAR, and also Mike Roberg and John Hoversten at CU,and Nestor Lopez at MIT Lincoln Laboratory for suggestionson this project.


REFERENCES

[1] B. H. Briggs, G. J. Phillips, and D. H. Shinn, “The analysis of observationson spaced receivers of the fading of radio signals,” Proc. Phys. Soc. Lond.Sect. B, vol. 63, no. 2, pp. 106–121, Feb. 1950.

[2] S. A. Cohn, C. L. Holloway, S. P. Oncley, R. J. Doviak, and R. J. Lataitis,“Validation of a UHF spaced antenna wind profiler for high-resolutionboundary layer observations,” Radio Sci., vol. 32, no. 3, pp. 1279–1296,1997.

[3] W. L. Ecklund, D. A. Carter, and B. B. Balsley, “A UHF wind profiler forthe boundary layer: Brief description and initial results,” J. Atmos. Ocean.Technol., vol. 5, pp. 432–441, Jun. 1988.

[4] B. L. Weber, D. B. Wuertz, R. G. Strauch, D. A. Merritt, K. P. Moran,D. C. Law, D. van de Kamp, R. B. Chadwick, M. H. Ackley, M. F. Barth,N. L. Abshire, P. A. Miller, and T. W. Schlatter, “Preliminary evaluation ofthe first NOAA demonstration network wind profiler,” J. Atmos. Ocean.Technol., vol. 7, no. 6, pp. 909–918, Dec. 1990.

[5] S. Fukao, T. Sato, T. Tsuda, S. Kato, K. Wakasugi, and T. Makihira, “TheMU radar with an active phased array system: 1. Antenna and poweramplifiers,” Radio Sci., vol. 20, pp. 1155–1176, Nov./Dec. 1985.

[6] W. K. Hocking, P. A. Taylor, P. S. Argall, I. Zawadzki, F. Fabry,G. Mcbean, R. Sica, H. Hangan, G. Klaassen, J. Barron, and R. Mercen,“A VHF wind profiler network in Ontario and Quebec, Canada Designdetails and capabilities,” in Proc. Amer. Meterol. Soc. Conf., 2009.

[7] J. Nash and T. J. Oakley, “Development of COST 76 wind profiler networkin Europe,” Phys. Chem. Earth, B: Hydrol., Oceans Atmos., vol. 26, no. 3,pp. 193–199, 2001.

[8] V. K. Anandan and D. B. V. Jagannatham, “An autonomous interferencedetection and filtering approach applied to wind profilers,” IEEE Trans.Geosci. Remote Sens., vol. 48, no. 4, pp. 1660–1666, Apr. 2010.

[9] C.-M. Gan, Y. H. Wu, B. M. Gross, M. Arend, F. Moshary, and S. Ahmed,“A comparison of estimated mixing height by multiple remote sensinginstruments and its influence on air quality in urban regions,” in Proc.IEEE IGARSS, Jul. 25–30, 2010, pp. 730–733.

[10] A. Muschinski, V. Lehmann, L. Justen, and G. Teschke, “Advanced radarwind profiling,” Meteorol. Z., vol. 14, no. 5, pp. 609–625, Oct. 2005.

[11] W. K. Hocking, “A review of Mesosphere-Stratosphere-Troposphere(MST) radar developments and studies, circa 1997–2008,” J. Atmos. SolarTerr. Phys., vol. 73, no. 9, pp. 848–882, Jun. 2011.

[12] F. M. Ralph, “Using radar-measured radial vertical velocities to distin-guish precipitation scattering from clear-air scattering,” J. Atmos. Ocean.Technol., vol. 12, no. 2, pp. 257–267, Apr. 1995.

[13] S. N. Mitra, “A radio method of measuring winds in the ionosphere,” J.Inst. Elect. Eng., vol. 1949, no. 11, p. 277, Nov. 1949.

[14] O. Røyrvik, “Spaced antenna drift at Jicamarca, mesospheric measure-ments,” Radio Sci., vol. 18, no. 3, pp. 461–476, 1983.

[15] R. A. Vincent, P. T. May, W. K. Hocking, W. G. Elford, B. H. Candy,and B. H. Briggs, “First results with the Adelaide VHF radar: Spacedantenna studies of tropospheric winds,” J. Atmos. Terr. Phys., vol. 49,no. 4, pp. 353–366, Apr. 1987.

[16] J. V. Baelen, T. Tsuda, A. Richmond, S. Avery, S. Kato, S. Fukao, andM. Yamamoto, “Comparison of VHF Doppler beam swinging and spacedantenna observations with the MU radar: First results,” Radio Sci., vol. 25,no. 4, pp. 629–640, 1990.

[17] B. H. Briggs, “The analysis of spaced sensor records by the corre-lation technique,” in Middle Atmosphere Program Handbook, vol. 13.Chicago, IL: Univ. Illinois, 1984.

[18] K. Le, R. Palmer, B. Cheong, T.-Y. Yu, G. Zhang, and S. Torres, “Onthe use of auxiliary receive channels for clutter mitigation with phasedarray weather radars,” IEEE Trans. Geosci. Remote Sens., vol. 47, no. 1,pp. 272–284, Jan. 2009.

[19] V. I. Tatarski, Wave Propagation in a Turbulent Medium. New York:McGraw-Hill, 1961.

[20] K. Wakasugi and S. Fukao, “Sidelobe properties of a complementary codeused in MST radar observations,” IEEE Trans. Geosci. Remote Sens.,vol. GRS-23, no. 1, pp. 57–59, Jan. 1985.

[21] D. Law, private communication, 2007.[22] Ansys Inc., Ansoft HFSS (High Frequency Structural Simulator) User

Guide, 2009.[23] J. R. James and P. S. Hall, Eds., Handbook of Microstrip Antennas.

London, U.K.: Peter Peregrinus Press, 1989.[24] Vaisala, Inc., Wind Profiling, The History, Principles, and Applications,

Woburn, MA, 2004.[25] A. C. Riddle, L. M. Hartten, D. A. Carter, P. E. Johnston, and

C. R. Williams, “A minimum threshold for wind profiler signal-to-noiseratios,” J. Atmos. Ocean. Technol.. doi:10.1175/JTECH-D-11-00173.1, inpress, 2012.

[26] Linear Technology Corp., LTC2255 Datasheet, 2005.[27] C. L. Martin, W. O. J. Brown, S. A. Cohn, and M. Susedik, “Next gen-

eration spaced antenna wind profiler technology,” in Proc. 11th Symp.Meteorol. Observ. Instrum., Jan. 16, 2001, p. 199.

[28] F. H. Raab, “Idealized operation of the class E tuned power amplifier,”IEEE Trans. Circuits Syst., vol. CAS-24, no. 12, pp. 725–735, Dec. 1977.

[29] T. B. Mader and Z. B. Popovic, “The transmission-line high-efficiencyclass-E amplifier,” IEEE Microw. Guided Wave Lett., vol. 5, no. 9,pp. 290–292, Sep. 1995.

[30] T. B. Mader, E. Bryerton, M. Markovic, M. Forman, and Z. Popovic,“Switched-mode high-efficiency microwave power amplifiers in a free-space power-combiner array,” IEEE Trans. Microw. Theory Tech., vol. 46,no. 10, pp. 1391–1398, Oct. 1998.

[31] Freescale Semiconductor, Inc., MRF6VP41KHR6 Datasheet, 2010.[32] International Telecommunications Union, Recommendation ITU-R

M.1085-1, Technical and Operational Characteristics of Wind ProfilerRadars for Bands in the Vicinity of 400 MHz, 1997.

[33] U.S. Dept. of Commerce, National Telecommunications and InformationAdministration, Manual of Regulations and Procedures for Federal RadioFrequency Management (Redbook), Radar Spectrum Engineering Criteria(RSEC), Section 5.5.5., Washington, DC, 2011.

Brad Lindseth (S’95–M’06–SM’10) received theB.S. degree in electrical engineering fromWashington University, St. Louis, MO. He iscurrently working toward the Ph.D. degree in elec-trical engineering from the University of Coloradoat Boulder, Boulder.

After working as an Engineer on MRI and nuclearmagnetic resonance systems, from 1999 to 2002,he joined the Vestibular Neuroscience Laboratory atWashington University and received an M.S. in elec-trical engineering in 2005 with a thesis on biological

magnetic field sensors in pigeons. From 2005 to 2007, he was with the AtomicDevices and Instrumentation Group at the National Institute for Standardsand Technology, Boulder, where he worked on chip-scale magnetometers. In2007, he joined the Earth Observing Laboratory at the National Center forAtmospheric Research, Boulder, where he is currently an Electrical Engineerworking on 449-MHz and 915-MHz wind profiler radars.

Mr. Lindseth was awarded first place in the 2011 NXP High Power RFDesign Challenge for a 2-kW radar pulse amplifier.

William O. J. Brown received the B.Sc., M.Sc.(Hons.), and Ph.D. degrees from the University ofCanterbury, Christchurch, New Zealand, in 1984,1986, and 1993, respectively.

He did postdoctoral fellowships at Kyoto Univer-sity’s Radio Atmospheric Science Center, Uji, Japanand at the Department of Atmospheric and OceanicSciences, McGill University, Montreal, QC, Canada.Since 1998, he has been at the National Centerfor Atmospheric Research, Boulder, CO. He is aProject Scientist in the Earth Observing Laboratory

and leads the Atmospheric Profiling Group in the In-situ-Sensing Facility.This group deploys integrated sounding system and GPS advanced upper airsounding facilities including wind profilers, radiosondes, and other sensors forthe meteorological research community at locations all around the world. Hisresearch focus is on development and applications of wind profiler radars.

Jim Jordan, photograph and biography not available at the time of publication.

Daniel Law, photograph and biography not available at the time of publication.


Terry Hock (M’81) is an engineer with the NationalCenter for Atmospheric Research. He specializes inthe measurement of winds through global position-ing system dropsondes. He led NCAR’s developmentof this revolutionary device that has extended atmo-spheric profiling capabilities.

Stephen A. Cohn has undergraduate degrees inphysics and astronomy. He received the Ph.D. degreein atmospheric science from MIT. He has been withthe National Center for Atmospheric Research since1994. Prior to becoming the In-Situ Sensing Facility(ISF) Manager in 2005, he was the Chief Scientistof the Integrated Sounding System group. His re-search interests range from advanced measurementtechniques for turbulence and wind to the study ofboundary layer flows in complex terrain.

Zoya Popovic (S’86–M’90–SM’99–F’02) receivedthe Dipl.Ing. degree from the University of Belgrade,Serbia, Yugoslavia, in 1985 and the Ph.D. de-gree from the California Institute of Technology,Pasadena, in 1990.

Since 1990, she has been with the University ofColorado at Boulder, Boulder, where she is currentlya Distinguished Professor and holds the HudsonMoore Jr. Chair in the Department of Electrical,Computer and Energy Engineering. In 2001, she wasa Visiting Professor with the Technical University

of Munich, Munich, Germany. Since 1991, she has graduated 40 Ph.D. stu-dents. Her research interests include high-efficiency, low-noise, and broadbandmicrowave and millimeter-wave circuits, quasi-optical millimeter-wave tech-niques for imaging, smart, and multibeam antenna arrays, intelligent RF frontends, and wireless powering for batteryless sensors.

Dr. Popovic was the recipient of the 1993 and 2006 Microwave Prizes pre-sented by the IEEE Microwave Theory and Techniques Society (IEEE MTT-S)for the best journal papers, and received the 1996 URSI Issac Koga Gold Medal.In 1997, Eta Kappa Nu students chose her as a Professor of the Year. She wasthe recipient of a 2000 Humboldt Research Award for Senior U.S. Scientistsfrom the German Alexander von Humboldt Stiftung. She was elected a ForeignMember of the Serbian Academy of Sciences and Arts in 2006. She was alsothe recipient of the 2001 Hewlett-Packard/American Society for EngineeringEducation Terman Medal for combined teaching and research excellence.

Documents

A New Portable 449-MHz Spaced Antenna Wind Proﬁler Radarecee.colorado.edu/microwave/docs/publications/2012/... · LINDSETHet al.:NEW PORTABLE 449MHzSPACED ANTENNA WIND PROFILER