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ACOUSTICALLY ACTUATED ULTRA-COMPACT
NEMS MAGNETOELECTRIC ANTENNAS
A Dissertation Presented
By
Hwaider Lin
to
The Department of Electrical and Computer Engineering
In partial fulfillment of the requirements
for the degree of
Doctor of Philosophy
in the field of
Electrical Engineering
Northeastern University
Boston, Massachusetts
December 2018
1
Acoustically Actuated Ultra-compact
NEMS Magnetoelectric Antennas
by
Hwaider Lin
W.M. Keck Laboratory for Integrated Ferroics, and Department of
Electrical and Computer Engineering, Northeastern University,
Boston, MA 02115, USA
Committees:
Prof. Nian X. Sun (Advisor)
Prof. Hossein Mosallaei
Prof. Marvin Onabajo
2
ABSTRACT
Antenna miniaturization is one of the fundamental challenges for decades.
Conventional small antennas use electric currents for radiation which relies on
electromagnetic wave resonance that leads to antenna sizes comparable to the
electromagnetic wavelength 0. Here we demonstrated a new antenna mechanism:
Acoustically actuated nanomechanical magnetoelectric (ME) antennas with released
ferromagnetic/piezoelectric thin film resonators, which can generate magnetic currents for
radiation and miniaturize the antenna size magnitude of 1 to 2 orders smaller with one
of the highest gains within all nano-scale passive antennas.
The ME antenna won the NASA Tech Briefs - Create the Future Design Contest: First
Prize (in Electronics/Sensors/IoT Category) with over 800 entries from 60 countries in
2018, which is sponsored by Comsol, Intel, Analog Devices, Mouser Electronics and is
featured in NASA Tech Briefs Magazine with more publicity and exposure to the industry
and investors. The publication in Nature Communications was widely cited in different
news media, including NATURE (Ultra-small antennas point way to miniature brain
implants), SCIENCE (Mini-antennas could power brain-computer interfaces, medical
devices), news on various websites and newspapers in different languages, TV interview.
These nano-antennas with ultra-high sensitivity, high selectivity of the magnetic field,
the integrated capability to CMOS technology, and ground plane immunity from the human
body, have great potential applications for bio-implantable antennas, biomedical
applications, internet of things, etc.
3
ACKNOWLEDGMENTS
I especially want to attribute all my achievements to my research advisor, Prof. Nian
X. Sun, for his mentorship in my doctoral study. He is the one that develops my knowledge
and problem-solving skills, and I appreciate all these efforts that led to my success.
I want to thank Prof. Hossein Mosallaei and Prof. Marvin Onabajo for being my
committees. I enjoyed the time with both professors as their teaching assistant.
I want to thank Prof. Greg Carman from TANMS in UCLA. I had a great learning
experience with him in TANMS since 2015. I also want to thank Prof. David Cheng from
California State University in Fullerton. We have known each other since 2009, and he is
always guiding me to the correct academic and career path since then. I appreciate both
professors being my reference contacts for my career.
I want to thank all the collaborators from different universities, Air Force Research
Laboratory, DARPA, NSF, and Raytheon. I would also like to show my great appreciation
for Dr. John Gianvittorio from Raytheon, Dr. Brandon M. Howe, and Dr. Michael E.
McConney both from AFRL that willing to be my reference contacts in the future.
I want to thank all my colleagues from Prof. Nian X. Sun’ group for providing me an
excellent atmosphere for research. Special thanks to Dr. Tianxiang Nan who co-authored
the first ME antenna paper with me in Nature Communications.
Finally, I would like to show my enormous gratitude to my parents and brother for
their unconditional love and meticulous support on my life. I have to say, without my
family, I will not have so many accomplishments. Thank you so much, and I love you!
4
CONTENTS
1 Introduction 12
1.1 Multiferroics and Magnetoelectric Coupling (ME)…………………………....12
1.2 The Scope of the Dissertation…………………………………………………17
1.3 Reference……………………………………………………………………...18
2 Experimental Methods 27
2.1 Thin Film Deposition………………………………………………………….27
2.2 Magnetic Hysteresis Measurements………………………………………….30
2.3 Ferromagnetic Resonance Spectrometer……………………………………...33
2.4 Reference……………………………………………………………………...39
3 Acoustically Actuated Ultra-compact NEMS Magnetoelectric Antennas 40
3.1 New Mechanism....……………………………………………………………40
3.1.1 Motivation……………………………………………………………….40
3.1.2 Theory……………………………………………………………………42
3.1.3 Modeling…………………………………………………………………45
3.1.4 Micro-fabrication………………………………………………………...49
3.2 Experimental Data…………………………………………………………….54
3.2.1 Magnetoelectric Coupling Demonstration……………………………….54
3.2.2 Modified Equivalent Circuit Modeling…………………………………...60
3.2.3 Magnetic Sensitivity……………………………………………………...62
3.2.4 Far Field Measurement…………………………………………………...66
3.3 Discussion…………………………………………………………………….72
5
3.3.1 Frequency Capability…………………………………………………….72
3.3.2 Antenna Gain...………………………………………………………….73
3.3.3 Antenna Efficiency……………………………………………………….76
3.3.4 ME Antenna Arrays…...…………………………………………………78
3.3.5 Minimalization Techniques………………………………………………80
3.4 NanoNeuroRFID……………………………………………………………...82
3.4.1 Research Strategy………………………………………………………...82
3.4.2 Proposed Architecture……………………………………………………84
3.4.3 Innovation………………………………………………………………...86
3.5 Summary……………………………………………………………………...88
3.6 Reference……………………………………………………………………...90
4 NEMS ME Bandpass Filters with Dual E- and H- Field Tunability 101
4.1 Introduction………………………………………………………………….101
4.2 Design and Fabrication………………………………………………………103
4.3 Results and Discussion……………………………………….……………108
4.3.1 Modified Equivalent Circuit Modeling…………………………………108
4.3.2 Delta E Effect…………………………………………………………110
4.3.3 Magnetoelectric Coupling………………………………………………112
4.3.4 Band-pass Filter Performance…………………………………………116
4.4 Summary………………………………………………………………….117
4.5 Reference……………………………………………………………………118
5 Conclusions 119
6
LIST OF FIGURES
1 Introduction
1-1 (a) Electric field dependence of the transmission coefficient (S21) spectra of
FeGaB/PZNPT (b) Magnetic hysteresis loops of the FeGaB/PZN-PT multiferroic
heterostructure under different external electric fields measured by VSM………….14
1-2 FMR spectra of Terfenol-D/PZN-PT at E=0 kV/cm (blue) and E=6 kV/cm (red)…15
2 Experimental Methods
2-1 Schematic of magnetron sputtering.
(http://www.nims.go.jp/mmu/tutorials/sputtering.html)..............................................27
2-2 X-ray reflectometry (XRR) spectra with different thin films thicknesses
(https://e-reports-ext.llnl.gov/pdf/799501.pdf)............................................................29
2-3 (a) Schematic of vibrating sample magnetometry (VSM). (b) Typical magnetic
hysteresis loop of FeGaB/PZT multiferroic heterostructure with different applied
electric fields…………………………………………………………………………32
2-4 (a) Schematic of Zeeman effect. (b) EPR spectrum with showing Hres and ∆Hpp.
Insect shows the absorption spectrum……………………………………………….36
2-5 (a) Schematic of a home-built broadband FMR system. (b) Resonant linewidth ∆Hpp
as a function of frequency for NiFe. (c) Resonance field HFMR as a function of
frequency for NiFe………………………………………………………………….38
7
3 Acoustically Actuated Ultra-compact NEMS Magnetoelectric Antennas
3-1 Illustrations of the nano-plate resonator (NPR) and thin film bulk acoustic wave
resonator (FBAR)……………………………………………………………………43
3-2 (a) Illustration and explanation of the new antenna mechanism. (b) Illustration and
explanation of the ground plane effect………………………………………………44
3-3 (a) Comsol direct magnetoelectric coupling simulation process flow. (b) Schematic
of the magnetoelectric nanoplate resonator (NPR) and the induced ME voltage
simulation. The RF field (HRF) is generated by an RF coil………………………….46
3-4 The structure and layers using five-masks micro-fabrication process flow of the ME
antenna……………………………………………………………………………….49
3-5 (a) Magnetic hysteresis loop and (b) Ferromagnetic resonance spectrum of
FeGaB/Al2O3 multilayers……………………………………………………………51
3-6 (a) The optical images of the fabricated NPR and FBAR. (b) The scanning electron
microscopy (SEM) image of the fabricated NPR and FBAR……………………….52
3-7 (a) Measured admittance curve of the ME NPR. (b) Simulated admittance curve of
the ME NPR. The inset indicates a contour extensional mode of vibration at resonance
with the applied RF voltage signal………………………………………………….55
3-8 (a) Calculated ME coupling coefficient (left axis) and the Measured induced ME
voltage (right axis) versus the frequency of HRF excitation. (b) Simulated induced ME
voltage……………………………………………………………………………….56
8
3-9 (a) Measured admittance curve of the non-magnetic NPR. (b) Measured induced
ME voltage versus the frequency of HRF excitation………………………………….58
3-10 The Modified Butterworth–van Dyke (MBVD) model………………………….61
3-11 (a) ME coupling coefficient αME of ME sensor as a function of bias DC magnetic
field (x-axis) and the RF driving frequency (y-axis). The dashed curve exhibits the
resonance frequency (highest intensity at each frequency sweep) versus bias magnetic
field. The bias magnetic field was swept from -5 mT to 5 mT. (b) The hysteresis loop
of αME obtained by sweeping the magnetic field back and force. The inset shows the
schematic of the ME NPR with the external bias magnetic field applied along its length
direction……………………………………………………………………………...63
3-12 Induced ME voltage as a function of magnetic field at the excitation frequency of
60.7 MHz (red) and 1 MHz (blue) indicates the detection limit…………………….65
3-13 (a) Return loss (S22) of ME FBAR. The inset shows the simulated out-of-plane
displacement of the disk at the resonance peak position. (b) Return loss (S22) curve of
the non-magnetic Al/AlN control FBAR…………………………………………….67
3-14 (a) Transmitting and receiving behavior (S12 and S21) of ME FBAR. (b) S12 and S21
of the non-magnetic Al/AlN control FBAR…………………………………………68
3-15 Antenna polar normalized gain charts: (a) and (b) The out-of-plane axis with in-
plane rotation; (c) and (d) The in-plane axis perpendicular to the ME antenna anchor
direction with out-of-plane rotation; (e) and (f) The in-plane axis along the ME antenna
anchor direction with out-of-plane rotation. The sinusoidal wave along 0 or 180
9
direction denotes the propagating H-field component of the EM waves……………70
3-16 Measured Resonance frequency as the function of one over width (1/w) for NPR
resonators and one over thickness (1/t) for FBAR resonators………………………..72
3-17 Simulated reflection coefficient (S11) of the small loop antenna. The inset shows the
schematic of the simulated small loop antenna………………………………………74
3-18 (a) Comparison of simulated induced voltages from a radio frequency magnetic
excitation with among one to three resonators arrays in series. (b) Comparison of return
loss of ME antenna with three resonators arrays in parallel for achieving broadband
performance. Insets show the displacement at resonance which indicates the resonate
mode of the ME antennas……………………………………………………………79
3-19 The maximum dimension of miniaturized antennas with different techniques vs.
frequency…………………………………………………………………………….81
3-20 Schematic of the wireless implantable nanoscale neural radio frequency
identification (NanoNeuroRFID) system with a bi-directional communication link for
a capacity of 100~1000 implanted elements………………………………………… 83
3-21 (a) Proposed architecture of the implantable NanoNeuroRFID with energy
harvesting, clock source, and RF transmission capability. (b) The architecture of the
RF transceiver for external wireless power transfer and time-shared neural
recording…………………………………………………………………………….84
10
4 NEMS ME Bandpass Filters with Dual H- and E- Field Tunability
4-1 Schematic of the layered structure of the NEMS ME band-pass filter…………103
4-2 (a) Simulated admittance amplitude curve of the NEMS coupled ring-shaped FBAR
resonator showing the electromechanical resonance frequency of ~92MHz. (b)
Simulated signal transmission from port 1 to port 2………………………………104
4-3 The fabrication process of NEMS ME band-pass filter…………………………106
4-4 Optical and SEM images of the fabricated NEMS ME band-pass filter with silicon
substrate released…………………………………………………………………107
4-5 (a) Measured Admittance curve and Butterworth–van Dyke (BVD) model fitting of
the fabricated NEMS magnetic field resonator. (b) The BVD equivalent electrical
circuit of the resonator………………………………………………………………109
4-6 (a) Measured Admittance curve at various bias DC magnetic fields. (b) Resonance
frequency and peak admittance amplitude as a function of the DC magnetic field…111
4-7 Resonance frequency as a function of DC Bias Voltage…………………………113
4-8 NEMS ME band-pass filter measured return loss S11 and insertion loss S21 at zero
bias field………………………………………………………………………….114
4-9 (a) S11 performance, and (b) S21 performance with different dc magnetic fields…115
11
LIST OF TABLES
1 Introduction
1-1 Classification of Magnetoelectric Coupling……………………………………….13
3 Acoustically Actuated Ultra-compact NEMS Magnetoelectric Antennas
4-1 Key Features of Conventional and ME Antennas…………………………………41
4-2 Miniaturized UHF Antennas Comparison…………………………………………89
12
1. Introduction
1.1 Multiferroics and Magnetoelectric Coupling (ME)
Both single-phase multiferroic materials and composites, which have two or more of
the ferroic properties (ferroelectricity, ferromagnetism, and ferroelasticity), provide great
interest and numbers of multiferroic devices [1]-[8] using different magnetoelectric (ME)
coupling. Multiferroic composites have ME coupling coefficients several orders of
magnitude higher than those of single phase multiferroics, which provide effective energy
conversion between electric and magnetic fields [9]-[14]. Table 1-1 shows the categorized
multiferroic devices based on the control mechanisms. Direct ME coupling (magnetic field
control of electrical polarization) can be used in energy harvesters, and magnetometers, etc.
[15]-[20]. Converse ME coupling (E-field control of magnetization switching) has been
used in Spintronics, ME Random Access Memory, and others [21]-[26]. Converse ME
coupling (E-field control of the magnetic permeability and spin waves) can be used in
voltage tunable inductors, tunable bandpass filters, tunable phase shifters, etc. [27]-[50].
High permittivity and high permeability multiferroic materials with high permeability and
high permittivity provide great opportunities for miniature antennas or other
RF/microwave devices [51]-[56]. Finally, the combination of direct and converse ME
coupling is required to realize magnetoelectric antennas used for both receiving and
transmitting behaviors [57].
Achieving strong ME coupling in multiferroic heterostructures becomes more critical
to get large tunability in multiferroic devices in which the performance is mainly dependent
13
on the multiferroic materials [58]. Significant progress has been made in multiferroic
heterostructures to support the ME coupling concept such as a large ferromagnetic
resonance frequency voltage tunability from 1.75 GHz to 7.57 GHz in a FeGaB/PZN-PT
(Pb (Zn1/3Nb2/3) O3–PbTiO3) heterostructure.
Fig. 1-1 (a) shows electrical tuning of transmission coefficients S21 in FeGaB/PZN-
PT heterostructure by sweeping the frequency with a network analyzer. The peak of the
curve shows the absorption when increasing the voltage, representing the electric field
dependence of FMR frequency with 5.82GHz tunable FMR frequency range. Notice that
in the PZN-PT single crystal, the rhombohedral phase transition to the tetragonal phase
causes the huge jump of the resonant frequency change between 5.8 and 6 kV/cm electric
field. The minimum frequency change above 6 kV/cm electric fields was consistent with
the strain-electric field relation of the PZN-PT single crystal [59]. A positive Heff along the
Table 1-1: Classification of Magnetoelectric Coupling
ME coupling Physical mechanism used for devices
Direct ME coupling H-field control of electric polarization
Converse ME Coupling
E control of magnetization switching
E control of permeability μ
E control of spin wave
No ME Coupling needed High μ and high ε
14
Fig. 1-1. (a) Electric field dependence of the transmission coefficient (S21) spectra of FeGaB/PZN-PT (b)
Magnetic hysteresis loops of the FeGaB/PZN-PT multiferroic heterostructure under different external
electric fields measured by VSM.
15
in-plane [01-1] direction was produced from the induced tensile strain in PZN-PT (011)
and the positive magnetostriction of FeGaB. With the applied magnetic field along the [100]
direction, the E-field dependence of magnetic hysteresis loops was shown in Fig. 1-1 (b).
A massive change in saturation magnetization from up to 700 Oe with the applied
electric field is due to the E-field induced compressive strain along [100] direction and the
significant positive magnetostriction, resulting in a negative Heff along this direction [60].
In Fig. 1-2, the largest E-field-induced Heff of 3500 Oe was achieved in Terfenol-D/PZN-
Fig. 1-2. FMR spectra of Terfenol-D/PZN-PT at E=0 kV/cm (blue) and E=6 kV/cm (red).
16
PT (011) structure due to the large magnetostriction constant in the magnetic phase [61].
The giant tunable magnetic anisotropy of multiferroic heterostructures provide excellent
opportunities for reconfigurable microwave multiferroic devices with ultra-low power.
17
1.2 The Scope of the Dissertation
⚫ Experimental Methods:
This section introduces different main experimental techniques to characterize
multiferroics materials for the AlN-based ME antenna.
⚫ Acoustically Actuated Ultra-compact NEMS Magnetoelectric Antennas
The main section of the dissertation is to introduce a new antenna approach
coupling the acoustic wave and the electromagnetic wave for magnetic currents
radiation. At the end of the section, a biomedical application (NanoNeuroRFID) for
the ME antenna will be conceptually introduced.
⚫ NEMS Bandpass Filters with Dual E- and H- Field Tunability
NEMS bandpass filters with ME resonator will be introduced in this section
showing other device using multiferroics and ME coupling with similar structure.
⚫ Conclusion
The final section will summarize the work of this dissertation.
18
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23
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43 X. Yang, Y. Gao, J. Wu, S. Beguhn, T. Nan, Z. Zhou, M. Liu, and N.X. Sun: Dual H-
and E-Field Tunable Multiferroic Bandpass Filter at Ku Band Using Partially
Magnetized Spinel Ferrites. IEEE Trans. Magn. 49, 5485 (2013).
44 M. Li, Z. Zhou, M. Liu, J. Lou, D. E. Oates, G. F. Dionne, M. L. Wang, and N. X.
Sun: Novel NiZnAl-ferrites and strong magnetoelectric coupling in NiZnAl-
ferrite/PZT multiferroic heterostructures. J. Appl. Phys. 46, 275001 (2013).
45 X. Yang, J. Wu, J. Lou, X. Xing, D. E. Oate, G. F. Dionne, and N. X. Sun: Compact
tunable bandpass filter on YIG substrate. Electron. Lett. 48, 1070 (2012).
46 Z. Zhou, S. Beguhn, J. Lou, S. Rand, M. Li, X. Yang, S. D. Li, M. Liu, and N. X. Sun:
Low moment NiCr radio frequency magnetic films for multiferroic heterostructures
with strong magnetoelectric coupling. J. Appl. Phys. 111, 103915 (2012).
24
47 J. Lou, G. N. Pellegrini, M. Liu, N. D. Mathur, and N. X. Sun: Equivalence of direct
and converse magnetoelectric coefficients in strain-coupled two-phase systems. Appl.
Phys. Lett. 100, 102907 (2012).
48 G. M. Yang, O. Obi, G. Wen, and N. X. Sun: Design of tunable bandpass filters with
ferrite sandwich materials by using a piezoelectric transducer. IEEE Trans. Magn. 47,
3732 (2011).
49 J. Lou, M. Liu, D. Reed, Y. H. Ren, and N. X. Sun: Electric field modulation of surface
anisotropy and magneto-dynamics in multiferroic heterostructures. J. Appl. Phys. 109,
07D731 (2011).
50 M. Liu, O. Obi, J. Lou, S. Stoute, Z. Cai, K. Ziemer, and N. X. Sun: Strong
magnetoelectric coupling in ferrite/ferroelectric multiferroic heterostructures derived
by low temperature spin-spray deposition. J. Phys. D 42, 045007 (2009).
51 G. M. Yang, O. Obi, and N. X. Sun: Enhancing ground plane immunity of dipole
antennas with spin–spray‐deposited lossy ferrite films. Microw. Opt. Technol. Lett.
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52 G. M. Yang, X. Xing, A. Daigle, O. Obi, M. Liu, J. Lou, S. Stoute, K. Naishadham,
and N. X. Sun: Planar annular ring antennas with multilayer self-biased NiCo-ferrite
films loading. IEEE Trans. Antennas Propag. 58, 648 (2010).
53 G. M. Yang, X. Xing, O. Obi, A. Daigle, M. Liu, S. Stoute, K. Naishadham, and N.
X. Sun: Loading effects of self-biased magnetic films on patch antennas with
substrate/superstrate sandwich structure. IET Microw. Antenna Propag. 4, 1172
(2010).
25
54 G. M. Yang, X. Xing, A. Daigle, M. Liu, O. Obi, S. Stoute, K. Naishadham, and N.
X. Sun: Tunable miniaturized patch antennas with self-biased multilayer magnetic
films. IEEE Trans. Antennas and Propag. 57, 2190 (2009).
55 G. M. Yang, R. H. Jin, G.B. Xiao, C. Vittoria, V. G. Harris, and N. X. Sun:
Ultrawideband (UWB) antennas with multiresonant split-ring loops. IEEE Trans.
Antennas Propag. 57, 256 (2009).
56 G. M. Yang, A. Daigle, M. Liu, O. Obi, S. Stoute, K. Naishadham, and N. X. Sun:
Planar circular loop antennas with self-biased magnetic film loading. Electron. Lett.
44, 332 (2008).
57 T. Nan, H. Lin, Y. Gao, A. Matyushov, G. Yu, H. Chen, N. Sun, S. Wei, Z. Wang, M.
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Acoustically actuated ultra-compact NEMS magnetoelectric antennas. Nat. Commun.
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60 M. Liu, S. Li, Z. Zhou, S. Beguhn, J. Lou, F. Xu, T. J. Lu, and N. X. Sun: Electrically
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multiferroic heterostructures. J. Appl. Phys. 112, 063917 (2012).
26
61 J. Lou, D. Reed, M. Liu, and N. X. Sun: Electrostatically tunable magnetoelectric
inductors with large inductance tunability. Appl. Phys. Lett. 94, 112508 (2009).
27
2. Experimental Methods
2.1 Thin Film Deposition
High-quality thin films were deposited using Physical Vapor Deposition (PVD) with
both DC and RF sources depending on thin film properties. The magnetron sputtering
manufactured by AJA International were used. It consists of the main chamber and a load
dock chamber, which enables a fast sample transferring. A low background pressure below
8 × 10−8 Torr can be reached in the main chamber using a CTI cryopump that operates at
11 K. The sputtering gas of ultrahigh purity Ar, N2, and O2 can be provided for metal,
Fig. 2-1. Schematic of magnetron sputtering. (http://www.nims.go.jp/mmu/tutorials/sputtering.html)
28
nitrides and oxides deposition respectively. The sputtering pressure is usually maintained
at 3 mTorr using a programmable mechanical valve with a fixed gas flow between 15-20
standard cubic cm/min (sccm). It is equipped with six sputtering sources including four
DC guns and two RF guns. 2-inch diameter targets which can be operated separately or
simultaneously are used for various compositional alloy and complex oxides. The system
provides the rotating substrate holder which enables the sputtered thin film with high
uniformity and homogeneity. It also provides a sample holder with the high-temperature
capability which can heat the sample to 1000°C for high-quality single crystals.
Fig. 2-1 shows the schematic of magnetron sputtering. Before the sputtering process,
a high vacuum must be obtained. Then, a controlled flow of noble gas, usually Ar, is
introduced into the chamber that will maintain a relatively higher vacuum of several mTorr
in the chamber. From this point, a negative voltage of several hundred volts is applied to
the target. Such a high voltage and low pressure will ionize the argon atoms and turn them
into argon ions and electrons. The negative voltage on the target attracts ions to the target
surface at speed, which will lose their kinetic energy to the target surface. If the energy is
high enough, the surface atoms become sputtered. When the sputtered atoms reach the
substrate surface, they come to condense together and form a dense film. With the aid of
magnets placed underneath the target, a much higher sputtering rate can be achieved due
to the fact that ions are confined to the surface of the target. The sputtering rate is typically
controlled by the DC or RF source power.
A calibration film with the thickness estimated around 50 nm is usually used for
deposition rate calibration. The X-ray reflectometry (XRR) is used for measuring the
calibration film thickness. Typical XRR spectra with various film thicknesses are shown
29
in Fig. 2-2 The intensity of reflection of the X-ray from the film is plotted as a function of
the incident angle 2θ. That angle is kept below 10°. The reflection spectrum shows periodic
fringes, which is resulted from the interference of the reflected X-ray beam from the
interfaces. The thickness of the film is inversely proportional to the gap of the fringes and
can be extracted from the fitting of the reflection spectrum.
Fig. 2-2. X-ray reflectometry (XRR) spectra with different thin films thicknesses
(https://e-reports-ext.llnl.gov/pdf/799501.pdf)
30
2.2 Magnetic Hysteresis Measurements
One of the most common ways to measure the static magnetic properties of a
ferromagnetic material is the measurements of the magnetic hysteresis loop. Ferromagnetic
materials, which exhibit spontaneous magnetization, have a non-zero magnetization when
the applied field external magnetic field is removed. The remained magnetization, at
which the external field is zero, is called the remanent magnetization (Mr). The
magnetic hysteresis loop is measured when an alternating magnetic field is applied to the
ferromagnetic material. In order to switch the magnetization back to zero, an external
magnetic field that is opposite to the field initially applied is needed. The strength of
that opposite magnetic field to drive magnetization to zero is called the magnetic
coercive field (Hc). Since the sign of remanent magnetization depends on the history of
the magnetic field applied, the ferromagnets will exhibit two distinct magnetic states.
This property of ferromagnets is utilized for magnetic memory devices such as hard drive
disk (HDD) and magnetic random access memory (MRAM). A magnetic field (Hk) is
required to fully saturate the materials to its saturation magnetization (Ms). This Hk
represents the uniaxial magnetic anisotropy along the applied field direction. The
hysteresis behavior of ferromagnetic materials is related to the existence of magnetic
domains. Experimental techniques such as vibrating sample magnetometry (VSM),
superconducting quantum interference device (SQUID), and magneto-optical Kerr effect
polarimetry (MOKE) are usually used to measure magnetic hysteresis loop.
In this dissertation, VSM is chosen as the main characterization technique due to its
high sensitivity in measuring in-plane magnetized films. The schematic of a typical
31
VSM is shown in Fig. 2-3 (a). The system usually consists of a pair of electromagnets
which provides alternating external magnetic field (low frequency, time constant > 0.3 s);
a vibrating stage which vibrates the sample at tens of kilohertz; two pair of pick-up coils
which can detect the periodic change in magnetic flux φ produced by the vibrating sample
due to the Faraday's law. In order to eliminate the background noise and increase the
detection limit, the induced voltage in coils is measured using a lock-in amplifier. The
reference frequency is set to be equal to the frequency of the vibrating stage. Low
temperature liquid-nitrogen cryogenic system and high-temperature oven tube can also be
applied.
Due to the small spacing of electromagnets (also the pick-up coils), samples are
usually diced into small square pieces. Prior to any measurements, the VSM needs to be
carefully calibrated using a Ni sphere standard sample with known magnetization at a
specific magnetic field. The magnetization 4πMs of the measured sample can be obtained
in the unit emu/cc by knowing the volume of the magnetic materials. To increase the
signal to noise ratio of the sample with a very small magnetic film thickness (around 1 nm),
several pieces of the sample can be attached on top of each other. In order to characterize
the magnetoelectric coupling of multiferroic heterostructure using VSM, an in-situ voltage
needs to be applied to the sample. Fig. 2-3 (b) shows the magnetic hysteresis loops of
FeGaB/PZT multiferroic heterostructure at different applied electric fields. Changes on
the Hc and Hk is observed indicating a voltage induced magnetization reorientation via
strain-mediated magnetoelectric effect.
32
Fig. 2-3. (a) Schematic of vibrating sample magnetometry (VSM). (b) Typical magnetic hysteresis loop of
FeGaB/PZT multiferroic heterostructure with different applied electric fields.
33
2.3 Ferromagnetic Resonance Spectrometer
Magnetization dynamics has been intensively studied for the precise quantification
of magnetic anisotropy, moment and damping constant of bulk and thin film magnetic
materials. Understanding of magnetization relaxation is essential for the application of
magnetic microwave and magnetic random access memory devices (MRAM). For example,
the switching speed of the magnetic memory element is strongly related to the damping
constant in such magnetic materials. A faster switching speed and higher memory density
may be achieved by controlling those key parameters.
From a macroscopic point of view, the magnetic moment would start to process
around the direction of the effective local field (Heff) with the application of a static
magnetic field )H0). The magnetization would align with Heff by considering the damping.
The dynamic response of magnetization is described by the Landau-Lifshitz-Gilbert
equation:
𝑑��
𝑑𝑡= −γ(�� × 𝜇0𝐻𝑒𝑓𝑓
) +𝛼
𝑀(�� ×
𝑑��
𝑑𝑡) (2.1)
where M is the magnetization, Heff is the effective magnetic field, and γ is the
gyromagnetic ratio which can be further described by γ = µBg/ћ. The first term of Eq.
2.1 corresponds to the precession of motion, and the second term corresponds to the
damping. The effective magnetic field Heff can be described by:
H𝑒𝑓𝑓 = −𝑑𝜉
𝑑𝑀 (2.2)
where 𝜉 is the magnetic free energy density which consists of the Zeeman energy of
34
external DC and RF magnetic fields, the demagnetization energy, the magnetocrystalline
energy. The static magnetic properties appearing in the Eq. 2.1 can be determined by
angular and frequency dependent FMR measurements. The resonance frequency can be
calculated as:
(𝜔
𝜇𝐵𝑔/ћ)2
= −1
𝑀2𝑠𝑖𝑛2(𝜃)[𝜕2𝐹
𝜕𝜃2
𝜕2𝐹
𝜕𝜙2 − (𝜕2𝐹
𝜕𝜃𝜕𝜙)2
] (2.3)
where ω is the microwave frequency, F is the total free energy, g is the g-factor, 𝜙 is the
azimuthal angle, and θ is the polar angle.
Several experimental techniques have been developed to carry out the response of the
magnetization dynamics [1]. Microwave cavity or coplanar waveguide (CPW) is used to
provide an RF magnetic field for inducing the magnetization procession. Such methods
have been used for bulk or thin film sample with millimeter dimensions. For
characterization of micro- or nano-scale magnetic devices, spin-torque ferromagnetic
resonance (ST-FMR) can be employed with a current induced spin torque. We use a
commercial Brucker electron paramagnetic resonance (EPR) spectrometer with an X-
band microwave cavity providing 9.75 GHz RF magnetic field. It utilizes the Zeeman effect
as shown in the schematic of Fig. 2-4 (a). With the application of the magnetic field, the
electron would have a lowest and highest energy state depending on the alignment of the
moment of the electron and the magnetic field. From quantum mechanics, the energy of
each spin state can be described as ±1 gµBB0, and the energy difference between the two
spin states is:
∆𝐸 = ℎ𝜐 = 𝑔𝜇𝐵𝐵0 (2-4)
35
From Eq. 2.4, one can tune the energy difference by changing the magnetic field
strength or microwave frequency. In the case of a microwave cavity, the magnetic field
becomes the only varying parameter with the fixed microwave frequency. There would be
a peak in the absorption spectrum when ∆E matches the energy of the radiation at a
specific magnetic field. This field is called the resonance field. The absorption power as a
function of the magnetic field is shown in the inset of Fig. 2-4 (b). For EPR spectrometer,
it uses lock-in technique to enhance the sensitivity. A magnetic field modulation is
applied using a pair of coils which provides a low-frequency sinusoidal signal. The
amplitude of the absorption signal could be modulated at the same frequency, and the
detected EPR signal is approximately linear over an interval with an amplitude
proportional to the slope of the absorption signal. Although a large modulation field can
increase the SNR by several orders of magnitude, a modulation field strength is usually
ten times smaller than the linewidth of measured materials. Otherwise, the EPR signal
broadens and becomes distorted. Fig. 2-4 (b) shows the EPR signal as a function of
magnetic field. It is represented as the first derivative of the absorption spectrum. In this
case, the resonance field corresponds to the intercept between the spectrum and the
baseline. The resonant linewidth ∆Hpp can also be derived from the field difference
between the resonant peak and dip. This ∆Hpp is closely associate with the damping
which may contain intrinsic such as Gilbert damping, and extrinsic terms such as two-
magnon scattering and sample inhomogeneities [2]-[3]. The resonant linewidth which
is linearly proportional to frequency can be expressed as:
Δ𝐻𝑝𝑝 = Δ𝐻0 +2𝜋𝜔𝛼
√3𝛾 (2-5)
36
where ∆H0 denotes the zero frequency offset of the linewidth due to the magnetic sample
inhomogeneities.
Fig. 2-4. (a) Schematic of Zeeman effect. (b) EPR spectrum with showing Hres and ∆Hpp. Insect shows the
absorption spectrum.
37
In order to precisely extract Gilbert damping constant, FMR has to be measured at
various microwave frequencies. A home-built broadband FMR spectrometer is used with
a broadband CPW as shown in Fig. 2-5 (a) [4]. The RF magnetic field can be generated
by an RF source through CPW. Similar to the EPR system, lock-in detection is used to
enhance SNR with a modulation field provided by a pair of coils. Fig. 2-5 (b) shows the
∆Hpp as a function of microwave frequency for a 50 nm NiFe sample. With a simple linear
fit described in Eq. 2.5, Gilbert damping constant 𝛼 = 0.0129 ± 0.0016 and
inhomogeneities ∆H0 = 0.66mT ± 0.06. Moreover, Fig. 2-5 (c) shows the resonance field
as a function of f, which can be fitted to Kittle equation:
𝑓 2𝜋⁄ = 𝛾𝜇0√(𝐻𝐹𝑀𝑅 + 𝐻𝑒𝑓𝑓)(𝐻𝐹𝑀𝑅 + 𝐻𝑒𝑓𝑓 + 𝑀𝑒𝑓𝑓) (2.6)
where magnetic anisotropy field Heff =1.54 ± 0.2 and effective magnetization 4πMeff = 0.96
± 0.04. It is worth noting that a bias electric field can be easily employed to a multiferroic
sample to study the magnetoelectric effect. Magnetic anisotropy field, as well as
damping coefficient, can be precisely measured at various electric fields / strain states.
38
Fig. 2-5. (a) Schematic of a home-built broadband FMR system. (b) Resonant linewidth ∆Hpp as a function
of frequency for NiFe. (c) Resonance field HFMR as a function of frequency for NiFe.
39
2.4 Reference
1 S. S. Kalarickal, P. Krivosik, M. Wu, C. E. Patton, M. L. Schneider, P. Kabos, T. J.
Silva, and J. P. Nibarger: Ferromagnetic resonance linewidth in metallic thin films:
Comparison of measurement methods. J. of Appl. Phys. 99, 093909 (2006).
2 J. Lindner, K. Lenz, E. Kosubek, K. Baberschke, D. Spoddig, R. Meckenstock, J. Pelzl,
Z. Frait, and D. Mills: Non-Gilbert-type damping of the magnetic relaxation in
ultrathin ferromagnets: Importance of magnon-magnon scattering. Phys. Rev. B 68, 6,
060102 (2003)
3 K. Zakeri, J. Lindner, I. Barsukov, R. Meckenstock, M. Farle, U. Von Hörsten, H.
Wende, W. Keune, J. Rocker, S. S. Kalarickal, K. Lenz, W. Kuch, K. Baberschke, and
Z. Frait: Spin dynamics in ferromagnets: Gilbert damping and two-magnon scattering.
Phys. Rev. B 76, 1, 104416 (2007).
4 S. Beguhn, Z. Zhou, S. Rand, X. Yang, J. Lou, and N. X. Sun: A new highly sensitive
broadband ferromagnetic resonance measurement system with lock-in detection. J.
Appl. Phys. 111, 07A503 (2012).
40
3. Acoustically Actuated Ultra-compact NEMS
Magnetoelectric Antennas
3.1 New Mechanism
3.1.1 Motivation
One of the key challenges on state-of-the-art antennas lies in their size miniaturization.
Conventional antennas rely on an EM wave resonance, and therefore typically have a size
of more than one-tenth of the EM wavelength 0. The limitation on antenna size
miniaturization has made it very challenging to achieve compact antennas and antenna
arrays, particularly at very-high-frequency (VHF, 30–300 MHz) and ultra-high-frequency
(UHF, 0.3–3 GHz) with large 0 [1]-[6]. New antenna concepts need to be investigated
with new mechanisms for the reduction of antenna size. Antennas are basically an array of
conductors that can generate the oscillating electric field and magnetic field through
oscillating electric currents which are required to ensure a high radiation altitude. However,
there is another path to attain radiation power, since electricity and magnetization are
always coupling together, which indicates the magnetic currents will also generate the
electromagnetic wave. On the other hand, strong strain-mediated ME coupling in
magnetic/piezoelectric heterostructures has been recently demonstrated which enables
efficient energy transfer between magnetism and electricity.
The strong ME coupling, if realized dynamically at radio frequencies (RF) in ME
41
heterostructures, could enable voltage induced RF magnetic currents that radiate EM waves,
and acoustically actuated nano-scale ME antennas with an entirely new receiving and
transmitting mechanism, for EM waves. However, despite the moderate interaction
between the surface acoustic wave and magnetization [7]-[9], strong ME effect has only
been demonstrated at kHz frequencies, or in a static or quasi-static process [10]-[11]. Here
one question naturally arises: Is it possible to realize efficient energy coupling between
bulk acoustic waves and EM waves in ME heterostructures at RF frequencies through ME
coupling?
Multiferroics research society has been studied in magnetoelectric (ME) antenna since
2012 [12]-[15] and was first theoretically described by Yao [16]. Here we successfully
demonstrated a novel ME antenna with 1 to 2 orders of magnitude miniaturization over
Table 3-1: Key Features of Conventional and ME Antennas
Conventional Antenna Magnetoelectric Antenna
EM Wave Resonance, Size
Comparable to 0, EM
Acoustic Resonance, Size
Comparable to 0, Acoustic
Electric Current Radiation Magnetic Current Radiation
(Ground Plane Immunity)
Gain: -68dBi for same-size dipole
antenna; -90dBi for same-size loop
antenna
Gain: -18dBi for Rchu=550µm ME
size, one of the highest gains
within all nano-scale antennas
42
state-of-the-art antennas [17]. This NEMS antenna can provide -18dBi antenna gain which
is 50 dB higher over the same-nano-size conventional antenna at the similar frequency.
This dissertation will introduce the new mechanism under the viewpoint of microwave
engineers, focus on the comparison between the ME antenna with the conventional
antennas, and propose a bio-medical application- NanoNeuroRFID. Table 3-1 points out
the key features difference between the conventional antenna with the ME antenna.
3.1.2 Theory
Here we demonstrate the nanoelectromechanical system (NEMS) antennas operating
at VHF and UHF frequencies based on the strong ME coupling between EM and bulk
acoustic waves in the resonant ME heterostructures (ferromagnetic/piezoelectric) which
realize both transmitting and receiving mechanisms. The antenna consists one layer of
piezoelectric material and one layer of magnetostrictive material, and it is based on the
bulk acoustic wave (BAW) resonator to transfer the dynamic strain across different layers.
In Fig. 3-1, two proposed antennas structures are shown with the excitation and
vibration direction. Both nano-plate resonators (NPR) and thin-film bulk acoustic wave
resonators (FBAR) have the same excitation but with different resonance mode providing
a variety of frequency coverages possibility. Fig. 3-2 (a) shows the illustration of the new
antenna mechanism. From the transmitting aspect, by applying RF electric field to the
NEMS resonator, the mechanical resonance would induce alternating strain wave/acoustic
wave that can be directly transferred to the upper ferromagnetic thin film. The acoustic
wave would then induce a dynamic change of the magnetization due to the strong
piezomagnetic constant and generate magnetic currents for radiation; Reciprocally, from
43
the receiving aspect, the RF magnetic field component of the electromagnetic wave can be
detected by the ferromagnetic layer and induce an acoustic wave on that layer. When this
acoustic wave transfer to the piezoelectric thin-film, the dynamic voltage/charge or RF
signal would be generated due to the direct piezoelectric coupling.
The acoustic wavelength is about 5 orders shorter than the electromagnetic
wavelength at the same frequency. Therefore, since the ME antennas are operating at the
acoustic resonant frequency instead of EM wave resonant frequency, these ME antennas
are expected to have sizes comparable to the acoustic wavelength, and the antennas can
dramatically shrink into ten to hundreds of times smaller. Furthermore, while this new
mechanism involves the coupling between the electromagnetic wave and acoustic wave,
Fig. 3-1. Illustrations of the nano-plate resonator (NPR) and thin film bulk acoustic wave resonator (FBAR).
44
the additional mechanical resonance can successfully solve the impedance mismatching
from conventional small antennas.
Fig. 3-2 (b) shows the illustration of the ground plane effect. A ground plane is a flat
or nearly flat horizontal conducting surface, which can also be the human body. Ground
plane plays major roles in determining its radiation characteristics including gain. However,
the imaging currents flowing in the opposite direction of planer antennas can cancel out
the radiation from the antennas. This is why most of the large profile antennas are in
vertical position and perpendicular to the surface such as vertical quarter wave dipole,
where the imaging currents serve as part of an antenna for in-phase radiation. The ME
Fig. 3-2. (a) Illustration and explanation of the new antenna mechanism. (b) Illustration and explanation of
the ground plane effect.
45
antennas use magnetic currents for radiation instead of electric currents. The in-phase
imaging current will provide 3dB gain enhancement while attaching on the ground. This
ground plane immunity property can provide a variety of applications on the metallic
surface and on the human body which is also considered as a ground plane.
3.1.3 Modeling
The coupling between the magnetic, elastic and electric field in the two different
magnetostrictive and piezoelectric materials should be taken into consideration for
analyzing the response of the ME structure. Simulations using finite element method (FEM)
software, Comsol Multiphysics, were carried out to analyze the frequency response of
magnetic fields, solid mechanics, and electrostatics modules. In Fig. 3-3 (a), the ME
composites were constructed into magnetostrictive, piezoelectric materials and air sub-
domains and simulated in the frequency domain for 3-D geometry to illustrate the modeling
principles for this complex problem. The simulation setup for induced voltage is illustrated
in Fig. 3-3 (b).
In the air phase, we assumed that a spatially uniform, sinusoidal wave magnetic field
is applied. The air model space is truncated by an infinite element domain region. When
using the infinite element domain features, the boundary conditions on the outside of the
modeling does not affect the solutions.
In the magnetostrictive material, the magnetic permeability and the magnetostrictive
strain show a nonlinear dependency on the magnetic flux and the mechanical stress/strains
in the ME composite. The constitutive equation of the magnetostrictive is shown as:
46
Fig. 3-3. (a) Comsol direct magnetoelectric coupling simulation process flow. (b) Schematic of the
magnetoelectric nanoplate resonator (NPR) and the induced ME voltage simulation. The RF field (HRF) is
generated by an RF coil.
47
𝐵 = 𝜇0[𝐻 + 𝑀(𝐻, 𝑆𝑚𝑒) + 𝑀𝑟] (3.1)
where 𝐵 and 𝑀𝑟 are the magnetic flux density and the remanent magnetization,
respectively; The dynamic magnetization 𝑀 is related to 𝐻 and 𝑆𝑚𝑒 which represent the
magnetic field and the mechanical stress, respectively. Assuming as an isotropic material,
the magnetostrictive strain 𝜀𝑚𝑒is modeled as the following quadratic isotropic form of the
magnetization field 𝑀:
𝜀𝑚𝑒 =3
2
𝜆𝑠
𝑀𝑠2 𝑑𝑒𝑣(𝑀⨂𝑀) (3.2)
where the magnetostrictive coefficient 𝜆𝑠 and the saturation magnetization 𝑀𝑠are set to be
70 ppm and 1114084 A/m from the experimental results of the FeGaB, respectively.
In the piezoelectric material, we assume a small signal behavior described by the
linear piezoelectric material model, in which we established constitutive relations in a
strain-charge form. Similarly, piezoelectric tensors and mechanical properties were
obtained from the built-in modules. The relation between the stress, electric field, and the
electric displacement field in a stress-charge form is given by the piezoelectric constitutive
equations:
σ = cε − eE (3.3)
D = cε + κE (3.4)
where σ and ε are the stress and strain tensors; E, and D are the electric field and electric
flux density; c, e, and κ are the stiffness, strain to electric field coupling constant and
48
permittivity, respectively. The solid mechanics model is described by the elastic
constitutive relations:
ε =1
2[(∇𝑢)𝑇 + ∇𝑢] (3.5)
σ = Cε (3.6)
∇σ = −ρ𝜔2 (3.7)
where 𝑢 is the displacement, 𝜌 is the density, 𝜔 is the angular frequency, and is 𝐶 the
elasticity matrix.
The purpose of the simulation is to demonstrate the capability of ME coupling and to
observe the resonance mode of the device using magnetostatic approximation only in the
near-field regime where the magnetostriction and piezoelectric modules in COMSOL are
reliable and widely used. However, simulations so far may not be able to capture the real
physics which contain many boundary conditions and anisotropic materials parameters.
For example, the magnetic FeGaB layer in the ME antenna shows a highly anisotropic
Young’s modulus with a ΔE effect of 160 GPa along the in-plane magnetic hard axis
direction, which is very hard to incorporate into any existing model such as Comsol. It is
very challenging to do 3-D real device structure for far-field simulation in the framework
of a three-dimensional (3D) ME antennas. It is beyond the scope but will be the focus in
the future. Recently, 3-D models with complete dynamic Maxwell equations and ME
coupling using MATLAB has been investigated and had great progress by Yao [18]-[19].
49
3.1.4 Micro-fabrication
Both nano-plate resonator (NPR) and thin-film bulk acoustic wave resonator (FBAR)
devices provide the integrated capability to CMOS technology and use the same five-masks
micro-fabrication process which is shown in Fig. 3-4. The process starts with a high
resistivity Silicon (Si) wafer (>10000 Ohm.cm). The Platinum (Pt) film was sputter-
deposited and patterned by lift-off on top of the Si substrate to establish the bottom
Fig. 3-4. The structure and layers using five-masks micro-fabrication process flow of the ME antenna.
50
electrodes. Then, the Aluminum Nitride (AlN) film was sputter-deposited, and vias were
etched by H3PO4 to access the bottom electrodes. After that, the AlN film was etched by
inductively coupled plasma (ICP) etching in Cl2 based chemistry to define the shape of the
resonant nano-plate. Next, the gold (Au) film was evaporated and patterned by lift-off to
form the top ground. Finally, the FeGaB/Al2O3 multilayer layer [20]-[21] was deposited by
a magnetron sputtering and patterned by lift-off. A 7960 A/m (100 Oe) in- situ magnetic
field bias was applied during the magnetic deposition perpendicular to the anchor direction
of the device to pre-orient the magnetic domains. Then, the structure was removed by XeF2
isotropic etching of the Silicon substrate to minimize the substrate clamping effect. For the
biomedical application, the ME antenna will be completely contact-less and encapsulated
with the biocompatible material
The FeGaB/Al2O3 multilayers have the stacking of [Fe7Ga2B1 (45 nm)/Al2O3 (5 nm)]
×10), which demonstrate eddy-current loss reduction, lower out of plane anisotropy, and
higher permeability in comparison with a single FeGaB layer of the same thickness [22].
The magnetic multilayers have a total thickness that is equal to the thickness of AlN thin-
film for achieving high sensitivity and the high-quality factor of the resonator at the same
time. The magnetic multilayers were sputter-deposited with a 5 nm Ta seed layer at the Ar
atmosphere of 3 mTorr with a background pressure of less than 1 × 10−7 Torr. The Ta seed
layer promoted the FeGaB thin-film growth exhibiting narrow resonance linewidth and
close-to-bulk magnetic moment [23]. XeF2 Si release process would etch the magnetic
material resulting in a very rough surface of the magnetic thin film. So Al with low density
is chosen as a capping layer of magnetic materials for protection. The FeGaB layer was co-
sputtered from FeGa (DC sputtering) and B (RF sputtering) targets. The Al2O3 layer was
51
Fig. 3-5. (a) Magnetic hysteresis loop and (b) Ferromagnetic resonance spectrum of FeGaB/Al2O3
multilayers.
52
deposited by RF sputtering using an Al2O3 target. The deposition rates are calibrated
with X-ray reflectivity.
In Fig. 3-5 (a), the FeGaB/Al2O3 multilayers with a magnetic coercive field <400 A/m
(0.5 mTesla) measured by vibration sample magnetometer (VSM) indicates a soft magnetic
Fig. 3-6. (a) The optical images of the fabricated NPR and FBAR. (b) The scanning electron microscopy
(SEM) image of the fabricated NPR and FBAR.
53
property. It is essential for achieving a large magnetostriction constant of 70 ppm which is
so far the record high value. Fig. 3-5 (b) shows the FMR spectrum taken at 9.5 GHz of
FeGaB/Al2O3 multilayers which gives a resonance frequency of 93 mT and magnetic
moment of 1.15 T based on the Kittel equation. The resonance linewidth of 4780 A/m (6
mTesla) measured by ferromagnetic resonance spectroscopy can also be obtained
demonstrating a good microwave property with a low magnetic loss. Instead of the
patterned devices, the reference sample is a full film with a lateral dimension of 5 mm by
5 mm. Note that there could be a variation in magnetic properties between the reference
sample and device due to the different shape anisotropy and stress state. Fig. 3-6 (a) shows
the optical images of the fabricated NPR and FBAR; Fig. 3-6 (b) shows the scanning
electron microscopy (SEM) image of the fabricated NPR and FBAR. The resonators are
released from the silicon substrate but mechanically supported and electrically contacted
by the two AlN/Pt anchors for optimized ME coupling with a minimum substrate clamping
effect to maximum the resonance performance.
54
3.2 Experimental Data
3.2.1 Magnetoelectric Coupling Demonstration
The performance of ME coupling was demonstrated through the NPR with an in-plane
contour mode (by means of d31 piezoelectric coefficient) of vibration excited with a
perpendicular electric field on the piezoelectric AlN layer. The length (L) and width (w) of
the FeGaB/AlN resonator are 200 µm and 50 µm, respectively. The use of a NEMS
resonator with an ultra-thin (thickness, t = 500 nm) AlN thin film enables efficient on-chip
acoustic transduction with ultra-low energy dissipation [24]-[25]. The electrical admittance
curve was characterized by using a network analyzer as shown in Fig. 3-7 (a) to study the
electromechanical properties of the ME NPR. The short-open-load calibration was
performed prior to the device measurements. The available power at the network analyzer
port was set to -12 dBm, and the IF bandwidth was 50 Hz. The devices were tested in an
RF probe station with a probe with ground-signal-ground configuration. The resonant
frequency corresponds to the contour mode of vibration excited in AlN, which can be
analytically expressed as
𝑓𝑟 ∝1
2𝑤√
𝐸
𝜌 (3.8)
where 𝑤 is the width of the resonator, E and ρ are the equivalent Young’s modulus and
equivalent density of the resonator, respectively [26]-[27]. Comsol simulation on the
admittance curve of the same NPR shown in Fig. 3-7 (b) shows good agreement with the
experimental result. The in-plane displacement distribution shown in Fig. 3-7 (b) inset
55
Fig. 3-7. (a) Measured admittance curve of the ME NPR. (b) Simulated admittance curve of the ME NPR.
The inset indicates a contour extensional mode of vibration at resonance with the applied RF voltage signal.
56
Fig. 3-8. (a) Calculated ME coupling coefficient (left axis) and the Measured induced ME voltage (right
axis) versus the frequency of HRF excitation. (b) Simulated induced ME voltage.
57
indicates a contour extensional mode of vibration at resonance with the applied RF voltage
signal. It is also notable that the Q-factor 930 of this ME resonator is much higher than the
conventional low-frequency ME heterostructures in previous reports [28]-[32].
Under HRF excitation with an amplitude about 60 nT (Wb/m2) from an RF coil along
the length direction of the resonator, the induced ME voltage output of the NPR device was
measured by an ultrahigh frequency lock-in amplifier (UHFLI), as shown in Fig. 3-8 (a).
A clear electromechanical resonance peak is shown in the ME voltage curve at 60.7 MHz
with a peak amplitude of 180 μV which match the resonance well in Fig. 3-7 (a). The
experimentally measured voltage agrees well with the simulated results of the ME voltage
with a peak amplitude of 196 μV as shown in Fig. 3-8 (b) The inset shows the simulated
in-plane displacement of the ME resonator excited by the HRF at resonance. The same mode
of vibration excited by the magnetic field and electric field demonstrates that the strain-
mediated ME coupling is dominating. A maximum ME coupling coefficient 𝛼𝑀𝐸 of 6
kVOe-1cm-1 can be obtained from [33]:
𝛼𝑀𝐸 = 𝜕𝑉𝜕𝐻𝑅𝐹𝑡⁄ (3.9)
where 𝑉 is the induced voltage and 𝑡 is the thickness of AlN. It is notable this ME coupling
coefficient is obtained without any DC bias magnetic field, and the value is comparable to
the recent reported values with the optimum bias magnetic field at much lower
electromechanical resonance frequencies of kHz [34]
58
Fig. 3-9. (a) Measured admittance curve of the non-magnetic NPR. (b) Measured induced ME voltage
versus the frequency of HRF excitation.
59
As a comparison, a non-magnetic NPR has also been tested as a control device to
confirm that the strain-mediated ME coupling is responsible for the observed voltage
output under the HRF excitation. A 500 nm Cu thin film was deposited on the AlN plate to
replace the ferromagnetic FeGaB layer as the top electrode. In Fig 3-9 (a), the Cu/AlN
based NPR exhibits a similar admittance behavior as the ME NPR. With the same HRF
excitation, the induced voltage of the Cu/AlN resonator at resonance shown in Fig 3-9 (b)
is extremely low, about two orders of magnitude smaller than the induced voltage in the
ME NPR. We can observe that the induced voltage spectrum profile of the Cu/AlN nano-
plate resonator is highly antisymmetric near its resonance frequency, which is different
from the symmetric ME voltage curve but similar to its admittance curve. This
antisymmetric line shape can be attributed to a weak inductive coupling effect between the
device ground loop and EM wave, which could also exist in the ME NPR device. However,
the symmetric ME voltage spectrum indicates that the inductive coupling effect has an
extremely low efficiency comparing to the ME coupling. Thus, the strong resonance peak
induced by the HRF in FeGaB/AlN NPR device results from the presence of high-
permeability FeGaB films [35] which couples to RF excitation magnetic field very
effectively, that is the ME coupling.
60
3.2.2 Modified Equivalent Circuit Modeling
In Fig. 3-10, the admittance amplitude of NPR can be fitted with Modified
Butterworth–van Dyke (MBVD) model [36] which is traditionally used to simplify and
characterize the piezoelectric resonator to extract the electromechanical parameters such
as resonance frequency, electromechanical coupling coefficient kt2, and Q-factor. MBVD
equivalent circuit consists of electrical components and equivalent mechanical components
connected in parallel. The electrical components include the device capacitance C0 which
is defined by the device geometry and a resistance R0p which is associated with the
dielectric loss. While the mechanical branch contains the motional capacitance Cm,
motional inductance Lm and motional resistance Rm, which can be expressed as:
𝑅𝑚 = 1 𝜔0𝑐0𝑘𝑡2⁄ 𝑄 (3-10)
𝐶𝑚 =8
𝜋2 𝐶0𝑘𝑡2 (3-11)
𝐿𝑚 = 1 𝜔02⁄ 𝐶𝑚 (3-12)
The series resistance Rs is serially connected to both branches as the electrical loss of the
electrodes. The resonance frequency occurs at the resonance frequency 2πω0, where the Cm
and Lm cancel with each other. The kt2 represents the efficiency of electrical and acoustic
energy conversion, and Q-factor defines the ratio of the energy stored in the vibrating
resonant structure to the energy dissipated per cycle by the damping processes. Note that
the kt2Q is the figure of merit (FOM) of an electromechanical resonator.
61
A high-quality factor of 930 in the air was extracted from the MBVD fitting at zero
bias magnetic field, while the quality factors reported in different magnetoelectric
resonators operating at low frequencies were around 100. The electrically floating
(FeGaB/Al2O3)×10 multilayers provide good confinement of the electric field within the
entire thickness of the AlN layer, which results in a high electromechanical coupling
coefficient 𝑘𝑡2 of 1.35%, comparable to what is typically achieved in conventional AlN
nano-plate resonator employing the similiar electrode configuration [37].
Fig. 3-10 The Modified Butterworth–van Dyke (MBVD) model.
62
3.2.3 Magnetic Sensitivity
The ME NPR with multi-finger interdigitated electrodes, which we have
demonstrated recently [38], were found to have negligibly small ME voltage in the same
measurement setup, that is over three orders of magnitude smaller at the electromechanical
resonance compared to the single-plate ME NPR. Single-finger ME resonators producing
high ME output voltage indicates the uniform RF excitation magnetic fields couple
strongly to the single nanoplate. While the negligibly ME voltage output in multi-finger
ME resonators is due to the fact that, the uniform HRF do not couple efficiently to the multi-
finger nano-plate resonators which produce non-uniform RF strain fields and non-uniform
magnetization fields. We further gain insight into the magnetization dependence of the
single-finger ME NPR shown in Fig. 3-11 by examining its ME coupling strength at
different bias magnetic fields. The induced ME voltage spectrum was measured with DC
bias magnetic fields swept from -5 mT to 5 mT along the resonator length direction (as
shown in the inset of Fig. 3-11 (b).
Fig. 3-11 (a) shows the ME coupling coefficient α as a function of the bias DC
magnetic field strength and the RF driving magnetic field frequency. At zero bias magnetic
field μ0HDC=0, the α is maximized at the fr of 60.7 MHz, which is in good agreement with
Fig. 3-7 (a). At μ0HDC = ±5 mT, fr is shifted to 60.72 MHz as shown in the dashed curve of
Fig. 3-10 (a). This can be attributed to the ΔE effect [39], that is the bias magnetic field
modifies the Young’s modulus of FeGaB and thus leads to different fr of the resonator.
Moreover, a hysteresis behavior of the ME coupling coefficient (at fr) was observed by
sweeping the DC magnetic field back and force, with a maximum of 6 kV cm-1 Oe-1 at
63
Fig. 3-11 (a) ME coupling coefficient αME of ME sensor as a function of bias DC magnetic field (x-axis)
and the RF driving frequency (y-axis). The dashed curve exhibits the resonance frequency (highest intensity
at each frequency sweep) versus bias magnetic field. The bias magnetic field was swept from -5 mT to 5
mT. (b) The hysteresis loop of αME obtained by sweeping the magnetic field back and force. The inset shows
the schematic of the ME NPR with the external bias magnetic field applied along its length direction.
64
±0.5 mT in Fig. 3-11 (b), which is consistent with the strain-mediated ME coupling
mechanism and the magnetic hysteresis of the FeGaB/AlN nanoplate. This provides
another direct evidence that the observed interaction between EM wave and acoustic
resonance in ME NPR results from the ME coupling between magnetostrictive FeGaB and
piezoelectric AlN in the resonant body.
From the experimental results, it is interesting to note that the strong ME coupling
coefficient at zero magnetic field and the relatively weak dependency of ME coupling
coefficient on bias magnetic field directly lead to robust self-biased ME sensor which is
critical for real applications. These are drastically different from conventional ME
heterostructures with electromechanical resonance frequencies in the kilohertz frequency
range which show near zero ME coupling at zero bias magnetic field [40]-[42], which is
worthwhile to investigate in the future. One of the reason might be attributed to the edge
curling wall under the self-bias condition for the magnetic/non-magnetic multilayers
(FeGaB/Al2O3) used as the magnetostrictive layer in ME antennas at megahertz frequency
range [43]-[44].
The detection limit of the NPR ME antennas for sensing weak HRF under zero bias
magnetic field was also characterized as shown in Fig. 3-12, where the induced voltage is
plotted as a function of HRF at two different excitation frequencies. At the resonance
frequency of 60.7 MHz (red), the linear curve scatters at 40 pT with a limit detection
voltage of 0.1 µV, indicating a detection limit of 40 pT for the NPR ME sensor. While at
the off-resonance frequency of 1 MHz (blue), the induced voltage randomly distributes
around the 0.1 µV, showing no sensitivity to 1 MHz magnetic field excitation with the
amplitude of 10-11 T - 10-7 T.
65
Fig. 3-12 Induced ME voltage as a function of magnetic field at excitation frequency of 60.7 MHz (red)
and 1 MHz (blue) indicates the detection limit.
66
3.2.4 Far Field Measurement
We further fabricated ME antennas that operate at the GHz range based on the out-of-
plane mode of thin-film bulk acoustic wave resonators (FBAR), where the resonant
frequency is defined by resonator thickness rather than the width. The antenna transmission
behavior of the FBAR ME antennas was tested in a far-field configuration at GHz range in
an anechoic chamber. For small antennas that the dimension is shorter than half of the
wavelength, the far-field region can be considered at > 2 × wavelength. A calibrated linear
polarization standard horn antenna and an ME FBAR based antenna with a diameter of 550
µm (Magnetic disk diameter of 200 µm) are connected to port 1 and port 2 of a network
analyzer, respectively for antenna measurements. The resonant frequency corresponds to
the thickness mode of vibration excited in AlN, which can be expressed as
𝑓𝑟 ∝1
2𝑡√
𝐸
𝜌 (3.13)
where 𝑡 is the thickness of the resonator, E and ρ are the equivalent Young’s modulus and
equivalent density of the resonator, respectively. The resonance frequency was found to be
2.53 GHz by measuring the reflection coefficient (S22) of the FBAR device as shown in
Fig. 3-13 (a) with a peak return loss of 10.26 dB. The inset shows the simulated out-of-
plane displacement of the FBAR indicating a thickness extensional mode of vibration.
A non-magnetic control device with 1000 nm Al/ 500 nm AlN has also been tested
with the same experimental setups in order to rule out any artificial EM coupling to the
ground loop of devices. In the non-magnetic control device, 1000 nm Al was used to
replace the 500 nm thick FeGaB multilayer for achieving a device resonance frequency
67
Fig. 3-13 (a) Return loss (S22) of ME FBAR. The inset shows the simulated out-of-plane displacement of
the disk at the resonance peak position. (b) Return loss (S22) curve of the non-magnetic Al/AlN control
FBAR
68
Fig. 3-14 (a) Transmitting and receiving behavior (S12 and S21) of ME FBAR. (b) S12 and S21 of the non-
magnetic Al/AlN control FBAR
69
near 2.5 GHz. As shown in Fig. 3-13 (b), the Al/AlN control device exhibits a similar S22
but better impedance matching with at 2.50 GHz. This indicates that both ME and control
antennas have the similar input energy to the antenna at a similar frequency.
The receiving and transmitting behavior of ME antennas corresponds to the S21 and
S12 parameters, respectively, as shown in Fig. 3-14 (a). A large and clear peak can be
observed from the ME antenna transmission behavior. However, as shown in Fig. 3-14 (b),
no evident S21 and S12 resonance peak can be observed in the measurements except a very
weak peak at 2.50 GHz with a peak amplitude ~20dB lower than the performance of ME
antenna. Similar to the Cu/AlN NPR control device, we can suggest that the ME coupling
effect strongly enhances the performance of the antenna transmission.
The radiation behavior of ME FBAR antenna was also tested by rotating the linearly
polarized standard antenna as shown in Fig. 3-15. The standard antenna can be rotated
along one of the three major axes of the ME antenna. The out-of-plane axis (with in-plane
rotation) in Fig. 3-15 (a) and (b), in-plane axis perpendicular to the ME antenna anchor
direction (with out-of-plane rotation) in (c) and (d), and in-plane axis along the ME antenna
anchor direction (with out-of-plane rotation) in (e) and (f). In all the schematics of Fig. 3-
15, the sinusoidal wave along 0 (or 180) direction denotes the propagating H-field
component of the incoming EM wave. All three polar gain charts in Fig. 3-15 (a), (c) and
(e) show the similar shape of the sideways figure eight due to the magnetic anisotropy of
the FeGaB/Al2O3 multilayer in the circular resonating disk of the ME FBAR. As shown in
Fig. 3-15 (a), the ME FBAR antenna has the highest gain when the Hrf is perpendicular to
the anchor direction of the antenna, and lowest gain when the Hrf is parallel to the anchor
direction. This is probably because the in-plane magnetic anisotropy of the FeGaB in the
70
Fig. 3-15 Antenna polar normalized gain charts: (a) and (b) The out-of-plane axis with in-plane rotation;
(c) and (d) The in-plane axis perpendicular to the ME antenna anchor direction with out-of-plane rotation;
(e) and (f) The in-plane axis along the ME antenna anchor direction with out-of-plane rotation. The
sinusoidal wave along 0 or 180 direction denotes the propagating H-field component of the EM waves.
71
circular disk of the FBAR is along the width direction of the ME antenna, and the highest
permeability and therefore strongest coupling between Hrf and ME antenna is achieved
along 0 or 180 direction in Fig. 3-15 (a). The other two rotation test configurations in Fig.
3-15 (c) and (e) show similar behavior, in which the antenna gain shows its maximum
value at 0° (or 180°). This is related to the in-plane anisotropy of the thin ferromagnetic
layer. All the rotational antenna gain measurements at different configurations demonstrate
that the high ME antenna gain originates from the strong magnetic coupling between the
magnetic field component of the EM wave and the FeGaB of the FeGaB/AlN
heterostructure in ME FBAR antennas.
72
3.3 Discussion
3.3.1 Frequency Capability
We measured the resonance frequency fr of various devices with different design
principles and geometries via a network analyzer. Fig. 3-16 plots the fr as the function of
1/w for NPR and 1/t for FBAR. As shown, the fr of NPR devices is inversely proportional
to the width for NPRs, and its fr is inversely proportional to the AlN thickness for FBAR.
Fig. 3-16 Measured Resonance frequency as the function of one over width (1/w) for NPR resonators and
one over thickness (1/t) for FBAR resonators.
73
All devices are fabricated on one single chip with the same fabrication, deposition
processes, and the same layered structure. This indicates that by simulation and device
geometry design, we can achieve a wide frequency band from tens of MHz (NPR with
large width) to tens of GHz (FBAR with thinner thickness) on one chip. A bank of multi-
frequency NEMS resonators can be connected to a CMOS oscillator circuit for the
realization of reconfigurable ME antenna arrays [45].
3.3.2 Antenna Gain
The antenna gain 𝐺𝐴 is calculated as -18dBi from the gain-transfer (gain-comparison)
method [46] which can be expressed as
𝐺𝐴 = 𝐺𝑟 + log10(𝑃𝐴 𝑃𝑟⁄ ) = 𝐺𝑟 + 𝑆21,𝐴 + 𝑆21,𝑟 (3-14)
where 𝐺𝑟 is the gain of the reference horn antenna, and 𝑃𝐴 and 𝑃𝑟 are the radiation power
of ME antenna and reference horn antenna. Although small antennas are inherently
inefficient, trade-offs among size, cost, and frequency defined from the application often
require that an antenna be physically small. However, the gain of the ME antenna is already
one of the highest gain among the nano-scale antennas at the smiliar frequency regium.
As we mentioned that ME antenna uses magnetic currents for radiation, we make the
gain compared with the same size small loop antenna to compare the antenna gain since
the small loop antenna, which is most often used as receive antennas for magnetic field
sensing or magnetic radiators, acts like a magnetic dipole. Small loop antennas have overall
circumference less than about one-tenth of a wavelength (C< 𝝀0/10). Low resistance and
high reactance make their impedance matching extremely difficult. ANSYS HFSS is used
74
to simulate the antenna gain of the small loop antenna. The small loop antenna has the same
dimension of the ME antenna with a=550 μm, where a is the smallest imaginary sphere of
radius enclosed the entire antenna structure including the ground loop. The small loop
antenna was designed as chip-scale devices and compatible with a lithographic fabrication
process. The substrate is a 2.2 μm thick AlN and the conductor is a 5 μm thick copper to
reduce the eddy current loss. In Fig. 3-17, the small loop antenna has a resonance at 34
GHz with a return loss of 22 dB; at 2.52 GHz (the resonance frequency of ME FBAR
antenna), the return loss is about 0.065 dB which will dramatically bring down the antenna
gain to −68.4 dBi which is 50dB lower than the ME antenna gain. This indicates that the
Fig. 3-17 Simulated reflection coefficient (S11) of the small loop antenna. The inset shows the schematic of
the simulated small loop antenna.
75
frequency of the same size conventional small loop antenna must be as high as tens of GHz
to achieve impedance matching.
Clearly, these miniaturized ME antennas have drastically enhanced antenna gain at
small size owing to the acoustically actuated ME effect based receiving/transmitting
mechanisms at RF frequencies. We note that the demonstrated ME antennas are purely
passive devices, no impedance matching circuit, or an external power source was used
during the measurement.
Here we consider another case from the calculation, the impedance of a small dipole
antenna having the same size as the ME antenna without considering the fabrication
challenge has the estimated resistance 𝑅 and the reactance 𝑋 are shown as
𝑅 =2𝜋2
3𝑐(
𝑙
𝜆)
2
(3-15)
𝑋 =−120𝜆
𝜋𝑙[ln (
𝑙
2𝑟) − 1] (3-16)
where 𝑙 and r are the length and radius of the dipole, respectively. We can get the estimated
dipole impedance 𝑍 = 0.02 + 𝑗14000 , which the reactance is extremely large.
Conventional small antennas would be very difficult to have design with proper impedance
matching to cancel out the high reactance, and very little power would be delivered from a
50 ohms source to a 0.02 ohms load.
However, the loss mechanism of ME antennas is determined by the radiation
resistance Rr and the mechanical resistance Rm related to the different mechanical damping
mechanisms of the magnetic and piezoelectric phases. Therefore, the impedance matching
76
is no longer only directly related to the radiation resistance of ME antennas, which is
different from conventional antennas. The ME antenna with the second acoustic
mechanism can successfully match the impedance and compensate the high reactance. The
measured impedance is 𝑍 = 68 + 𝑗25.
Ferrite chokes are applied on RF cables for ME antenna measurements. As potentially,
current leakages from coaxial cables/electrodes which connects the DUT and VNA will
contribute EM emissions to antennas which might affect the gain from 5 to 10 dB in small
antenna measurements. Investigation for precise ME antenna measurement needs to be
done in the future to rule out more contribution from the cable by different methods such
as optical-link or balun RF feeding.
3.3.3 Antenna Efficiency
The input power into the ME antenna is taken as the power irradiating the resonator
due to the horn antenna. This input power is the product of the power density (simulated
from Comsol) and the effective area (Aeff) of the ME antenna. The effective area (Aeff) can
be estimated to be 1.8 × 10−5 𝑚2 following the equation
𝐴𝑒𝑓𝑓 =𝜆2𝐺
4𝜋 (3-17)
where 𝐺 is the calculated gain from the previous section. The output power from the device
can be calculated using the S21 parameter measured from the network analyzer. Therefore,
the efficiency of the ME antenna is the ratio of the device output power to the power
captured by the device from the transmitting horn antenna which is 0.438% (-23.58dB).
77
We can also estimate the radiation power of the ME antenna with a simple magnetic
dipole model for a conceptual understanding. Assuming the input power can completely
switch all magnetic dipole moment for radiation. The magnetic dipole moment (m0) can be
expressed as
𝑚0 = 𝑀𝑠𝜋𝑟2𝑡 (3-18)
where 𝑀𝑠 is the saturation magnetization, and 𝑟 and 𝑡 are the radius and thickness of the
magnetic film. We obtain a radiation power 𝑃 from the magnetic film to be 2.8 × 10−18𝑊
and 0.28% (-25.53dB) radiation efficiency following the equation shown as
𝑃 =𝜇0𝜔4𝑀𝑚0
2
12𝜋𝑐3 (3-19)
Due to the in-plane uniaxial anisotropy with high sensitivity along the width direction
of the circular resonating magnetic disk. The ideal directivity 𝐷 of 6dB can be roughly
assumed by integrating the magnetic power density
𝐷 =∫ ∫ ∫ 𝜌
𝜋0 sin 𝜃 sin 𝜑𝑑𝜃𝑑𝜑𝑑𝜌
𝜋0
𝑝0
∫ 𝜌𝑑𝜌𝑝
0
(3-20)
where 𝑃(𝜌, 𝜑, 𝜃) is the magnetic power density in spherical coordinates. The directivity 𝐷
of 5.63dB and 7.58dB from two different method, which are both comparable to the ideal
value, can be calculated from the equation
𝐷 = 𝐺𝐴 𝜉𝑟𝑎𝑑⁄ (3-21)
The commonly used definition of an electrically small antenna is an antenna that
meets the requirement ka < 1, where k is the wave number 2π/λ, and a is the radius of the
78
“Chu sphere.” In 1948, Chu [47] derived the minimum possible Q for an omnidirectional
antenna enclosed in a Chu sphere. Though Chu’s contributions would provide the reference
for engineers that refined these limits, he restricted his analysis to a specific type of
“omnidirectional” antenna, which is quite different from this new mechanism - ME antenna,
where the directivity is estimated to be 6dB. From different estimations show that our
experimental results are of the correct order-of-magnitude.
3.3.4 ME antenna Arrays
ME antenna arrays can be designed to improve the performance especially for
multiband and wideband applications since the electromechanical resonance frequency of
is inversely proportional to the width or thickness of the resonator. ISM communication
band frequency ~27MHz is selected here for discussion. From simulation results, we can
design to use ME antenna arrays in series for achieving higher output and in parallel with
different width for multiband and wideband applications. From Fig. 3-18 (a), the simulated
induced voltage from an RF magnetic excitation, clearly, three magnetoelectric antenna
resonators arrays can lead to more than tripled output voltage comparing to single
resonators due to non-linear coupling; Since the resonance frequency is mainly depending
on the width of the resonator, by slight change of the widths, arrays in parallel could widen
the bandwidth for the desired frequency bandwidth as shown in Fig. 3-18 (b). Using series
and parallel combinations can provide a variety of specifications for different applications.
79
Fig. 3-18 (a) Comparison of simulated induced voltages from a radio frequency magnetic excitation with
among one to three resonators arrays in series. (b) Comparison of return loss of ME antenna with three
resonators arrays in parallel for achieving broadband performance. Insets show the displacement at
resonance which indicates the resonate mode of the ME antennas.
80
3.3.5 Minimalization Techniques
Antenna design is a compromise among volume, bandwidth, gain, and efficiency. The
best compromise choice is usually attained when most of the available volume is excited
for radiation [48]-[53]. There are several techniques for antenna miniaturization. Loading
the antenna with high permittivity and permeability material shortens the effective
wavelength and leads to lower resonance frequency to assist in antenna miniaturization
[54]-[59]; modifying and optimizing the antenna geometry and shape is the most widely
used method including slot loading, bending, folding, meandering, and fractal loading,
etc.[60]-[67]; utilizing lumped components compensates the large reactive impedance of
the electrically small antennas [68]-[74]; Using artificially engineered electromagnetic
metamaterials compensates the large reactive impedance of the electrically small antennas
[75]-[80]; Inducing a variable dipole by mechanical motion provides another matching
approach [81]. The maximum dimension of the miniaturized antennas with different
miniaturization techniques are presented against the operation frequency as shown in Fig.
3-19. Since the acoustic wavelength is much less that of the EM wave resonance, the ME
antennas are much smaller than state-of-art compact antennas. Size miniaturization of the
novel ME antennas is not due to high permittivity or high permeability of the materials,
which is different from conventional magnetodielectric antenna approaches.
81
Fig. 3-19 The maximum dimension of miniaturized antennas with different techniques vs. frequency.
82
3.4 NanoNeuroRFID
3.4.1 Research Strategy
The demonstration of the ME antennas has excited the bio-medical society due to the
high magnetic field sensitivity and the highest antenna gain within all nano-scale antennas
with ground plane immunity from the human body. This provides many possible
applications such as miniature brain implants, brain-computer interfaces, and medical
devices. Here we bring up a biomedical application example for ME antenna that we are
currently working on, the NanoNeuroRFID. Fig. 3-20 shows the schematic representation
and the application illustration. The overarching goal of this project is to create a truly
novel approach for recording and manipulating neural activity: wireless implantable and
addressable nanoscale neural radio frequency identification (NanoNeuroRFID) devices for
large-scale neural magnetic recording and modulation.
At the core of these NanoNeuroRFIDs is the magnetoelectric antenna array which is
able to both sense subtle shift in magnetic fields and to generate steep magnetic field
gradients. The size of these antennas allows them to be integrated with RF integrated
circuits for both data and energy transmission in a package. This small size and the fact
that the entire device is wireless for both data and power transfer will allow for the
NanoNeuroRFIDs be either incorporated on thin sheets or implanted directly into brain
tissue. Additionally, this system would fuel the development of compact closed loop
chronic implants for deep brain stimulation (DBS) and for brain-computer interface (BCI)
applications. A demonstration of even an equivalence would launch the use of these
83
advanced, lower cost, microfabricated NanoNeuroRFID devices for a wide range of basic
neuroscience, neurological and neuropsychiatric diseases.
Fig. 3-20 Schematic of the wireless implantable nanoscale neural radio frequency identification
(NanoNeuroRFID) system with a bi-directional communication link for a capacity of 100~1000 implanted
elements.
84
3.4.2 Proposed Architecture
Fig. 3-21 (a) Proposed architecture of the implantable NanoNeuroRFID with energy harvesting, clock
source, and RF transmission capability. (b) The architecture of the RF transceiver for external wireless
power transfer and time-shared neural recording.
85
The Nano-RFID is designed to perform three major functionalities: (1) wireless
magnetoelectric resonant energy harvesting from the extracranial RF source; (2) neural
magnetic sensing; (3) neural magnetic stimulation; all with a very simple circuit
architecture such that the system area and power consumption are small enough for a brain
implant. Energy harvesting will remove the need for batteries. Once enough energy is
harvested, the system can perform neural magnetic sensing and neural magnetic
stimulation through the magnetoelectric antenna. An on-chip real-time clock (RTC) is
developed, which will synchronize the nano-RFID as well as keep time for transmitting
data and neural stimulation at regular intervals or as needed. Fig. 3-21 shows the simple
architecture of the proposed implantable NanoNeuroRFID system including the Nano-
RFID and the external transceiver.
In Fig. 3-21 (a), an on-chip RF rectifier circuit will be used to harvest energy from the
received RF signal. The rectifier will generate a DC output voltage stored on a capacitor,
which will act as a buffer to supply energy to the on-chip circuits, including during RF
transmission. The ME antenna will be reconfigured as an RF oscillator with a higher output
power mode for neural stimulation. The advantage of having stored energy on the capacitor
provides us with an ability to perform neural stimulation with the known energy and for a
known duration of time in a very precise manner. In the presence of multiple
NanoNeuroRFIDs, the external data acquisition device needs to synchronize the data
collection in a time-shared fashion. A precise on-chip real-time clock is needed for the
NanoNeuroRFID, which can initiate neuron sensing and RF transmission at a precise
interval of time in the correct time-slot provided by the external data device.
86
The transceiver architecture is developed to communicate with NanoNeuroRFIDs
wirelessly using a compact external device showing in Fig. 3-21 (b). In the transmitter
section, an oscillator and amplifier will drive the output power for wireless power transfer
to the Nano-RFIDs with the magnetoelectric antennas and energy harvesting circuits. The
Nano-RFIDs are simultaneously charged through wireless power transmission from the
external transceiver. Afterward, the external transceiver will receive data transmissions
from NanoNeuroRFIDs, where the “on” signal indicates the detection of neuron firing and
the “off” signal indicates neuron inactivity. A code programmed into the timing circuit of
each NanoNeuroRFID will serve as its address in the time-sharing scheme. The external
transceiver is designed with commercial off-the-shelf components and assembled on a
custom printed circuit board to allow timely completion. Similar to other single-chip
transmitters and receivers for medical applications, the design is anticipated to be
implemented as a custom system-on-a-chip.
3.4.3 Innovation
Compared to neural electrical sensing based on differential voltages from neural
probes or wireless implants, these wireless NanoNeuroRFIDs are based on neural magnetic
sensing and have several advantages: (1) neural magnetic sensing is not referential, which
enables significantly more compact NanoNeuroRFIDs; (2) neural magnetic sensing
enables sensing individual neuronal activity, which allows for better separation of
individual neuronal activity and detection of more neurons; (3) it is easy to create safe and
cheap NanoNeuroRFID implants coated with bio-compatible polymer films such as
Parylene; (4) the same technology can be used for animals and human, allowing for direct
87
comparisons and easier translation of animal to human information; and (5) compact size
allows for distributed, addressable, high channel count use in the parenchyma, pial surface,
extradural, and in both central and peripheral nervous tissue. In summary, these
NanoNeuroRFIDs will be the first kind of untethered implants for large-scale neural
magnetic recording and modulation, which provide unprecedented opportunities for (1)
large-scale neural network recording capabilities in vitro and in vivo with 100 – 1000 of
individual recorder/stimulators; (2) new tools for circuit manipulation; (3) large-scale
neural magnetic sensing and stimulation; and (4) directly linking neural activity to behavior.
88
3.5 Summary
Recent miniaturized antennas comparison at VHF and UHF range is summarized in
Table 3-2. Conventional antennas have been developed for decades and act as critical
components widely used in smartphones, tablets, radio frequency identification systems,
radars, wireless communication, etc. Here we proposed a future antenna miniaturization
mechanism, Magnetoelectric Antennas, to open up more possibilities of application due to
their unique and particular properties. We have demonstrated the ME antennas based on
NPR and FBAR structures with new acoustically actuated receiving and transmitting
mechanism. These ME antennas are excellent sensors and radiators for EM waves.
Different modes of vibration are designed, which are controlled by the geometry design of
the ME antennas for realizing both VHF (60 MHz) and UHF (2.525 GHz) operation
frequencies. Both NPR and FBAR resonator antennas can be fabricated on the same Si
wafer with the same microfabrication process, which allows for the integration of
broadband ME antenna arrays from tens of MHz (NPR with large W) to tens of GHz (FBAR
with thinner AlN thickness) on one chip by simulation and device geometry design. A bank
of multi-frequency MEMS resonators can be connected to a CMOS oscillator circuit for
the realization of reconfigurable antennas. With the advantages of the high magnetic field
sensitivity in near field and the highest passive antenna gain within all nano-scale antennas
at the similar frequency range, the ME antenna with 1 to 2 orders reduced size, integrated
capability to CMOS technology, and ground plane immunity from metallic surface or the
human body has a bright future for bio-implantable, wearable antennas, internet of things,
etc.
89
Table 3-2: Miniaturized UHF Antennas Comparison
f (GHz) T Footprint Gain (dBi) IC GPI Ref
0.403 0/585 0.01680 × 0.01680 -32 No No [82]
0.433 0/8 0.1250 × 0.1250 0 No No [83]
0.8 0/235 0.9600 × 0.4880 7 No No [84]
0.915 0/417 0.180 × 0.100 -1 No No [85]
0.915 0/216 0.1280 × 0.0600 0.91 No No [86]
1.574 0/1.31 0.27370 × 0.22630 3.57 No No [87]
1.649 0/142 0.0770 × 0.0770 0.8 No No [88]
2.41 0/24.9 0.1450 × 0.1370 2.3 No No [89]
2.45 0/4.3 0.120 × 0.120 9.5 No No [90]
2.45 0/19.7 0.1020 × 0.1020 4.34 No No [91]
2.45 0/610000 0.0360 × 0.0360 -18 Yes No [92]
2.53 0/98745 0.00670 × 0.00590 -18 Yes Yes ME
t = thickness, IC = Integrated Capability, GPI = Ground Plane Immunity
90
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4. NEMS ME Bandpass Filters with Dual H- and
E- Field Tunability
4.1 Introduction
RF Band-pass filters based on thin-film bulk acoustic resonators (FBAR) show a very
high-quality factor, compact size, low loss, preferable temperature stability and processing
compatible with the CMOS IC, which are widely used in smartphones, E-readers, etc.
However, these FBAR based band-pass filters have a fixed frequency of operation, which
lead to bulky and expensive electronics for reconfigurable communication systems.
Strong magnetoelectric (ME) coupling between its piezoelectric and piezomagnetic
phase, which can lead to various device application at room temperature have attracted an
increasing amount of research interest. Ultra-sensitive magnetometer has been
demonstrated based on the direct ME coupling or magnetic field control of electrical
polarization. By converse ME coupling, the magnetic anisotropy of the ferromagnetic
phase can be controlled by the applied voltage across the piezomagnetic layer. A significant
shift of the magnetic anisotropy field has been observed and probed by ferromagnetic
resonance with the applied voltage on the ferroelectric PMN-PT substrate. This strong
tunability can be used in the application of voltage controlled spintronics and
reconfigurable microwave/RF components with ultra-low power consumption.
Recently, researchers have demonstrated a variety of magnetoelectric sensors [1]-[3]
such as sensors with a high magnetoelectric coefficient of 737 V/cm.Oe at the electro-
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mechanical resonance frequency of 753 Hz using FeCoSiB/AlN thin film heterostructures
at a bias magnetic field of 6Oe; Magnetoelectric nano-plate resonator (NPR), due to the
delta-E effect (the modulation of the Young’s modulus of magnetic materials with
magnetic field) led to a new detection mechanism for ultra-sensitive self-biased RF NEMS
magnetoelectric sensor with a low limit of detection of DC magnetic fields of 300
picoTelsa.
In this chapter, I will present the miniaturized RF tunable band-pass filters based on
magnetoelectric NEMS coupled ring-shaped FBAR resonators with contour mode of
transmission [4]. Due to the strong magnetoelectric effect between the piezomagnetic
FeGaB and piezoelectric AlN thin film on the resonant body, the acoustic wave can be
strongly coupled with the radiated electromagnetic wave. A return loss of -11.15 dB and
insertion loss of 3.57 dB with a high-quality factor of 252 can be achieved at 93.165MHz.
The band-pass filters perform sensitive magnetic field dependence with ~0.5% magnetic
field tunability of the operation frequency.
103
4.2 Design and Fabrication
Thin-film bulk acoustic wave resonators (FBAR) with the reduction of size and power
consumption has caught the attention of many industrial and academic research groups all
over the world. Specifically, by applying RF electric field to the MEMS resonator, the
mechanical resonance would induce alternating strain wave/acoustic wave. When this
acoustic wave transfers to port 2, the dynamic voltage/charge would be generated due to
the direct piezoelectric coupling. The resonance frequency of FBAR resonators operating
in fundamental, longitudinal mode is mainly determined by the thickness of the acoustic
layer. However, in these coupled ring-shaped resonators, the contribution of the
transmission is from contour mode which is mainly determined by the width of the coupled
Fig. 4-1. Schematic of the layered structure of the NEMS ME band-pass filter.
104
Fig. 4-2. (a) Simulated admittance amplitude curve of the NEMS coupled ring-shaped FBAR resonator
showing the electromechanical resonance frequency of ~92MHz. (b) Simulated signal transmission from
port 1 to port 2.
105
structure showing in Fig. 4-1. The demonstrated band-pass filter is based on highly
sensitive magnetoelectric resonators with piezomagnetic FeGaB/Al2O3 multilayer and
piezoelectric AlN thin film heterostructures.
Phase locking of two coupled resonators with two ring structures with a gap of 2um
was designed. This phase-locking technique has been already achieved in phased-locked
spin torque nano-oscillators exhibiting enhanced quality factor Q [5]-[6]. The simulation
simply uses the same piezoelectric module in section 3.1.3 but in both time and frequency
domain. Different designs and shapes were simulated and tested. Two coupled ellipse rings
the structure was selected here because of the high-quality factor and the clear peak without
spurious peaks due to the smooth curve from both simulation and measured results. Fig. 4-
2 (a) is the simulated admittance amplitude curve of the NEMS coupled ring-shaped FBAR
resonators showing the designed electromechanical resonance frequency of ~92MHz. Fig.
4-2 (b) is the illustration of the transmission from port 1 to port 2 by displacement
simulation as time goes by.
The NEMS magnetoelectric band-pass filter was fabricated using the same five-mask
microfabrication process as the ME antenna. A 50 nm thick Platinum (Pt) film was sputter-
deposited on top of the Si substrate to define the bottom electrodes in Fig. 4-3 (a). Then,
the 200 nm AlN film was sputter-deposited and vias etched by H3PO4 to access the
electrodes in Fig. 4-3 (b). After that, the AlN film was etched by Inductively Coupled
Plasma (ICP) etching to define the shape of the resonator in Fig. 4-3 (c). A 100 nm thick
gold (Au) film was evaporated to form the top ground in Fig. 4-3 (d). Finally, 200 nm thick
FeGaB/Al2O3 multilayer layer was deposited by Physical Vapor Deposition (PVD) with a
100 Oe in-situ magnetic field bias applied during the deposition along the anchor direction
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of the device to pre-orient the magnetic domains. Then, the structure was released by XeF2
isotropic etching of the Silicon substrate in Fig. 4-3 (e). The complete removal of Si
substrate underneath the RF NEMS resonator results to a strong magnetoelectric coupling
and a high-quality factor of the mechanical resonance by diminishing the substrate-
clamping effect. The complete ME band-pass filter optical and Scanning Electron
Microscopy (SEM) images are shown in Fig. 4-4.
Fig. 4-3 The fabrication process of NEMS ME band-pass filter.
107
Fig. 4-4 Optical and SEM images of the fabricated NEMS ME band-pass filter with silicon substrate
released.
108
4.3 Results and Discussion
4.3.1 Modified Equivalent Circuit Modeling
The on-chip NEMS band-pass filter was measured by network analyzer connecting to
the electrode pads by RF probes. The transmission parameter S11 was acquired and
converted to admittance amplitude. The available power at the network analyzer port was
set to -12 dBm, and the IF bandwidth was 50 Hz which results to a trace noise magnitude
of 0.002 dB. The admittance curve and the Modified Butterworth–van Dyke (MBVD)
model fitting of the NEMS magnetic field resonator are depicted in Fig. 4-5 (a) showing
an electromechanical resonance frequency of 93 MHz which is similar to the simulation in
Figure 4-2 (a). Fig. 5 (b) shows the MBVD equivalent electrical circuit of the resonator, in
which Q is the quality factor of the resonator, Rs is the resistance of metal electrodes and
contact resistance, Rop is the parasitic resistance from substrate, C0 is the device capacitance
from the Pt/AlN/FeGaB stack, Cm, Lm, and Rm are the motional capacitance, inductance,
and resistance, respectively.
The resonance frequency of the ME NEMS resonator can be expressed by:
𝑓0 =1
2𝑤𝑒𝑞√
𝐸𝑒𝑞
𝜌𝑒𝑞 (4-1)
Which weq is the equivalent width of the ring electrode, Eeq is the equivalent Young’s
Modulus, and ρeq is the equivalent density of the resonator.
109
Fig. 4-5. (a) Measured Admittance curve and Butterworth–van Dyke (BVD) model fitting of the fabricated
NEMS magnetic field resonator. (b) The BVD equivalent electrical circuit of the resonator.
110
4.3.2 Delta E Effect
The admittance curve Strong ME coupling in the AlN/(FeGaB/Al2O3) ×10 based RF
NEMS resonator was demonstrated in a DC bias field with the induced change in the
electromechanical resonance frequency, which was attributable to the bias magnetic field
induced Young’s modulus change in FeGaB, or the delta-E effect [7]. A magnetostrictive
strain can be induced in the FeGaB layer under a DC magnetic field through the delta-E
effect, which led to a changed Young’s modulus of the FeGaB film, and therefore a
changed equivalent Young’s modulus of the NEMS magnetoelectric resonator. The
electromechanical resonance frequency and the admittance amplitude of the AlN resonator
were varied through DC magnetic fields.
The delta-E effect is determined by the total anisotropy:
𝛫𝑡𝑜𝑡 = 𝛫𝜎 + 𝛫𝑠ℎ𝑎𝑝𝑒 + 𝛫𝑖𝑛𝑑 (4-2)
Where 𝛫𝜎 is magnetoelastic anisotropy, Kshape is the shape anisotropy, and Kind is the
induced anisotropy. The extreme small DC magnetic field was applied along the length
direction of the device by a home-made Helmholtz coil driven by a precision current source.
Fig. 4-6 (a) shows the admittance curve of the NEMS ME band-pass filter at various
DC bias magnetic fields (0 Oe and 90 Oe) applied along the width direction of the resonator.
Both resonance frequency and peak admittance amplitude are plotted in Fig. 4-6 (b)
exhibited a similar trend with DC bias magnetic field, which first decreased with the
increase of bias field, reaching minimum values at a bias field of 90 Oe. For the NEMS
band-pass filter, the change of Young’s module reached the maximum under the ~90Oe.
111
Fig. 4-6. (a) Measured Admittance curve at various bias DC magnetic fields. (b) Resonance frequency and
peak admittance amplitude as a function of the DC magnetic field.
112
The admittance versus DC magnetic field can be expressed as:
𝑑𝑌
𝑑𝐻=
𝑑𝑌
𝑑𝑓
𝑑𝑓
𝑑𝐻=
𝑑𝑌
𝑑𝑓(
𝑑𝑓
𝑑𝐸
𝑑𝐸
𝑑𝐻− 𝜈
𝑑𝑓
𝑑𝑊
𝑑𝑊
𝑑𝐻) (4-3)
Where Y is the admittance amplitude, E is the Young’s modulus of the resonator, 𝜈 is the
Poisson’s ratio of the magnetic materials, and W is the width of the ring electrodes. The
dY/df term is the slope of admittance amplitude versus frequency between series and
parallel resonances where it reaches the maximum; while df/dH can be seen as the
resonance frequency tunability to the DC magnetic fields. The frequency sensitivity can be
simplified from (4-1) and (4-3) equations as:
𝑑𝑓
𝑑𝐻=
𝑑𝑓
𝑑𝐸
𝑑𝐸
𝑑𝐻=
𝑓
2𝐸
𝑑𝐸
𝑑𝐻 (4-4)
We can observe that higher resonance frequency would results in higher tunability to
the magnetic field. The 93.165 MHz resonant NEMS band-pass filter was fabricated to
reach high sensitivity due to the sensitivity to frequency relation.
4.3.3 Magnetoelectric Coupling
Converse magnetoelectric coupling was also achieved through electric field induced
effective magnetic field by applying different DC voltages superimposed to the RF signal
via a bias Tee on the piezoelectric AlN layer. However, the shifting of the frequency due
to magnetostrictive strain is negligible comparing with delta-E effect since the
magnetostriction coefficient of FeGaB of 70 ppm is much smaller than the delta-E effect
induced percentage change in Young’s modulus of magnetostrictive soft magnetic films
which can be up to 20%~30%. An applied DC voltage on the AlN layer led to a
113
piezoelectric strain in the AlN and in FeGaB, which resulted in a strain induced the
effective magnetic field in FeGaB layer. A positive DC voltage led to the decreased
resonance frequency of the NEMS resonator; while a negative DC voltage resulted in
enhanced resonance frequency, which can be attributed to the change of the stiffness of the
resonator by the induced piezoelectric stress [8]. When the piezoelectric strain in AlN was
transferred to the magnetic materials, an effective anisotropy field 𝐻𝑒𝑓𝑓 can be expressed
as:
𝐻𝑒𝑓𝑓 = −3𝜆𝜎
𝑀𝑠 (4-5)
Fig. 4-7. Resonance frequency as a function of DC Bias Voltage.
114
Where 𝜎 is the in-plane stress transferred from piezoelectric AlN layer to magnetostrictive
FeGaB layer; 𝜆 is the in-plane magnetostriction coefficient for magnetic phase; 𝑀𝑠 is the
saturation magnetization. The linear relationship between the magnetic transition fields and
the applied voltages results from the linear piezoelectricity of AlN, indicating a converse
magnetoelectric coupling between the piezomagnetic phase and the piezoelectric phase.
The tunability of 2.3 kHz/1V is achieved as shown in Fig. 4-7.
Fig. 4-8. NEMS ME band-pass filter measured return loss S11 and insertion loss S21 at zero bias field.
90 91 92 93 94 95 96-12
-11
-10
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
s11
s21
Frequency (MHz)
s1
1 (
dB
)
-35
-30
-25
-20
-15
-10
-5
0
s2
1 (
dB
)
115
Fig. 4-9. (a) S11 performance, and (b) S21 performance with different dc magnetic fields.
116
4.3.4 Band-pass Filter Performance
S-parameters performance with applying dc magnetic field is shown in Fig. 4-8. The
resonant frequency is at 93.165MHz with insertion loss of 3.57dB and return loss of ~ -
11.15dB. Ultra-sensitive frequency tunability of ~50kHz/10Oe with a high-quality factor
of 252 can be achieved showing in Fig. 4-9. Unlike the piezomagnetic coefficient which is
almost zero at zero bias magnetic field, the change of Young’s modulus due to magnetic
domain wall motion is not zero at zero bias magnetic field, which makes the NEMS
magnetoelectric band-pass filter a self-biased device.
117
4.4 Summary
In summary, the integrated tunable RF band-pass filter based on NEMS
Magnetoelectric resonators is designed, fabricated, and measured. Ultra-sensitive
frequency Dual H- and E- field tunability of 50kHz/10Oe and 2.3 kHz/1V with a high-
quality factor of 252 are achieved. The tunable RF band-pass filters based on NEMS
magnetoelectric Resonators are compact, power efficient and readily integrated with
CMOS technology. It represents a new class of tunable ultra-sensitive magnetometers and
filters for DC magnetic fields, and will definitely be the future highlight in RF and
microwave reconfigurable communication systems.
118
4.5 Reference
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fields using magneto-electric sensors at room temperature. Appl. Phys. Lett. 88,
062510 (2006).
2 G. Sreenivasulu, U. Laletin, V. M. Petrov, V. V. Petrov and G. Srinivasan: A
permendur-piezoelectric multiferroic composite for low-noise ultra-sensitive
magnetic field sensors. Appl. Phys. Lett. 100, 173506 (2012).
3 T. Nan, Y. Hui, M. Rinaldi, and N. X. Sun: Self-Biased 215MHz Magnetoelectric
NEMS Resonator for Ultra-Sensitive DC Magnetic Field Detection. Sci. Rep. 3, 1985
(2013).
4 H. Lin, T. Nan, Z. Qian, Y. Gao, Y. Hui, X. Wang, R. Guo, A. Belkessam, W. Shi, M.
Rinaldi, N. X. Sun: Tunable RF band-pass filters based on NEMS magnetoelectric
resonators. IEEE MTTS Int. Microw. Symp., San Francisco, CA, May 22-27 (2016).
5 F.B. Mancoff, N.D. Rizzo, B.N. Engel, and S. Tehrani: Phase-locking in double-
point-contact spin-transfer devices. Nature 437, 393, (2005).
6 S. Kaka, M. R. Pufall, W. H. Rippard, T. J. Silva1, S. E. Russek, and J. A. Katine:
Mutual phase-locking of microwave spin torque nano-oscillators. Nature 437, 389-
392 (2005).
7 A. Ludwig, and E. Quandt: Optimization of the ΔE effect in thin films and multilayers
by magnetic field annealing. IEEE. Trans. Magn. 38, 2829-2831 (2002).
8 R. B. Karabalin, M. H. Matheny, X. L. Feng, E. Defaÿ, G. Le Rhun, C. Marcoux, S.
Hentz, P. Andreucci, and M. L. Roukes: Piezoelectric nanoelectromechanical
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119
5. Conclusion
Multiferroic composite materials and devices with two or more of the ferroic properties
have attracted interests due to the demonstrated unique functionalities and ME coupling
performance which is several orders of magnitude higher than single phase multiferroics.
Progress has been made to demonstrate giant voltage control of the ferromagnetic
resonance frequency in the composite structures. The integrated RF/microwave devices are
compact, lightweight, power efficient, and provide excellent opportunities for new
reconfigurable RF/microwave applications for spintronics, and magnetic field sensing.
Advancements in multiferroic composite materials will depend on achieving strong ME
coupling for devices in which the performance is dependent on the material.
The future antenna miniaturization has also been demonstrated by ME antennas, which
has been a critical obstacle for conventional antennas due to the EM wavelength. The
strong ME coupling induces RF magnetic currents to radiate EM waves at a wide range of
frequencies allowing for acoustically actuated ME antennas with receiving and
transmitting capabilities on the nanoscale. The operating frequencies span from several
MHz to tens of GHz, which are controlled by the calculated geometric design of the
resonators with in-plane or out-of-plane vibrational modes. Future designs will focus on
impedance matching and geometry optimization of ME antenna arrays to achieve higher
gain for increased communication reliability.
Novel Implantable Smart Magnetoelectric NanoRFIDs for Large-Scale Neural
Magnetic Recording and Modulation are also introduced. These NanoNeuroRFIDs will be
the first kind of untethered implants for large-scale neural magnetic recording and
120
modulation, which provide unprecedented opportunities for (1) large-scale neural network
recording capabilities in vitro and in vivo with 100-1000 of individual recorder/stimulators;
(2) new tools for circuit manipulation; (3) large-scale neural magnetic sensing and
stimulation; and (4) directly linking neural activity to behavior.
The ME antenna won the NASA Tech Briefs - Create the Future Design Contest: First
Prize (in Electronics/Sensors/IoT Category) with over 800 entries from 60 countries in
2018, which is sponsored by Comsol, Intel, Analog Devices, Mouser Electronics and is
featured in NASA Tech Briefs Magazine with more publicity and exposure to the industry
and investors. The publication in Nature Communications was widely cited in different
news media, including NATURE (Ultra-small antennas point way to miniature brain
implants), SCIENCE (Mini-antennas could power brain-computer interfaces, medical
devices), news on various websites and newspapers in different languages, TV interview.
These NEMS Antennas open up more possibilities of application due to their unique
and particular properties. With the advantages of the high magnetic field sensitivity in near
field and the highest antenna gain within all nano-scale passive antennas at the similar
frequency range, the antenna with 1 to 2 orders reduced size, integrated capability to
CMOS technology, and ground plane immunity from metallic surface or the human body
has a bright future for bio-implantable, wearable antennas, internet of things, etc.
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