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ADVANCE CONTROL SYSTEM ENGINEERING


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ADVANCED CONTOL SYSTEMS

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ADVANCED CONTROL SYSTEMS (PEEC5414) Module ‐I : (15 Hours) Discrete ‐ Time Control Systems :

Introduction: Discrete Time Control Systems and Continuous Time Control Systems, Sampling

Process. Digital Control Systems: Sample and Hold, Analog to digital conversion, Digital to analog conversion. The Z‐transform : Discrete ‐Time Signals, The Z‐transform, Z‐transform of Elementary functions, Important properties and Theorms of the Z‐transform. The inverse Ztransform, Z‐Transform method

for solving Difference Equations.

Z‐Plane Analysis of Discrete Time Control Systems: Impulse sampling & Data Hold, Reconstruction of Original signals from sampled signals: Sampling theorm, folding, aliasing. Pulse Transfer function:

Starred Laplace Transform of the signal involving Both ordinary and starred Laplace Transforms;

General procedures

for

obtaining

pulse

Transfer

functions,

Pulse

Transfer

function

of

open

loop

and

closed loop systems.

Mapping between the s‐plane and the z‐plane, Stability analysis of closed loop systems in the z‐plane: Stability analysis by use of the Bilinear Transformation and Routh stability critgion, Jury

stability. Test. Book No. 1: 1.1; 1.2; 1.4; 2.1; 2.2; 2.3; 2.4; 2.5; 2.6; 3.2; 3.4; 3.5; 4.2; 4.3. Module ‐II : (15 Hours) State Variable Analysis & Design:

Introduction: Concepts of State, State Variables and State Model (of continuous time systems):

State Model of Linear Systems, State Model for Single ‐Input ‐Single ‐Output Linear Systems,

Linearization of the State Equation. State Models for Linear Continuous – Time Systems: State ‐Space Representation Using Physical Variables, State – space Representation Using Phase Variables,

Phase variable formulations for transfer function with poles and zeros, State – space Representation using Canonical Variables, Derivation of Transfer Function for State Model. Diagonalization:

Eigenvalues and Eigenvectors, Generalized Eigenvectors.

Solution of State Equations: Properties of the State Transition Matrix, Computation of State Transition Matrix, Computation by Techniques Based on the Cayley‐Hamilton Theorem, Sylvester’s Expansion theorm. Concepts of Controllability and Observability: Controllability, Observability, Effect of Pole ‐zero Cancellation in Transfer Function. Pole Placement by State Feedback, Observer

Systems. State Variables and Linear Discrete – Time Systems: State Models from Linear Difference

Equations/z ‐transfer Functions, Solution of State Equations (Discrete Case), An Efficient Method of Discretization and Solution, Linear Transformation of State Vector (Discrete ‐Time Case), Derivation

of z‐Transfer Function from Discrete ‐Time State Model. Book No. 2: 12.1 to 12.9. Module ‐III : (12 Hours) Nonlinear Systems :

Introduction : Behaviour of Non linear Systems, Investigation of nonlinear systems. Common Physical Non Linearities: Saturation, Friction, Backlash, Relay, Multivariable Nonlinearity. The Phase Plane Method: Basic Concepts, Singular Points: Nodal Point, Saddle Point, Focus Point, Centre or Vortex Point, Stability of Non Linear Systems: Limit Cycles, Construction of Phase


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Trajectories: Construction by Analytical Method, Construction by Graphical Methods. The Describing Function Method: Basic Concepts: Derivation of Describing Functions: Dead ‐zone and Saturation, Relay with Dead ‐zone and Hysteresis, Backlash. Stability Analysis by Describing Function

Method: Relay with Dead Zone, Relay with Hysteresis, Stability Analysis by Gain ‐phase Plots. Jump Resonance. Liapunov’s Stability Analysis: Introduction, Liapunov’s Stability Critrion: Basic Stability

Theorems, Liapunov Functions, Instability. Direct Method of Liapunov & the Linear System: Methods of constructing Liapunov functions for Non linear Systems.

Book No. 2: 13.1 to 13.4; 15.1 to 15.10.

Text :

1. Discrete ‐Time Control System, by K.Ogata, 2nd edition (2009), PHI. 2. Control Systems Engineering, by I.J. Nagrath and M.Gopal., 5th Edition (2007 / 2009), New Age

International (P) Ltd. Publishers.

Reference : 1. Design of Feedback Control Systems by Stefani, Shahian, Savant, Hostetter, Fourth Edition (2009),

Oxford University Press. 2. Modern Control Systems by K.Ogata, 5th Edition (2010), PHI. 3. Modern Control Systems by Richard C. Dorf. And Robert, H.Bishop, 11th Edition (2008), Pearson

Education Inc. Publication.

4. Control Systems (Principles & Design) by M.Gopal, 3rd Edition (2008), Tata Mc.Graw Hill

Publishing Company Ltd. 5. Control Systems Engineering by Norman S.Nise, 4th Edition (2008), Wiley India (P) Ltd.


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Module 1: Discrete-Time Control systems

Lecture Note 1 (Introduction)Continuous time Control System:

In continuous time control systems, all the system variables are continuous signals. Whether

the system is linear or nonlinear, all variables are continuously present and therefore known

(available) at all times. A typical continuous time control system is shown in Figure below.

(Closed loop continuous-time control system )

Discrete time Control System:

Discrete time control systems are control systems in which one or more variables can change

only at discrete instants of time. These instants, which may be denoted by kT(k=0,1,2,…)

specify the times at which some physical measurement is performed or the times at which

the memory of a digital computer is read out.


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mode is the hold mode in which the capacitor voltage holds constant for a specified time

period.

(sample and hold circuit)

Digital to Analog Conversion:

Digital-to-analog conversion is simple and effectively instantaneous. Properly weighted

voltages are summed together to yield the analog output. For example, in Figure below, three

weighted voltages are summed. The three-bit binary code is represented by the switches.

Thus, if the binary number is 110 2, the center and bottom switches are on, and the analog

output is 6 volts. In actual use, the switches are electronic and are set by the input binarycode.

(Digital to Analog converter)


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Analog to Digital Conversion:

The process by which a sampled analog signal is quantized and converted to a binary numberis called analog to digital conversion. The A/D converter performs the operation of sample-

and –hold, quantizing and encoding.

The simplest type of A/D converter is the counter type. The basic principle on which it works

is the clock pulses are applied to the digital counter in such a way that the output voltage of

the D/A converter (that is part of the feedback loop in the A/D converter) is stepped up one

least significant bit at a time, and the output voltage is compared with the analog input

voltage once for each pulse. When the output voltage has reached the magnitude of the input

voltage, the clock pulses are stopped. The counter output voltage is then the digital output.

(Counter type analog to digital converter)


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Lecture Note 3(The Z-Transform)

Z-transform:Z-transform is a mathematical tool commonly used for the analysis and synthesis of discrete

time control systems. The role of Z-transform in discrete time systems is similar to that of the

Laplace transform in continuous systems. In considering Z-transform of a time function x(t),

we consider only the sampled values of x(t), i.e., x(0) , x (T), x (2T)....... where T is the

sampling period.

For a sequence of numbers x(k)

The above transforms are referred to as one sided z-transform. In one sided z-ransform, we

assume that x(t) = 0 for t < 0 or x(k) = 0 for k < 0. In two sided z-transform, we assume that

− 1 < t < 1 or k =,±1,±2,±3, ........

The one sided z-transform has a convenient closed form solution in its region of convergence

for most engineering applications. Whenever X(z), an infinite series in z -1, converges outside

the circle |z| = R , where R is the radius of absolute convergence it is not needed each time to

specify the values of z over which X(z) is convergent.

I.e. for |z| > R convergent and for |z| < R divergent.


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Z-Transforms of some elementary functions:(i) Unit step function is defined as:

Unit step sequence is defined as

Assuming that the function is continuous from right, the Z-transform is:

The above series converges if |z| > 1.

(ii) Unit ramp function is defined as:

Again

The Z-transform is:

The above series converges if |z| > 1.

(iii) Exponential function is defined as:

Again

The Z-transform is:


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Lecture Note 4(Theorems & properties of Z-Transform)

Table for z- and s-transform:

Important properties & theorems of z-transform:

1. Multiplication by a constant: , where

2. Linearity: If , then

3. Multiplication by a k :

4. Real shifting: and


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5. Complex shifting:

6. Initial value theorem:

7. Final value theorem:


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Lecture Note 5(Inverse Z-Transform)

Inverse Z-transforms: When F(z), the Z-transform of f(kT) or f(k), is given, the operation that determines the

corresponding x(kT) or x(k) is called inverse Z-transform. It should be noted that only the

time sequence at the sampling instants is obtained from the inverse Z-transform. So the

inverse Z-transform of F(z) yield a unique f(k) but does not yield a unique f(t).

The inverse Z- transform can be obtained by using

(i) Power series method:

Since F(z) is mostly expressed in the ratio of polynomial form

We can immediately recognize the sequence of f(k) if F(z) can be written in form of series

with increasing powers of z -1 .This is easily done by dividing the numerator by the

denominator.

I,e

Here the values of f(k) for k=0,1,2,… can be determined by inspection.

(ii) Partial fraction expansion method:

To expand F(z) into partial fractions, we first factor the denominator polynomial of F(z) and

find the poles F(z):

We then expand F(z)/z into partial fractions so that each term is easily recognizable

in a table of Z-transforms. I.e.


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Taking the inverse Z-transform we get

Lecture Note 6(Z-plane Analysis, sampling &hold)

Impulse Sampling:In case of an impulse sampling, the output of the sampler is considered to be a train of

impulses that begin with t=0, with the sampling period equal to T and the strength of each

impulse equal to sampled value of the continuous-time signal at the corresponding sampling

instant as shown in figure below.


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(Impulse sampler)

The sampler output is equal to the product of the continuous-time input x(t) and the train of

unit impulses δT(t).

The train of unit impulses δT(t) can be defined as

Now the sampler output can be expressed as

In the impulse sampler the switch may be thought of closing instantaneously every sampling

period T and generating impulses x(kT) δ(t-kT).The impulse sampler is a fictitious sampler

and it does not exist in the real world.

Data Hold:

Data-hold is a process of generating a continuous-time signal h(t) from a discrete-time

sequence x(kT).A hold circuit converts the sampled signal into a continuous-time signal,

which approximately reproduces the signal applied to the sampler.

The signal h(t) during the interval kT ≤ t ≤ (k+1)T can be expressed as

Since,

So,


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If the data-hold circuit is an nth-order polynomial extrapolator, it is called an nth order hold.

Zero-Order Hold:

It is the simplest data-hold obtained by putting n=0 in eq(1), gives .It

indicates that the circuit holds the value of h( kT) for kT≤ t < (k + 1) T until the next sample

h((k + 1) T) arrives.

(sampler and zero order hold)

The accuracy of zero order hold (ZOH) depends on the sampling frequency. When T→ 0 , the

output of ZOH approaches the continuous time signal. Zero order hold is again a linear

device which satisfies the principle of superposition.

(Impulse response of ZOH)


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The impulse response of a ZOH, as shown in the above figure can be written as

Lecture Note 7(Reconstruction of signal)

Data Reconstruction:

To reconstruct the original signal from a sampled signal, there is a certain minimum

frequency that the sampling operation must satisfy. Such a minimum frequency is specified

by the sampling theorem.

Sampling Theorem:


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If the sampling frequency ωs = 2 /T ( where T is the sampling period) and ω1 is the highest

frequency component present in the continuous- time signal x(t),then it is theoretically

possible that the signal x(t) can be constructed completely from the sampled signal x (t) if

ωs 2 ω1.

Considering the frequency spectrum of the signal x(t) as shown in the figure below

Then the frequency spectra of the sampled signal can be expressed as


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The above equation gives the frequency spectrum of the sampled signal x (t) as shown in

figure below in which we can see that frequency spectrum of the impulse sampled signal is

reproduced an infinite number of times and is attenuated by the factor 1/T.


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The above figure represents the frequency spectrum for ωs 2 ω1 without any overlap and the

continuous time signal x(t) can be reconstructed from the impulse-sampled signal x (t).

For ωs 2 ω1,as shown in figure below, the original shape of |X(j ω)| no longer appears in the

plot of | X (jω)| because of the superposition of the spectra. As a result the continuous time

signal x(t) can not be reconstructed from the impulse-sampled signal x (t).


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Lecture Note 8(Folding & Aliasing)

Folding:

The overlapping of the high frequency components with the fundamental component in the

frequency spectrum is sometimes referred to as folding and the frequency ωs/2 is often

known as folding frequency . The frequency ωs is called Nyquist frequency .. The figure


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below shows the region where folding error occurs.

Aliasing:

If the sampling rate is less than twice the input frequency, the output frequency will be

different from the input which is known as aliasing . The output frequency in that case is

called alias frequency and the period is referred to as alias period.

Figure-1 shows the frequency spectra of an impulse-sampled signal x (t), where ωs 2 ω1.

Now the frequency spectrum at an arbitrary frequency ω2 in the overlap region includes

components not only at frequency ω2 but also at frequency ωs- ω2 (in general n ωs ± ω2, where

n is an integer).


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(Figure-1:frequency spectra of an impulse-sampled signal)

When the composite spectrum is filtered by a low pass filter, the frequency component at

ω=nωs ± ω2 will appear in the output as if it were a frequency component at ω= ω2. The

frequency ω=nωs ± ω2 is called an alias of ω2 and this phenomena is called aliasing.

Lecture Note 9(Pulse Transfer function)

Pulse Transfer Function:The Pulse transfer function relates z-transform of the output at the sampling instants to the Z-

transform of the sampled input.

(Block Diagram of a system with sampled input and output )

Now

Again


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Using real convolution theorem

G(z) as given by the above equation ,the ratio of output C(z) and the input R(z) is called the

pulse transfer function .

Starred Laplace Transform of the Signal involving Both

Ordinary and starred Laplace Transform:

(Block Diagram of an Impulse sampled system)

The output of the system is C(s ) = G (s )R (s ). The transfer function of the above system is

difficult to manipulate because it contains a mixture of analog and digital components.

Thus,it is desirable to express the system characteristics by a transfer function that relatesr (t) to c (t), a fictitious sampler output as shown in the above figure.

Now


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Since c(kT) is periodic,

Siliarly,

Again,

Since is periodic, .Thus

By defining

Then

The above equation is referred as the starred transfer function . By substituting in

the previous expression we will directly get the z-transfer function G (z) as


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Lecture Note 10(Pulse T.F. of Open Loop &Closed Loop Systems)

Pulse Transfer function of Cascaded Elements:

(Cascaded elements separated by a sampler)

The input-output relations of the two systems G1 and G2 are described by D(z)=G 1(z)R(z)

and C(z) = G2(z)D(z), So the input-output relation of the overall system is C(z) =

G1(z)G2(z)R(z).

So it can be concluded that the z-transfer function of two linear system separated by asampler are the products of the individual z-transfer functions.


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(Cascaded elements not separated by a sampler)

Here the input-output relations of the two systems G1 and G2 are described by C(s ) = G1(s )G2(s )R (s ) and the output of the fictitious sampler is

C(z) = Z [G1(s )G2(s )]R(z) z-transform of the product G1(s )G2(s ) is denoted as Z [G1(s )G2(s )] = G1G2(z) = G2G1(z)

One should note that in general G1G2(z) ≠G1(z)G2(z), except for some special cases. The

overall output is thus, C(z) = G1G2(z)R(z).

Pulse Transfer Functions of Closed Loop Systems:

(closed loop system with a sampler in the forward path)

For the above closed loop system, the output of the sampler is regarded as an input to the

system. The input to the sampler is regarded as another output. Thus the input-output

relations can be formulated as


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Taking starred Laplace transform on both sides

Or

Since

Lecture Note 11(Mapping between s & z-plane)

Mapping between s-plane and z-plane:

The absolute stability and relative stability of the LTI continuous-time closed loop control

system are determined by the locations of closed loop poles in the s-plane. Since the complex

variables z and s are related by , the pole and zero locations in the z-plane are related

to the pole and zero locations in the s-plane. So the stability of LTI discrete-time time closed

loop control system are determined in terms of location of poles of the closed-loop pulse

transfer function.

A pole in the s plane can be located in the z plane through the transformation

Since

So

As is negative in the left half of the s-plane, so the left half of s-plane corresponds to


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(shown by region A inside of an unit circle).

For the axis ,it corresponds to .That is the imaginary axis in the s-plane

corresponds to the unit circle in the z-plane (shown by region B).

For positive values of in the right half of the s-plane corresponds to

.(shown by region C outside the unit circle)

(mapping regions of s-plane on to the z-plane)

Thus, a digital control system is

(1) Stable if all poles of the closed-loop transfer function are inside the unit circle on

the z-plane,

(2) Unstable if any pole is outside the unit circle and/or there are poles of multiplicity

greater than one on the unit circle, and

(3) Marginally stable if poles of multiplicity one are on the unit circle and all other

poles are inside the unit circle.


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Lecture Note 12(Stability Analysis)

Stability Analysis of closed loop system in z-plane:

Consider the following closed loop pulse transfer function of an linear time invariant single-

input-single-output discrete time control system

The stability of the above system can be determined from the location of closed loop poles in

z-plane which are the roots of the characteristic equation

1. For the system to be stable, the closed loop poles or the roots of the characteristic

equation must lie within the unit circle in z-plane. Otherwise the system would be

unstable.

2. If a simple pole lies at , the system becomes marginally stable. Similarly if a

pair of complex conjugate poles lies on the unit circle in the z-plane, the system ismarginally stable. Multiple poles on unit circle make the system unstable.


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Lecture Note 13(Routh Stability criterion)

Stability Analysis by use of Bilinear Transformation and

Routh Stability Criterion:It is a frequently used method in stability analysis of discrete time system by using bilinear

transformation coupled with Routh stability criterion. This requires ransformation from z-

plane to another plane called w-plane. The bilinear transformation defined by

where a , b , c , d are real constants. If we consider a = b = c = 1 and d = − 1, then thetransformation takes a form


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or ,

This transformation maps the inside of the unit circle in the z-plane into the left half of the w-

plane. Let the real part of w be α and imaginary part be β, so that . The inside of

the unit circle in z-plane can be represented by

Thus inside of the unit circle in z-plane maps into the left half of w-plane and outside of the

unit circle in z-plane maps into the right half of w-plane. Although w-plane seems to be

similar to s -plane, quantitatively it is not same.

In the stability analysis using bilinear transformation, we first substitute in the

characteristics equation

and simplify it to get the characteristic equation in w-plane

Once the characteristics equation P(z)=0 is transformed as Q (w) = 0, Routh stability

Criterion is applied in the same manner as in continuous time systems.


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Lecture Note 14(Jury Stability Test)

Jury Stability Test:

Assume that the characteristic equation P(z) is a polynomial in z as follows,

Where .Then the Jury table can be formed as shown below


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Where,

This system will be stable if the following conditions are all satisfied:

1.

2.

3. for n even and for odd

4. , , …….,


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Module – 2: State Space Analysis

Lecturer: 1

: Introduction to State Space analysis

: Understanding the Concepts of State, State Variable and State Model of continuous time

system

: State Model of Linear systems

Analysis and design of the non-linear, time variant, multivariable continuous system can be

easily done using Modern control system i.e. state – space – based methods, where system

model are directly written in the time domain. Its performance in real application where plant

model is uncertain has been improved using robust control system, i.e. combining the modern

state space and classical frequency domain techniques.


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Where is state vector, is input vector, A is system matrix

and B is input matrix. Similarly is output vector, C is output

matrix and D is transmission matrix.

Lecturer: 2

: State Model of Linear SISO systems

: Linearization of the State Equation

Linearization of the state Equation:

The state equation of a general time – invariant, non-linear continuous

system can be linearized, by considering small variations about an equilibrium point (x o, uo)

i.e. , where , is assumed initial state point and input vector

respectively. is state space.

Since the derivatives of all the state variables are zero at the equilibrium point, the system

continues to lie at the equilibrium point unless otherwise disturbed. The state equation can be


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linearized about the operating point by expanding it into Talyor series and

neglecting terms of second and higher order.

The linearized component equation can be written as the vector matrix equation

Where

A=

, B =

All the partial derivatives in the matrices A and B defined above (called the Jacobian

matrices) are evaluated at the equilibrium state .

Lecturer: 3

: State space representation using physical variable

Introduction: In this representation the state variables are real physical variables, which can

be measured and used for manipulation or for control purposes. The approach generally

adopted is to break the block diagram of the transfer function into subsystems in such a way

that the physical variables can be identified.

How to solve a problem

1. For an electrical network, apply any of the Network analysis (such as KCL, KVL, Mesh

and Node).


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2. Write the equations such that all the elements of the network have been considered.

3. The equation will be having derivative terms for energy storage elements.

4. Try to convert each equation to single order.5. Finally, write the state space equation.

For a mechanical system

1. For Mechanical system, convert the physical Model into its equivalent electrical network

using Force voltage or Force current analogy.

2. Electrical network can be solved as above.

Example:1

Example: 2

Example: 3


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Solution: Example.1


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Solution: Example.2,3


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Lecturer: 4

: State space representation using Phase Variable or Observable canonical form and

Alternate Phase Variable or Controllable Canonical form.

: Phase variable formulations for transfer function with poles and zeros.

Introduction: It is often convenient to consider the output of the system as one of the state

variable and remaining state variable as derivatives of this state variables. The state variables

thus obtained from one of the system variables and its (n-1) derivatives, are known as n-

dimensional phase variables.

Steps to be followed:

1. Obtain the differential equation for the given Physical Model.

2. Obtain the transfer function from the given differential equation.

An example has been solved both in phase and alternate phase variable form.


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Lecturer: 5

: State space representation using Canonical Variables (Normal form or Jordan Normal Form

Representation).


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: Derivation of Transfer function for State Model

Lecturer: 6


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: Eigen values and Eigenvectors

Introduction : - Eigen values and eigen vector are special form of square matrix. While theeigen value parameterizes the diagonal properties of the system (time scale, resonance

properties, amplifying factor etc), the eigenvector define the vector coordinate of the normal

mode of the system. Each eigenvector is associated with particular eigen values. The general

state of the system can be expressed as a combination of eigenvector.

Eigen vector can be explained as a vector ‘ ’ such that matrix operator ‘A’ transforms it to a

vector ‘ ’ ( is a constant) i.e. to a vector having the same direction in state space as the

vector .

Such a vector X is a solution of the equation

Or or

Now, the above equation has a non-trivial solution of and only if

or


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Examples has been solved for different cases :


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: Time domain solution of State Equation

: Properties of the State Transition Matrix


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Lecturer: 8

: Computation of State transition matrix


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Lecturer: 9

: Computation by Techniques based on the Cayley – Hamilton Theorem


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Lecturer: 10

: Computation by Techniques based on the Sylvester’s Expansion theorem


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Lecturer: 11

: Concept of Controllability and Observability

CONCEPTS OF CONTROLLABILITY AND OBSERVABILITY

Controllability

A system is said to be controllable if it is possible to transfer the system state from any initial

state x( ) to any desired state x(t) in specified finite time by a control vector u(t).

A system is completely controllable if all the state variables are controllable.

Controllability criterion

Controllability is a property of the linkage between the input and the state & thus involves the

matrices A and B.

For an order linear time invariant system, controllable can be determined from the matrix

shown below

=[B AB B ……… B ] is of rank n.

This test for controllability is called Kalman’s test.

Observability

A system is said to be observable if each state x(t) could be determined from measurement of

the output over a finite interval of time 0 ≤ t ≤ .

Observability criterion

Observability is a property of linkage between the output & the state & thus involves the

matrices A and C.

For an order linear time invariant system is completely observable if and only if the

observability matrix

= [ ] is of rank n.

This is Kalman’s test for observability.

Solved Problems

1) A dynamic system S represented by the state equation

ẋ= + r


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Check whether the system is completely controllable.

Ans:- A = , B =

AB = = , B = =

The controllability matrix is

=[B AB B ] = and the determinant of ≠ 0 and the

rank of matrix is 3. Hence the system is completely controllable.

2) The state equation for a system is

ẋ= + r

Check whether the system is completely controllable.

Ans:- A = , B =

AB = =

The controllability matrix

=[B AB ] = and the determinant of = 0 and the rank of

matrix is 1. Hence the system is not completely controllable.

3) A linear system is described by a state model

ẋ= + r and y = x

Determine whether the system is completely observable.

Ans:- A = , B = , =

= =

The observability matrix

= [ ] = and determinant of ≠ 0 and the rank of

matrix is 2. Hence the system is completely observable.


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Lecturer: 12

: Effect of Pole-zero Cancellation in Transfer Function

POLE-ZERO CANCELLATIONS AND STABILITY

Consider the linear time-invariant system given by the transfer function.

Recall that this system is stable if all of the poles are in the OLHP, and these poles are the

roots of the polynomial D(s). It is important to note that one should not cancel any common

poles and zeros of the transfer function before checking the roots of D (s). Specifically,

suppose that both of the polynomials N(s) and D (s) have a roots =a f or some complex (or

real)number a. One must not cancel out this common zero and pole in the transfer function

before testing for stability. The reason for this is that, even though the pole will not show up

in the response to the input, it will still appear as a result of any initial conditions in the

system, or due to additional inputs entering the system (such as disturbances). If the pole and

zero are in the CRHP, the system response might blow up due to these

Initial conditions or disturbances, even though the input to the system Is bounded, and this

would violate BIBO stability

To see this a little more clearly, consider the following example. Suppose the transfer

function of a linear system is given by

Nothing that s 2+2s-3=(s+3)(s-1) suppose we decided to cancel out the common pole and zero

at s=0 to obtain


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Based on this transfer function we might conclude that the system is stable, since it only has a

pole in OLHP . what we should actually dois look at the original denominator D(s) , and

correctly conclude that the system is un stable because one of the pole in the CRHP. To seewhy the pole zero cancellation hide the stability of the system first writes out the differential

equation corresponding to the transfer function we obtain

Taking Laplace transformation on both side and taking initial condition in count

Rearranging the equation we get

Note that the denominator polynomial in each of the terms on the right hand sides is equal to

D(s) (the denominator of the transfer function). For simplicity, suppose that y(0)=y 0 (for

some real number y 0) y˙(0) = 0 and u(0) = 0.The partial fraction expansion of the term

is given by

and this contributes the term to the response of the system. Note

that the e t term blows up, and thus the output of the system blows up if y 0 is not zero, even if

the input to the system is bounded.


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A nonlinear system, when excited by a sinusoidal input, may generate several harmonics

in addition to the fundamental corresponding to the input frequency. The amplitude of the

fundamental is usually the largest, but the harmonics may be of significant amplitude in manysituations.

Another peculiar characteristic exhibited by nonlinear systems is called jump

phenomenon. For example, let us consider the frequency response curve of spring-mass

damper system. The frequency responses of the system with a linear spring, hard spring and

soft spring are as shown in Fig. 6.2(a), Fig. 6.2(b) and Fig. 6.2(c) respectively. For a hard

spring, as the input frequency is gradually increased from zero, the measured responsefollows the curve through the A, B and C, but at C an increment in frequency results in

discontinuous jump down to the point D, after which with further increase in frequency, the

response curve follows through DE. If the frequency is now decreased, the response follows

the curve EDF with a jump up to B from the point F and then the response curve moves

towards A. This phenomenon which is peculiar to nonlinear systems is known as jump

resonance. For a soft spring, jump phenomenon will happen as shown in fig. 6.2(c).

Fig. 6.2(a) Fig. 6.2(b) Fig. 6.2 (c)

When excited by a sinusoidal input of constant frequency and the amplitude is increased from

low values, the output frequency at some point exactly matches with the input frequency and

continue to remain as such thereafter. This phenomenon which results in a synchronization or

matching of the output frequency with the input frequency is called frequency entrainment

or synchronization.


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Classification of Nonlinearities:

The nonlinearities are classified into i) Inherent nonlinearities and ii) Intentional

nonlinearities.The nonlinearities which are present in the components used in system due to the inherent

imperfections or properties of the system are known as inherent nonlinearities. Examples are

saturation in magnetic circuits, dead zone, back lash in gears etc. However in some cases

introduction of nonlinearity may improve the performance of the system, make the system

more economical consuming less space and more reliable than the linear system designed to

achieve the same objective. Such nonlinearities introduced intentionally to improve the

system performance are known as intentional nonlinearities. Examples are different types ofrelays which are very frequently used to perform various tasks. But it should be noted that the

improvement in system performance due to nonlinearity is possible only under specific

operating conditions. For other conditions, generally nonlinearity degrades the performance

of the system.


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Lecture-2

Saturation, Friction, Backlash, Relay, Multivariable Nonlinearity.

Common Physical Non Linearities:The common examples of physical nonlinearities are saturation, dead zone, coulomb friction,

stiction, backlash, different types of springs, different types of relays etc.

Saturation: This is the most common of all nonlinearities. All practical systems, when driven

by sufficiently large signals, exhibit the phenomenon of saturation due to limitations of

physical capabilities of their components. Saturation is a common phenomenon in magnetic

circuits and amplifiers.

Dead zone: Some systems do not respond to very small input signals. For a particular rangeof input, the output is zero. This is called dead zone existing in a system. The input-output

curve is shown in figure.

Backlash: Another important nonlinearity commonly occurring in physical systems ishysteresis in mechanical transmission such as gear trains and linkages. This nonlinearity is

somewhat different from magnetic hysteresis and is commonly referred to as backlash. In

servo systems, the gear backlash may cause sustained oscillations or chattering phenomenon

and the system may even turn unstable for large backlash.


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Relay: A relay is a nonlinear power amplifier which can provide large power

amplification inexpensively and is therefore deliberately introduced in control systems. A

relay controlled system can be switched abruptly between several discrete states which are

usually off, full forward and full reverse. Relay controlled systems find wide applications in

the control field. The characteristic of an ideal relay is as shown in figure. In practice a relay

has a definite amount of dead zone as shown. This dead zone is caused by the facts that relay

coil requires a finite amount of current to actuate the relay. Further, since a larger coil currentis needed to close the relay than the current at which the relay drops out, the characteristic

always exhibits hysteresis.

Multivariable Nonlinearity: Some nonlinearities such as the torque-speed characteristics of a

servomotor, transistor characteristics etc., are functions of more than one variable. Such

nonlinearities are called multivariable nonlinearities.


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Lecture-3,4

Basic Concepts, Singular Points: Nodal Point, Saddle Point, Focus Point, Centre or Vortex

Point

The Phase Plane Method:

This method is applicable to second order linear or nonlinear systems for the study of the

nature of phase trajectories near the equilibrium points. The system behaviour is qualitatively

analysed along with design of system parameters so as to get the desired response from the

system. The periodic oscillations in nonlinear systems called limit cycle can be identified

with this method which helps in investigating the stability of the system.Singular points : From the fundamental theorem of uniqueness of solutions of the state

equations or differential equations, it can be seen that the solution of the state equation

starting from an initial state in the state space is unique. This will be true if f 1(x1, x 2) and

f 2(x1,x2) are analytic. For such a system, consider the points in the state space at which the

derivatives of all the state variables are zero. These points are called singular points . These

are in fact equilibrium points of the system. If the system is placed at such a point, it will

continue to lie there if left undisturbed. A family of phase trajectories starting from different

initial states is called a phase portrait. As time t increases, the phase portrait graphically

shows how the system moves in the entire state plane from the initial states in the different

regions. Since the solutions from each of the initial conditions are unique, the phase

trajectories do not cross one another. If the system has nonlinear elements which are piece-

wise linear, the complete state space can be divided into different regions and phase plane

trajectories constructed for each of the regions separately.

Nodal Point: Consider eigen values are real, distinct and negative as shown in figure (a). For

this case the equation of the phase trajectory follows as z 2=c (z 1)k

1

Where k1= ( ) 0 so that the trajectories become a set of parabola as shown in figure

(b) and the equilibrium point is called a node. In the original system of coordinates, these

trajectories appear to be skewed as shown in figure (c).


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If the eigen values are both positive, the nature of the trajectories does not change, except that

the trajectories diverge out from the equilibrium point as both z 1(t) and z 2(t) are increasing

exponentially. The phase trajectories in the x1-x2 plane are as shown in figure (d). This typeof singularity is identified as a node, but it is an unstable node as the trajectories diverge from

the equilibrium point.

Where z 1(t)= z 1(0) z 2(t)= z 2(0)

Saddle Point:

Consider now a system with eigen values are real, distinct one positive and one negative.

Here, one of the states corresponding to the negative eigen value converges and the one

corresponding to positive eigen value diverges so that the trajectories are given by z 2 = c(z 1)-k

or (z 1)

k z2 = c which is an equation to a rectangular hyperbola for positive values of k. The

location of the eigen values, the phase portrait in z 1 – z 2 plane and in the x 1 – x 2 plane are as

shown in figure. The equilibrium point around which the trajectories are of this type is called

a saddle point


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Centre or Vortex Point:

Consider now the case of complex conjugate eigen values with zero real parts.

ie., λ 1, λ 2 = j ω

= = from which + d = 0

Integrating the above equation, we get R 2

which is an equation to a circle ofradius R. The radius R can be evaluated from the initial conditions. The trajectories are thus

concentric circles in y 1-y2 plane and ellipses in the x 1-x2 plane as shown in figure. Such a

singular points, around which the state trajectories are concentric circles or ellipses, are called

a centre or vortex .


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Lecture-5Stability of Non Linear Systems: Limit Cycles

Limit Cycles:

Limit cycles have a distinct geometric configuration in the phase plane portrait, namely, that

of an isolated closed path in the phase plane. A given system may have more than one limit

cycle. A limit cycle represents a steady state oscillation, to which or from which all

trajectories nearby will converge or diverge. In a nonlinear system, limit cycles describes the

amplitude and period of a self sustained oscillation. It should be pointed out that not allclosed curves in the phase plane are limit cycles. A phase-plane portrait of a conservative

system, in which there is no damping to dissipate energy, is a continuous family of closed

curves. Closed curves of this kind are not limit cycles because none of these curves are

isolated from one another. Such trajectories always occur as a continuous family, so that there

are closed curves in any neighborhoods of any particular closed curve. On the other hand,

limit cycles are periodic motions exhibited only by nonlinear non conservative systems.

As an example, let us consider the well known Vander Pol’s differential equation

This describes physical situations in many nonlinear systems.

In terms of the state variables and we obtained

= x 2

= - x 1

The figure shows the phase trajectories of the system for μ > 0 and μ < 0. In case of μ > 0 we

observe that for large values of x1(0), the system response is damped and the amplitude of

x1(t) decreases till the system state enters the limit cycle as shown by the outer trajectory. On

the other hand, if initially x1(0) is small, the damping is negative, and hence the amplitude of


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x1(t) increases till the system state enters the limit cycle as shown by the inner trajectory.

When μ < 0, the trajectories moves in the opposite directions as shown in figure.

A limit cycle is called stable if trajectories near the limit cycle, originating from outside orinside, converge to that limit cycle. In this case, the system exhibits a sustained oscillation

with constant amplitude. This is shown in figure (i). The inside of the limit cycle is an

unstable region in the sense that trajectories diverge to the limit cycle, and the outside is a

stable region in the sense that trajectories converge to the limit cycle.

A limit cycle is called an unstable one if trajectories near it diverge from this limit cycle. In

this case, an unstable region surrounds a stable region. If a trajectory starts within the stable

region, it converges to a singular point within the limit cycle. If a trajectory starts in the

unstable region, it diverges with time to infinity as shown in figure (ii). The inside of an

unstable limit cycle is the stable region, and the outside the unstable region.

In the phase plane, a limit cycle is defied as an isolated closed curve . The trajectory has to be

both closed, indicating the periodic nature of the motion, and isolated, indicating the limiting

nature of the cycle (with near by trajectories

converging or diverging from it).


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Depending on the motion patterns of the trajectories in the vicinity of the limit cycle, we can

distinguish three kinds of limit cycles

• Stable Limit Cycles : all trajectories in the vicinity of thelimit cycle converge to it as t → ∞ (Fig. 2.10.a).

• Unstable Limit Cycles : all trajectories in the vicinity of

the limit cycle diverge to it as t → ∞ (Fig. 2.10.b)

• Semi-Stable Limit Cycles : some of the trajectories in

the vicinity of the limit cycle converge to it as t → ∞ (Fig. 2.10.c)


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Lecture-6

Construction by Analytical Method, Construction by Graphical Methods

Construction of Phase Trajectories:

Consider the homogenous second order system with differential equations

This equation may be written in the standard form

+2 + x=0

where ζ and ωn are the damping factor and undamped natural frequency of the

system.Defining the state variables as x = x 1 and = x 2, we get the state equation in the state

variable form as

= x 2

=

These equations may then be solved for phase variables x 1 and x 2. The time response plots of

x1, x2 for various values of damping with initial conditions can be plotted. When the

differential equations describing the dynamics of the system are nonlinear, it is in general not

possible to obtain a closed form solution of x 1, x 2. For example, if the spring force is

nonlinear say (k 1 x + k 2 x3) the state equation takes the form

= x 2

=

Solving these equations by integration is no more an easy task. In such situations, a

graphical method known as the phase-plane method is found to be very helpful. The

coordinate plane with axes that correspond to the dependent variable x 1 and x 2 is called


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phase-plane. The curve described by the state point (x 1,x2) in the phase-plane with respect to

time is called a phase trajectory. A phase trajectory can be easily constructed by graphical

techniques.Isoclines Method:

Let the state equations for a nonlinear system be in the form

=f 1(x1,x2)

= f 2(x1,x2)

When both f 1(x1,x2) and f 2(x1,x2) are analytic.

From the above equation, the slope of the trajectory is given by

Therefore, the locus of constant slope of the trajectory is given by

f 2(x1,x2) =M f 1(x1,x2)

The above equation gives the equation to the family of isoclines. For different values of M,

the slope of the trajectory, different isoclines can be drawn in the phase plane. Knowing the

value of M on a given isoclines, it is easy to draw line segments on each of these isoclines.

Consider a simple linear system with state equations

=

Dividing the above equations we get the slope of the state trajectory in the x 1-x2 plane as

For a constant value of this slope say M, we get a set of equations

which is a straight line in the x 1-x2 plane. We can draw different lines in the x 1-x2 plane for

different values of M; called isoclines. If draw sufficiently large number of isoclines to cover

the complete state space as shown, we can see how the state trajectories are moving in the

state plane. Different trajectories can be drawn from different initial conditions. A large

number of such trajectories together form a phase portrait. A few typical trajectories are

shown in figure given below.


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The Procedure for construction of the phase trajectories can be summarised as below:

1.For the given nonlinear differential equation, define the state variables as x 1 and x 2 and

obtain the state equations as

= x 2f

2.Determine the equation to the isoclines as

3. For typical values of M, draw a large number of isoclines in x 1-x2 plane

4. On each of the isoclines, draw small line segments with a slope M.

5. From an initial condition point, draw a trajectory following the line segments

With slopes M on each of the isoclines.

Delta Method:

The delta method of constructing phase trajectories is applied to systems of the form

Where may be linear or nonlinear and may even be time varying but must be

continuous and single valued.


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With the help of this method, phase trajectory for any system with step or ramp or any time

varying input can be conveniently drawn. The method results in considerable time saving

when a single or a few phase trajectories are required rather than a complete phase portrait.While applying the delta method, the above equation is first converted to the form

In general depends upon the variables x, and t, but for short intervals the changes

in these variables are negligible. Thus over a short interval, we have

, where δ is a constant.

Let us choose the state variables as x 1 = x; ,then

Therefore, the slope equation over a short interval is given by

With δ known at any point P on the trajectory and assumed constant for a short interval, we

can draw a short segment of the trajectory by using the trajectory slope dx 2/dx 1 given in the

above equation. A simple geometrical construction given below can be used for this purpose.

1. From the initial point, calculate the value of δ.

2. Draw a short arc segment through the initial point with (- δ, 0) as centre, thereby

determining a new point on the trajectory.

3. Repeat the process at the new point and continue.

Example : For the system described by the equation given below, construct thetrajectory starting at the initial point (1, 0) using delta method.

Let , and ,then

The above equation can be rearranged as


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Where

At initial point δ is calculated as δ = 0+1-1 = 0. Therefore, the initial arc is centered at point

(0, 0). The mean value of the coordinates of the two ends of the arc is used to calculate the

next value of δ and the procedure is continued. By constructing the small arcs in this way the

complete trajectory will be obtained as shown in figure.


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Lecture-7,8

The Describing Function Method, Derivation of Describing Functions

Describing Function Describing functions provide a method for the analysis of nonlinear systems that is

closely related to the linear-system techniques involving Bode or gain-phase plots. It is

possible to use this type of analysis to determine if limit cycles (constant-amplitude periodic

oscillations) are possible for a given system. It is also possible to use describing functions to

predict the response of certain nonlinear systems to purely sinusoidal excitation, although this

topic is not covered here.2

unfortunately, since the frequency response and transient response

of nonlinear systems are not directly related, the determination of transient response is not

possible via describing functions.

Certain approximate techniques exist which are capable of determining the behavior of

wider class of systems then is possible by phase plane method. These techniques are known

as the describing function techniques .Let us consider the block diagram of a nonlinear

system shown in figure, where in the blocks G(S) represent the linear elements, while the

block N represents the nonlinear elements.

Let us assume that x to be non linearity is sinusoidal,

x = X

With such an input, the output of nonlinear element will in general be non sinusoidal periodic

function which may be expressed in terms of Fourier series as follows:

y =

In the absence of external input (r=0), the output y of N is feedback to its input through the

linear element G(S) tandem. It can be assumed to a good degree of accuracy that all the

harmonics of y are filtered out in the process such that input x(t) to the non linear element N


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is mainly contributed by the fundamental component of y ,i.e.(t) remains sinusoidal. Under

such conditions the harmonic content of y can be thrown away for the purpose of analysis and

the fundamental component of y, i.e.,

The non linearity can be replaced by a described function which is defined to be the

complex function embodying amplification and phase shift of the fundamental frequency

component of y relative to x, i.e.

Derivation of Describing Functions

The describing function of a non linear element is given by

Where X =amplitude of the input sinusoid; Y1=amplitude of fundamental harmonic

component of the output; and =phase shift of fundamental harmonic component of the

output with respect to input.

Therefore, for computing the describing function of a nonlinear element, we are simply

required to find the fundamental harmonic component of its output for an input x= X

.The fundamental component of the output can be written as

Where the coefficients A 1 and B 2 of the Fourier series are

B1= ………… (1)


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A1= …………. (2)

The amplitude and phase angle of the fundamental component of the output are given by

Y1=

Illustrative derivation of describing functions of commonly encountered non linearities are

given below

Dead-zone and Saturation:Idealized characteristics of a nonlinearity having dead-zone and saturation and its response to

sinusoidal input are shown in figure. The output wave form may be described as follows:

Where and

Using equations 1 and 2 recognizing that the output has half-wave and quarter-wave

symmetries, we have


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It is found that the describing functions of saturation and dead-zone nonlinearity are

frequency-invariant having zero phase shifts. In fact all nonlinearities, whose input output

characteristics are represented by a planner graph, would result in describing functionsindependent of frequency and amplitude dependent.

Lecture-9

Relay with Dead-zone and Hysteresis, Backlash

Relay with Dead-zone and Hysteresis:


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=

=

=

Therefore

……………….. (6)

It is seen that is a function of X, the input amplitude and is independent of frequency.

Further, it being memory type nonlinearity, has both magnitude and angle .Equation (6)

is the general expression for the describing function of a relay element, from which the

describing functions of various simplified relay characteristics given below;

1. Ideal relay,(fig a) letting D=H=0 in eqn.(6)

…………. (7)

2. Relay with dead-zone (fig b).letting H=0 in eqn. (6)


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; X>D/2

3. Relay with hysteresis (fig c).Letting H=D in eqn. (6)

; X>H/2

Backlash

The characteristics of backlash nonlinearity and its response to sinusoidal input are shown in

fig. The output may be described as follows:

y =x - b/2 ;

=X - b/2 ;

=x + b/2 ;

=-X + b/2 ;

=x - b/2 ;

Where


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B1=

= + +

=

A1=

= + +


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Therefore,


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Lecture-10

Stability Analysis by Describing Function Method, Jump Resonance

Describing functions are most frequently used to determine if limit cycles (stable-

amplitude periodic oscillations) are possible for a given system, and to determine the

amplitudes of various signals when these oscillations are present.

Describing-function analysis is simplified if the system can be arranged in a form

similar to that shown in Fig. 6.9. The inverting block is included to represent the inversionconventionally indicated at the summing point in a negative-feedback system. Since the

intent of the analysis is to examine the possibility of steady-state oscillations, system input

and output points are irrevelant. The important feature of the topology shown in Fig. 6.9 is

that a single nonlinear element appears in a loop with a single linear element. The linear

element shown can of course represent the reduction of a complex interconnection of linear

elements in the original system to a single transfer function.

The system shown in Fig. 6.10 illustrates a type of manipulation that simplifies the use of

describing functions in certain cases. A limiter consisting of back-to-back Zener diodes is

included in a circuit that also consisting of back-to-back Zener diodes is included in a circuit


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that also contains an amplifier and a resistor-capacitor network. The Zener limiter is assumed

to have the piecewise-linear characteristics shown in Fig. 6.10b.

The describing function for the nonlinear network that includes R1, R2, C, and the limiter

could be calculated by assuming a sinusoidal signal for v b and finding the amplitude and

relative phase angle of the fundamental component of VA. The resulting describing function

would be frequency.


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Once a system has been reduced to the form shown in Fig. 6.9, it can be analyzed by means

of describing functions. The describing-function approximation states that oscillations may

be possible if particular values of E i and exist such that

Gives one stable and one unstable limit cycle. The left most intersection corresponds to the

stable one.


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Jump Resonance

Consider a nonlinear system shown in figure where in the nonlinear part has been replaced byits describing function.

Let r (t) = R

X (t) = R

It immediately follows that

=

We can also write

=

Further we can write

In order to simplify the analysis we assume

(i)

(ii) i.e independent of frequency

So we can write


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L ecture-11Basic Stability Theores, Liapunov Functions, Instability

Consider a dynamical system which satisfies

x˙ = f ( x, t ) x(t 0) = x0 x R n. (4.31)

We will assume that f ( x, t ) satisfies the standard conditions for the existence and uniqueness

of solutions. Such conditions are, for instance, that f ( x, t ) is Lipschitz continuous with respect

to x, uniformly in t , and piecewise continuous in t . A point x* Rn is an equilibrium point of

(4.31) if

F ( x* , t ) ≡ 0. Intuitively and somewhat crudely speaking, we say an equilibrium point is

locally stable if all solutions which start near x* (meaning that the initial conditions are in a

neighborhood of x ) remain near x for all time. The equilibrium point x is said to be locally

asymptotically


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stable if x is locally stable and, furthermore, all solutions starting near x tend towards x as

t → ∞ . We say somewhat crude because the time-varying nature of equation (4.31)

introduces all kinds of additional subtleties. Nonetheless, it is intuitive that a pendulum has a

locally stable

equilibrium point when the pendulum is hanging straight down and an unstable equilibrium

point when it is pointing straight up. If the pendulum

is damped, the stable equilibrium point is locally asymptotically stable.

By shifting the origin of the system, we may assume that the equilibrium

point of interest occurs at x = 0. If multiple equilibrium points exist, we will need to study

the stability of each by appropriately shifting the origin.

4.1 Stability in the sense of Lyapunov

The equilibrium point x = 0 of (4.31) is stable (in the sense of Lyapunov)

at t = t 0 if for any > 0 there exists a δ (t 0 , ) > 0 such that

< δ < , t ≥ t 0………….(4.32)


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Lyapunov stability is a very mild requirement on equilibrium points. In particular, it does not

require that trajectories starting close to the origin tend to the origin asymptotically. Also,stability is defined at a time instant t 0. Uniform stability is a concept which guarantees that

the equilibrium point is not losing stability. We insist that for a uniformly stable equilibrium

point x , δ in the Definition 4.1 not be a function of t 0, so that equation (4.32) may hold for

all t 0. Asymptotic stability is made precise in the following definition:

4.2 Asymptotic stability

An equilibrium point x* = 0 of (4.31) is asymptotically stable at t = t 0 if

1. x * = 0 is stable, and

2. x * = 0 is locally attractive; i.e., there exists δ (t 0) such that

< δ = 0 (4.33)

As in the previous definition, asymptotic stability is defined at t 0.Uniform asymptotic stability

requires:

1. x * = 0 is uniformly stable, and

2. x * = 0 is uniformly locally attractive; i.e., there exists δ independent

of t 0 for which equation (4.33) holds. Further, it is required that the convergence in equation

(4.33) is uniform.


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Finally, we say that an equilibrium point is unstable if it is not stable. This is less of a

tautology than it sounds and the reader should be sure he or she can negate the definition of

stability in the sense of Lyapunov to get a definition of instability. In robotics, we are almostalways interested in uniformly asymptotically stable equilibria. If we wish to move the robot

to a point, we would like to actually converge to that point, not merely remain nearby. Figure

4.7 illustrates the difference between stability in the sense of Lyapunov and asymptotic

stability. Definitions 4.1 and 4.2 are local definitions; they describe the behavior of a system

near an equilibrium point. We say an equilibrium point x* is globally stable if it is stable for

all initial conditions x0 R n. Global stability is very desirable, but in many applications it can

be difficult to achieve. We will concentrate on local stability theorems and indicate where it is

possible to extend the results to the global case. Notions


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Basic theorem of Lyapunov

Let V ( x, t ) be a non-negative function with derivative ˙V along the trajectories of the system.

1. If V ( x, t ) is locally positive definite and ˙V ( x, t ) ≤ 0 locally in x and for all t, then the

origin of the system is locally stable (in the sense of Lyapunov).

2. If V ( x, t ) is locally positive definite and decrescent, and ˙V ( x, t ) ≤ 0 locally in x and for allt, then the origin of the system is uniformly locally stable (in the sense of Lyapunov).

3. If V ( x, t ) is locally positive definite and decrescent, and − ˙V ( x, t ) is locally positive

definite, then the origin of the system is uniformly locally asymptotically stable.

4. If V ( x, t ) is positive definite and decrescent, and − ˙V ( x, t ) is positive definite, then the

origin of the system is globally uniformly asymptotically stable.


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The conditions in the theorem are summarized in Table 4.1.

Theorm-1

Consider the system

Suppose there exists a scalar function v(x) which for some real number satisfies the

following properties for all x in the region

(a) V(x)>0; x that is v(x) is positive definite scalar function.

(b) V (0) = 0

(c)V(x) has continuous partial derivatives with respect to all component of x

(d) (i.e dv/dt is negative semi definite scalar function)

Then the system is stable at the origin.

Theorem-2

If the property of (d) of theorem-1 is replaced with (d) , (i.e dv/dt is negative

definite scalar function),then the system is asymptotically stable.

It is intuitively obvious since continuous v function>0 except at x=0, satisfies the condition

dv/dt


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The linear system is asymptotically stable in-the-large at the origin if and only if

given any symmetric, positive definite matrix Q, there exists a symmetric positive definite

matrix P which is the unique solutionATP + PA = -Q ……….13.10

Proof

To prove the sufficiency of the result of above theorem, let us assume that a symmetric

positive definite matrix P exists which is the unique solution of eqn.(13.11). Consider the

scalar function.

V(x) = xT Px

Note that V( x) > 0 for x 0V ( 0) = 0

And

The time derivate of V(x) is

(x) = T P X + x T P

Using eqns. (13.9) and (13.10) we get

V(x) = xTATPx +x TPAx

= xT

(AT

P + PA)x = -xTQx

Since Q is positive definite, V(x) is negative definite. Norm of x may be defined as

Then

V(X) =

V(X)

The system is therefore asymptotically stable in-the large at the origin.

In order to show that the result is also necessary, suppose that the system is asymptotically

stable and P is negative definite, consider the scalar function

Therefore V( x) = xT Px……………………………….13.11

(x) =

=


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Page 113

>0

There is contradiction since V(x) given by eqn. (13.11) satisfies instability theorem.

Thus the conditions for the positive definiteness of P are necessary and sufficient forasymptotic stability of the system of eqn. (13.9).

Methods of constructing Liapunov functions for Non linear Systems.

As has been said earlier ,the liapunov theorems give only sufficient conditions on system

stability and furthermore there is no unique way of constructing a liapunov function except in

the case of linear systems where a liapunov function can always be constructed and both

necessary and sufficient conditions Established .Because of this draw back a host of methods

have become available in literature and many refinements have been suggested to enlarge theregion in which the system is found to be stable. since this treatise is meant as a first exposure

of the student to the liapunov direct method, only two of the relatively simpler techniques of

constructing a liapunov’s function would be advanced here.

Krasovskiis method

Consider a system

Define a liapunov function as

V = f TPf

Where P=a symmetric positive definite matrix.

Now

………13.13

J = is jacobian matrix

Substituting in eqn (13.13), we have

=


114/114


( ) f

Let Q P + PJ

Since V is positive definite, for the system to be asymptotically stable, Q should be negative

definite. If in addition, the system is asymptotically stable in-the-

large.

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