JOURNAL OF TELECOMMUNICATIONS, VOLUME 17, ISSUE 1, NOVEMBER 2012 1

Analysis and Design of Narrow Band Filters for Telecommunications Applications

Adiba Elfadl 1, 2, Lekbir Belrhiti 1, Hamid Bouyghf 1 and Seddik Bri 1

Abstract—In this paper, we have developed the microwave theory and analyzed the filter design for telecom applications. The software ADS (Advanced Design System) is used for design bandpass filter with microstrip lines coupled (McLin) Chebychev type of order 5 at the center frequency f = 2.4 GHz with 10% bandwidth. The topology of a DBR filter (Dual Behavior Resonator) of order 2 of a chain 2.14 GHz is simulated. The design of a filter types that Tchebysheve 0.1dB ripple in the pass band with a standard bandwidth of 2% and a center frequency fo = 5.8 GHz using a ring resonator and the ring resonator side coupled to access quarter wavelength for a center frequency of 94 GHz.

Index Terms— Design Narrow, Band Filter, Microstrip Technology, ADS software, Telecommuincations.

—————————— u ——————————

1 INTRODUCTION Filters are essential components of existing telecommu-

nications systems. They are found in different parts of the front-end radio, to select the useful signal, on the one hand, but also to filter the harmonics generated by non-linear components (amplifiers, oscillators, mixers). The spectral congestion and the proximity of the allocated bands have led to increased demands in performance of these filters, including a level of increased selectivity and lower insertion loss constraints are prevailing [1],[2],[3]. An ideal filter is a system that transmits without any dis-tortion signal whose spectrum is useful in the range of pulses and totally eliminates any signal whose spectrum is outside the range. In practice the ideal filter synthesis is impossible; one can show that it leads to a noncausal realization. The filters will have made a different transmittance ideal. The filters are needed in telecommunication systems because they can separate the useful signal component of spurious components. The very important development of RF equipment imposes new constraints on microwave chan-nels. The filter function must meet new specifications, whether technical or financial: - How the RF spectrum with strong filtering constraints: center frequency, bandwidth, selectivity; - Consideration of specific parameters such as adaptation input / output; - For passive filters: taking into account losses (related to noise factor); - For active filters: taking into account the gain, consump-tion of non-linearity, noise; - Size reduction (gain of congestion) and cost, especially for mass production.

To change the structure of the low pass filter to structure a  bandpass centered, simply give the arm series and par-allel. This is obtained by placing in series, series resonant circuits and, in parallel, parallel resonant circuits. There are many topologies of band pass filters, which differ in frequency to meet specifications. Classically, there are three families of band pass filters, broadband filters and mid band and narrow band filters. We will present the classical topologies of band pass filter. The syntheses as-sociated permit, from a template filter, to define an ideal prototype filter characterized by sets of impedances and electrical lengths needed to achieve the desired function [4]. Our paper consists of two parts: Part or the general prin-ciple of filtering and as a filter in the microwave tele-communications system and the state of the art planar microwave filters are presented. Then we mention the criteria for selection of bandpass filter topologies such as broadband filters and mid-band and narrow band filters [5],[6],[7]. In the end of this section, we study the technol-ogy of planar filters (microstrip technology coplanar technology). In the second part, we used the software ADS (Advanced Design System) that allows to perform simulations using various libraries of active and passive microwave. Simulation results of two bandpass filters with coupled lines in parallel, also the design of a filter DBR (Dual Behavior Resonator) of order 2. Finally, we simulated a filter dual-mode ring and designing a filter ring W-band (94 GHz).

———————————————— 1.  Instrumentaion  and  Materials  group,  Electrical  Engineering  Department,  High  School  of  Technology:  ESTM,  Moulay  Ismail  University,  B.  P  3103,  

Meknès  –  Morocco.    2.  Systems  and  Telecommunications  Engineering  Decision  Laboratory,  Ibn  Tofail  University,  Faculty  of  Sciences,  B.P.  133,  Kenitra  -­‐‑  Morocco    

© 2012 JOT www.journaloftelecommunications.co.uk

2

2 THEORY OF THE MICROWAVE FILTER The broadband filters are characterized by bandwidths between 80% and 20%. Among the broadband filter to-pologies, we can distinguish the filters stubs and folded stub filters.The filter stubs (short circuit (CC) or open cir-cuit (OC)) are certainly the most classic. The synthesis of such structures proposed by G. Matthaei developed from a prototype low pass with quarter-wave inverters [8],[9]. The resonators are made from the quarter-wave stub short-circuited. This synthesis allows design elements of the filter whose schematic diagram is shown in fig.1.

 Fig. 1. Band-pass filter quarter wave stubs

The  synthesis  of  this  type  of  filter  is  proposed  by  G.  Matthaei  [10].  The  electrical  characteristics  of   the  filter  are:  center  fre-­‐‑quency  fo,  the  relative  bandwidth  w,  Za  the  impedance  level  of  standardization  and  gi  coefficients  of   the  Chebyshev  nor-­‐‑malized  low-­‐‑pass  filter:    

11 2 22. . 1C d ga

π ωθ⎧ ⎡ ⎤⎪ ⎢ ⎥⎣ ⎦⎨⎪⎩

= −

=   (1)  

Where d is a dimensionless parameter of freedom for ad-justing the level of impedance elements. It should keep the value of this parameter in a range between 0.1 and 1. From these parameters, we define the following interme-diate values:

2

1,2

, 1 0

12.. 2

1, 10

0 1

.

.

.

.

2 2. .tan( ), 1 1, 1 21... 1

ao

a

k k a

a k kk n

n n a n

a n

o

J CgY g

J g CY g g

J C ggY g g

J g Cak kNk k Yk n aθ

+

+= −

− +

−

⎧=⎪

⎪⎪⎪ =⎪⎪⎪⎨

=⎪⎪⎪⎪ ⎡ ⎤ ⎡ ⎤⎪ ⎢ ⎥ ⎢ ⎥⎪ ⎢ ⎥ ⎢ ⎥⎣ ⎦⎪ ⎣ ⎦⎩

+= ++ = −

                     (2)

 

It is then possible to calculate values of admittances characteristics of various lines making up the filter. The equations determining these values for stubs are:

1,21 0 1 1 1,2

1, , 11, , 12.. 1

11 0 1 1 1,

. .(1 ). .tan( )

.( . . . ).tan

a aa

k k k kk a k k k kk n

a a

nn a n n a n n

a

JY g Y d g Y N

Y

J JY Y N N

Y Y

JY Y g g d g g Y NY

θ

θ

− +− += −

−+ −

⎧ ⎡ ⎤= − + −⎪ ⎢ ⎥

⎣ ⎦⎪⎪ ⎡ ⎤⎪

= + − −⎨ ⎢ ⎥⎣ ⎦⎪

⎪ ⎡ ⎤⎪ = − + −⎢ ⎥⎪ ⎣ ⎦⎩

            (3)  

     

The use of stubs topology does not allow achieving nar-row band filters. Indeed, for a selective bandwidth, im-pedance levels quickly become very inevitably leading to feasibility problems. The narrow band filters are charac-terized by a bandwidth less than 20%. We present inter-esting topologies to perform the function in the narrow band microwave frequency range.

2.1  The Coupled Line Filter Quarter Wave Topology best known for the low band filter is based on the use of half-wave resonators and quarter wave coupled lines  (Figure 2). The filter order is equal to N-1 (where N is the number of coupled lines). For such filters the level of selectivity is closely linked to levels of coupling. The syntheses governing such filters have been developed by Matthaei or Cohn [10],[11]. The values of the different filter elements are based on the impedances of even and odd models for each of the coupled lines.  

 Fig. 2. Bandpass filter coupled line quarter wave

Let   f0,   w,   Za,   and   gi,   respectively, the center frequency, relative bandwidth, the impedance of the filter and the normalization coefficients of Chebyshev normalized low-pass prototype. For the input and output of the filter (k=0 and k=n):

  , 11

, 1, 1

, 1 , 1

, 1 , 1 , 1 , 1

2

, 11, 1 , 1

1.

( ) . . 1

( ) ( )

( ) ( ) ( ) ( )

tan( ) ( ) . .2

2

k kk k

a k kk koo a

a

a ak k k koe a oo

b b a ak k k k k k k koo oe oo oe

b a k kk k k koe oe a a

a

Jg g

Jh

Z

Jh

Z Z

Z Z ZZ Z Z Z

Z Z Z ZZθ

+

+

++

+ +

+ + + +

++ +

⎧ =⎪⎪⎪ ⎛ ⎞⎪ = +⎜ ⎟⎪ ⎝ ⎠⎪⎪

= −⎨⎪

= + −⎪⎪

⎡ ⎤⎪ ⎛ ⎞⎢ ⎥= + + −⎜ ⎟⎜ ⎟⎢ ⎥⎝ ⎠⎢ ⎥

⎩

⎪⎪⎪

(4)                                  

                       

   

3

   For the other sections of the filter (1≤k≤n-1) :

, 1

1

22

, 11, 1

, 1, 1 , 1

, 1, 1 , 1

1.

tan4

( ) .

( ) .

.

.

k k

k ka

k kk k

a

k kk koe k ka

a

k kk koo k ka

a

Jg g

J

JZ

JZ

Z

M Z

hZZ M

hZZ M

θ

+

+

+

+

++ +

++ +

⎧=⎪

⎪⎪

⎛ ⎞⎪= + ⎜ ⎟⎪ ⎜ ⎟⎪ ⎝ ⎠⎨

⎪ ⎛ ⎞= +⎪ ⎜ ⎟

⎪ ⎝ ⎠⎪

⎛ ⎞⎪ = −⎜ ⎟⎪ ⎝ ⎠⎩

                                                               (5)                  

We define a dimensionless parameter h freedom to adjust the impedance level of the different sections of coupled lines. This parameter has a theoretical value for setting levels close to impedance of 50 Ohms; this value is given by the following expression:

2

011

2tan

1

⎟⎟⎠

⎞⎜⎜⎝

⎛+

=

Za

Jh

θ

             (6)

   

2.2    The Ring Filter The pseudo-elliptic filter ring presented here was de-volloped by MK Mohd Salleh et al. [12],[13]. It consists of a resonator whose perimeter is equal to a wavelength excited by two quarter-wave lines coupled identical, thus facilitating the design phase and adjustment. Such a filter is characterized by the presence of two distinct modes of propagation that create a transmission zero of both sides of the bandwidth. Discrimination of the two modes of propagation is generally provided by the introduction of discontinuities in the ring. Here it is assured by control-ling the level of energy coupling in the ring. The synthesis of this topology has parameters of freedom to easily con-trol the electrical characteristics of the filter.

2.3    Dual Behavior Resonator Filter (DBR) The resonator DBR (Dual Behavior Resonator) is a topo-logy developed by C.Quendo LEST. This topology is composed of two paralleled stubs open circuit terminated having an electrical differ either in terms of electrical length is impedance. Both stubs have dual behavior, both bandpass and bandstop filters [14],[15],[16],[17]. The be-havior notch has two stubs that create two transmission zeros at two different frequencies. Bandpass behavior is provided by the constructive recombination occurring between the two zeros (Figure 3). The elementary resona-tor is characterized by a pole (bandwidth) and two transmissions zeros, these three parameters are totally independent in the general cases.

   

Fig. 3. Resonator topology DBR  

A filter of order N DBR is obtained by cascading N re-sonators. In the case of a filter of order N, the N pole this electrical response in the pass band, N zeros in the stop band and lower N zeros in the upper stopband. Figure 4 shows two examples of fourth-order filters. For both examples, the answers have the same bandwidth, howe-ver attenuated bands are different. In the first case all ze-ros are combined while in the second case the zeros are distinct. This feature gives the DBR filter flexibility during the design [18],[19],[20].  

   

Fig. 4. Topology if the DBR filter  

The combline topology consists of a network of parallel resonators coupled, loaded on the hand by a short circuit termination, and on the other hand by a capacitor. Access is made by coupling between the lines to short circuit termination (0, n +1) and the resonators (1, n). The use of capacitive load slightly reduces the length of the resona-tors, the resonators and has a length less than λg/4 to the resonant frequency. Combline filters have an asymmetric electrical response. Indeed, the transmission response of such a filter is sligh-tly more than low frequency high frequency. There is no yet comprehensive summary for this types of filter synthesis in essentially below shows the calculation of capacity. The synthesis of a filter Combline is as follows: Let θo the electrical length of the resonator center frequency, w is the filter bandwidth, ε the permittivity of the propagation medium absolute and relative permittivity εr, determine the normalized admittances characteristics (Yaj /Ya) for obtaining a coefficient of vacuum optimal quality of the resonator. Like h, Yaj parameters appear as one degree of freedom for this synthesis. Once this setting is chosen, it is sufficient to determine:  

4

20 0 0

1..

1

1

0 1

1

1

, 1

12.. 1

cot .csc.2

.

.

..

.

j aj

a aj n

aT

a

Tn a

a n n

j j

j j A A

a k kj n

b YY Y

bwYG

Y g gbnw

G YY g g

b bJ Y YwY g g

θ θ θ

=

+

+

+

+= −

⎧ +=⎪

⎪⎪⎪⎪

=⎪⎪⎪⎨⎪⎪ =⎪⎪⎪⎪

=⎪⎪⎩

         (7)  

                                                                                                               The capacitance per unit of standard lengths of the reso-

nator is then for transformers input and output:

0 1376.7. . 1

376.7.1 . 1

a T

ar

a Tn

ar

C Y GY

Y GCnY

ε ε

ε ε

⎧ ⎛ ⎞⎪ = −⎜ ⎟⎪ ⎝ ⎠⎪⎨⎪ ⎛ ⎞+

= −⎜ ⎟⎪⎝ ⎠⎪⎩                                                                                (8)

 

For resonators 1 and n:

1 01 1 120

10

376.7.. 1 .tan( )

376.7. 1,. 1 .tan( )

a a T

a a ar

a Tn n

a a ar

Y Y CC G JY Y Y

Y G CCn Yan Jn nY Y Y

θε εε

θε εε

+

⎧ ⎡ ⎤= − + − +⎪ ⎢ ⎥

⎣ ⎦⎪⎨

⎡ ⎤−⎪ = − + − +⎢ ⎥⎪⎣ ⎦⎩

    (9)

 

     

 The values mutual capacitances per unit length are:

01 0376.7.

376.7.

376.7.. .tan

, 1 1

, 1 , 10

1, 1

C CY

C Y C

C JYY

ar

n n a nr

j j j jaarj n

ε εε

ε εε

θε ε

⎧⎪ = −⎪⎪⎪⎪

= −⎨⎪⎪

⎛ ⎞⎪ = ⎜ ⎟⎪ ⎜ ⎟⎝ ⎠⎪⎩

+ +

+ +

= −

                       (10)                                                                                                          

                 Values capacitive loads are:

0

0,1

cotωθ

⎟⎟⎠

⎞⎜⎜⎝

⎛=

=a

ajanj

sj Y

YYC

                               (11)

     

 The interdigitated topology is composed of a network of coupled liner quarter-wave alternating having end open circuit, short circuit. In the same way as for comblines filters, access is achieved by coupling between the lines terminated open circuit (0, n +1) and resonators (1, n). The electrical response of the interdigital filter is perfectly symmetrical and allows for better electrical characteristics in terms of phase. However, the synthesis of this type of filter is complex which creates difficulties in controlling the filter parameters.

2.4 Technologies of Planar Microstrip Filters In application where the signal carrying high power is not an essential parameter, the use of planar technologies is a solution to address the problems of congestion and vol-

ume weight structures. Among the planar technologies, we can distinguish technologies microstrip, coplanar, and multilayer membrane, each with its own specificities as a geometrical point of view than electricity. Used to make very microwave circuits, the microstrip structure comprises a conductive metal deposited on the upper face of a dielectric substrate and a ground plane on the underside. The fundamental mode of propagation of such a propagation medium is not TEM (transverse elec-tromagnetic) because the cross section is not homogene-ous. However, since the amplitudes of the longitudinal components of electric and magnetic fields are low enough to be neglected, it is called a quasi-TEM. This then makes it possible the modeling of the structure in the form of a transmission line characteristic impedance Zc in an equivalent homogeneous medium characterized by a relative effective permittivity εeff. From an industrial point of view, the technological pro-cess for etching of microstrip circuits is relatively simple to implement (similar to the industry 'printed circuit' in lower frequency). However, if the postponement of com-ponents in series is simple, its parallel implementation is more complex given the presence of ground plane on the underside. Indeed, the realization of a short-circuit ground return calls for sophisticated drilling techniques, as well as methods of metallization can be complex. In addition to these difficulties in implementation, the influ-ence of plated through holes on the electrical performance of the circuit is not negligible because of parasitic effects they show. The advantage of the coplanar technology is can connect an element to ground without going through in the sub-strate, which limits the parasitic effects. Another ad-vantage of this technology is the ease of integration and deferral of other structures, such as MMIC, with delay flip-chips. The presence of two modes is one of the main disadvantages of such technology. Although the com-bined use of both modes is not negligible, it is usually the odd mode (quasi-TEM) is used because low dispersion. To filter the even mode (TE quasi) it is necessary to force the potential between the two ground planes to the same value by the use of air bridges

3 SIMULATIONS RESULTS

3.1 The Simulation Software ADS (Agilent Device System)

Advanced Design System, developed by Agilent EEsof EDA is software for modeling and design of electronic systems for microwave and radio frequencies. The appli-cations are vast and include another in the field of mobile phones, wireless networks, communications systems, radar and satellite. The software provides opportunities for design and simu-lation for the areas of RF and microwave and is divided into two modules Analog RF and Digital Signal Proces-sing Designer Designer that can interact with each other: - The design of monolithic integrated circuits (MMICs) and hybrid (with Surface Mounted Components).

5

- The design of new architectures for future telecommuni-cations standards.

3.2 Design A Band Pass Filter Coupled To Mi-crostrip Lines (MCLIN)

The specifications of the band pass filter Chebychev type of order 5 to design the center frequency f=2.4 GHz are summarized in the table below (table 1):

TABLE 1 CHARACTERISTICS OF THE BAND PASSFILTER AT 2.4 GHZ

 Type  of  Filter   Chebyshev  

Center  frequency   2.4  GHz  Bandwidth  at  3dB   10%  Maximum  ripple   <0.1  dB  Stop  bandwidth   2*3  dB  Stop  band  attenuation   ≥  36  dB  Source  impedance   50  Ohms  Load  impedance   50  Ohms  Implementation   Quarter-­‐‑wave  coupled  lines  

 The characteristics of the substrate are described in the table below, using lines McLin (Coupled Microstrip Lines).

TABLE 2 CHARACTERISTICS COUPLED MICROSTRIPS LINES

Thickness  of  the  

dielectric  H=31  .5  mm  

Relative  permittivity  of  the  substrate  

εr=2.31  

Magnetic  permeability   µμr=1  Dielectric  loss   Tan  (δ)=0.0009  

Thickness  of  copper   T=1.2  mm  

Conductivity  of  copper   4.1  x10+7  S/m  

 In the above table dimensions coupled microstrip lines were synthesized using "LineCalc" a utility tool ADS (table 3).

TABLE 3 DIMENSIONS OF THE MICROSTRIP LINE

n   Width  (mm)  

Separation  (mm)  

Length  (mm)  

 1   69.23   6.23   904.46  2   89.4   36.96   884.25  3   90.57   49.82   883.06  4   90.57   49.42   883.06  5   89.4   36.96   884.25  6   69.23   6.232   904.46  

  The design of the filter in ADS software is presented in figure 5 and the layout in figure 6:

 Fig. 5. Design of the band-pass filter coupled microstrip lines

 Fig. 6 Layout of the band-pass filter coupled microstrip lines

The reflection coefficient is shown in figure 7 and the transmission coefficients in figure 8.

 

 Fig. 7. Reflection coefficients of band-pass filter to microstrip lines coupled

 Fig. 8.  Transmission coefficients of the band-pass filter coupled mi-crostrip lines

The response of the filter coefficients at the end of S12 and S11 is shown in figure 9.

 

   

Fig. 9. Responses of the band pass filter at fo = 2.4 GHz

6

From the above curves (Figures 8 and 9), we can see that: The filter response is centered on 2.4 GHz with 16.259 dB. The two transmission coefficients S12, S21 are equal -1 .396 dB at the central frequency with bandwidth equals 10%. This meets the specifications of the load being asked (fig-ure 10).  

   

Fig. 10. Phase and amplitude of reflection coefficients S12 , S11 filter at fo =2.4GHz

Both figures above show us that we have: 1. A gain which is equal to 0.185 dB at the center fre-

quency for the reflection coefficient S11, with a phase shift of 58.56°at the same frequency.

2. A maximum gain is 0.85 and a phase shift order -22 ° for S12 at the center frequency. For frequencies outside the band phase shift introduced by the fil-ter decrease linearly as a function of frequency.

3. 3 Dual Behavior Resonator: Order 2 DBR The filter is that we must develop a band pass filter of order 2 DBR at 2.14 GHz with a RF channel passive trans-ceiver This topology is easily achievable in microstrip technology, is perfectly suited to our problem . And that the lines are based micro-strip lines using implanted un-der ADS LineCalc tool. The table below shows the charac-teristics of the substrate (table 4):

TABLE 4 CHARACTERISTICS OF THE SUBSTRATE IN 2 DBR

Thickness  of  the  dielectric   H=0.762  mm  

Relative   permittivity   of   the  substrate  

εr=3.5  

Magnetic  permeability   µμr=1  Dielectric  loss       Tan  (δ)=0.002  Thickness  of  copper       T=17.5  µμm  

Conductivity  of  copper   10+7  S/m  

 The filter conception by ADS is given in figures 11 and 12:

 Fig. 11. Phase Diagram of the order 2 DBR band-pass filter  

 Fig. 12. Response of the order 2bandpass DBR filter

From the above figure 13 with the filter response DBR we find that:

1. DBR filter of order 2 we have achieved, we provided each DBR two transmission zeros on both sides of the bandwidth.

2. These zeros are disjoint (DBR different) the first two zeros low frequency respectively at 1.99 GHz and 2.02 GHz. The two zeros posi-tioned at 2.42 GHz and 2.77 GHz at high fre-quency.

3. The response of the filter at the centered fre-quency 2.21 GHz is -20.22 dB, with the bandwidth that is equal to 6.78%.

Using Momentum is a 2D electromagnetic simulator im-planted under ADS and based on the method of mo-ments. One can make an electromagnetic simulation of DBR bandpass filter of order 2. This electromagnetic si-mulation tool is very accurate in microstrip technology, and measurement results are often very close simulations. 3.4 Design of a dual mode filter ring We will propose a design method of dual-mode filters, disrupted at 135° doors, which uses the concepts of ad-mittance inverter. Suppose we want to design a filter type that has Tchebysheve 0. 1dB ripple in the pass band, with a normalized bandwidth of 2% and a center frequency fo = 5.8 GHz and ends at Z = 50 Ω loads. With these cons-traints, the tables give g1 = 0.8430 and 9; g2=0622. Using equation:

                                                        2.11

12

ggw

bJk ==                       (12)                                                                                                                                                          

So, the coupling coefficient k= -0.0276. The frequencies fp

7

and fi can be calculated by using equation:

kff p−

=10      ,  

kffi+

=10                          (13)                                                                                                                    

We find   5.721pf GHz= and GHzfi 881.5= .  Since, we used perturbation at 135°, such as oni ff = .  If Y0a  =  1/50 1−Ω   , the value of the perturbation C135=  0.047  pf.    The value of the capacity C is calculated by: 0

02a

n

YC f=  C  =  1.70  pf. Then, the value of b1 and 01J is calculated us-ing the equation:  

0 01

0

1 101

1

a nY fbf

G b wJg

π⎧ =⎪⎪⎨⎪ =⎪⎩

            (14)

 

We find b1 = 0.064 and 301 5.498.10J −= .    Now we need to

find a pair that allows for this inverter admittance. If we assume that Cc1 = Cc3 = 0, Cc2 = 0.163 pf and °−== 35.1521 φφ .  To achieve the desired filter, we use a

ring resonator which has an impedance   Y0a   =   1/50 1−Ω  and the central frequency   GHzff oni 881.5==   dis-turbed at 135°, with the perturbation capacity 0.094 Pf. The coupling must be done with a capaci-ty pfCc 163.02 = .  Must add more transmission lines of

electrical length °−= 35.152φ ,   at central frequency f = 5.8 GHz. Table 1 gives the results of some filters Che-bytshev designs.

TABLE 5 CONCEPTION RESULTS

 Renflement  

 (dB)  0.1   0.1   0.2   0.2   0.5   0.5  

g1   0.84   0.84   1.03   1.03   1.40   1.40  

g2   0.62   0.62   0.67   0.67   0.70   0.70  

w(℅)   1.0   2.0   1.0   2  .0   1.0   2.0  

k( 310− )  -­‐‑13.8   -­‐‑27.6   -­‐‑11.9   -­‐‑23.9   10.0   -­‐‑20.0  

)( 10

−ΩaY   0.02   0.02   0.02   0.02   0.02   0.02  

( )nf GHz   5.84   5.88   5.83   5.87   5.83   5.86  

)(2 135 fFC   47.65   95.5   41.2   82.6   34.6   69.3  

)(2 fFCc   110   163   98.7   144   84   122  

(deg)2φ   -­‐‑10.92   -­‐‑15.34  

-­‐‑9.86   -­‐‑13  .87   -­‐‑8.55   -­‐‑12.00  

 The filter structure as designed by ADS software (figure 13) and the response in figures 14, 15.

 

Fig. 13.   Design of the double mode filter at f=5.8 GHz with ren-flement 0 .1 dB

 

 Fig. 14. Response of the double filter at fo = 5.8 GHz    

 Fig. 15. Coefficient S11 at fo = 5.8 GHz

 Fig. 16. S11 phase and S21 dual-mode filter centered at fo = 5.8 GHz

 Fig. 17. Coefficients S11 and S22 of the double filter at fo = 5.8 GHz

From the two curves (Figure 16 and Figure 17), we de-duce that: The response of the filter is 21.55 dB at the cen-tered frequency 5.8 GHz. The width of the bandwidth of the filter is equal to 3.8%. This difference from what it has taken in support of the specifications of the load that the effect may be due to the discontinuity between the differ-ent lines,

1. The two reflection coefficients have the same amplitude and phase at the resonant frequency.

2. A maximum gain which is equal at the center

8

frequency. 3. For frequencies outside the band phase shift in-

troduced by the filter decreases linearly as a function of frequency.

3.5 Design of the band-pass filter coupled in paral-lel lines The band-pass filter 12 GHz is that we want to design based on the transmission line. So we must set the sub-strate to be used. For this, we must first insert an element substrate, MSUB, using the palette TLINE-Microstrip. Characteristics of the substrate are described in the table below (table 6):

TABLE 6 CHARACTERERISTICS OF THE SUBSTRATE MSUB

Characteristic   rε   H  (mm)   T  (mm)     tan( )δ

 Values   3.78   15   0.1   0.00  

 We used simple microstrip lines (MLIN) and coupled (MCFIL) to simulate the filter in ADS software (figure 18) and its response is shown in figure 19:

 Fig. 18. Diagram of the band-pass filter in parallel coupled line centered at 12 GHz

   

Fig. 19. Responses band pass filter centered at fo = 12 GHz

 Fig. 20. Amplitude and phase of S21 and S11 reflection coefficient (dB) filter fo = 12GHz

 Fig. 21. Coefficients S11 and S22 of the filter at f0 = 12 GHz

1. The simulation results of the band-pass filter coupled in parallel lines (Figures 20, 21 and 23) show that: The reflection coefficient S11 is -58.71 dB at fre-quency 11.98 GHz,with a bandwidth is equal to 10.68%.

2. The maximum gain of 1 at the frequency 12 GHz, with a linear decrease of the phase between [9 - 9.8] GHz and [13.4 - 15] GHz.

3. For the reflection coefficients have the same am-plitude and phase at the center frequency. It can automatically generate the layout of the circuit of the bandpass filter coupled to parallel lines as shown in the figure below (Figure 22).

 

 Fig. 22.  Layout of the pass-band filter  with lines coupled in parallel  

3.6 Design and synthesis filter ring The filter ring has an interest in addressing functions in the W-band (94 GHz) due to its degrees of freedom to independently control its characteristic parameters (fre-quency, bandwidth, frequency zeros). This filter has a pseudo-elliptical response characterized by a center fre-quency between two transmission zeros. We define the  following parameters: . of r ftz f=   :   Frequency of the first

9

transmission zero:

2

0

2

0

2.cos1

2.sin

⎟⎟⎠

⎞⎜⎜⎝

⎛+

⎟⎟⎠

⎞⎜⎜⎝

⎛

=

ff

ff

Ttz

tz

zπ

π

        (15)

          (20)  

With   rf   the ratio between the frequency of the transmis-sion zero and the first center frequency and  fo  is the Cen-ter frequency and Z0e is the characteristic impedance of the coupled lines the even mode and Zo is the normaliza-tion impedance.From these parameters, we define the following intermediate values:

 

( )2 2 2 2

20

3 2 20

2 20 0 0

2

(1 )( 1)

10

4 ( 2) 4 ( )

2 ( ( 2) )

( ( 2) 2 )

Q z o

y

o z e o z z Q o

Z e Q

oe Q

S x T Z

x

P xZ T Y Z x T x S Z

Q T Y Z Z x S

R Y Zo Zo x S

⎛ ⎞⎜ ⎟⎝ ⎠

⎧ = − −⎪⎪⎪⎪ = −⎪⎪

= − − + −⎨⎪

= − +⎪⎪

= − − +⎪⎪⎪⎩

                                             (16)  

These parameters are used to calculate the characteristic impedance of the quarter-wave lines and odd mode char-acteristic impedance of the coupled lines:

20 0 0 0 2

0

0

2

2

(2 )

Z Z e e Qe

re

Z oeoo

zr oe

r

P Q RT x T Y Z Y Z S xY x

ZxY

T YY TZ YZ

⎧ ⎛ ⎞+ ++ + − + −⎪ ⎜ ⎟⎜ ⎟⎪ ⎝ ⎠=⎪⎪

⎨⎪

=⎪−⎪

⎪⎩

    (17)

 

We obtain the design values in Table 7:

TABLE 7 DESIGN OF THE RING FILTER

 fo(GHz)  

fr   Y(dB)   Zoe(Ω )   Zo(Ω )  

94   0.86   0.03   90   50  

After calculation:  Zoo=14.88  Ω  and Zr=  59.59  Ω .  Filter design in ADS software (figure 23) and its response (figure 24).

 Fig. 22. Design of the ring filter at fo=94 GHz

 Fig. 23.  Response of the order 2 ring filter at 94 GHz

 Fig. 24.  Coefficient S11 and transmission S22 of the ring filter at 94 GHz

 Fig. 25.  Coefficient  S11  and  S22  of  the  filter  at  fo  =  12  GHz  

According to the previous figures (Figures 23, 24 and 25),we note that the response of the filter at 94 GHz is -21.697 dB, with the bandwidth of 8 .5% and transmission zeros LF and HF positioned to 80.75 GHz and 107.3 GHz.

4. CONCLUSION This paper has been to analyze and filter design for tele-com applications narrowband, using a simulation tool powerful software ADS (Advanced Design System) which allows simulations with various libraries active and passive microwave. Then, we studied the synthesis of five topologies narrowband filter. Design a bandpass fil-ter with microstrip lines coupled (McLin) Chebychev type of order 5 at the center frequency f = 2.4 GHz with 10% bandwidth. The topology of a dual Behavior Resonator filter order 2 of a chain 2.14GHz passive RF transceiver is analysed. This topology has a good performance with additional degree of freedom introduced by the transmission line and the end we generate the filter mask DBR. The designs of a filter type that Tchebysheve 0. 1dB ripple in the pass band with a standard bandwidth of 2% and a center fre-quency fo= 5.8 GHz using a ring resonator. The Design of a band-pass filter coupled in parallel lines at a center frequency fo=12 GHz based microstrip lines, and using Momentum simulator we arrived to generate the layout of the filter it can be used for the production of a electromagnetic simulation.

10

The application of synthesis to design the ring resonator side coupled to access quarter wavelength for a center frequency of 94 GHz. This filter has a pseudo-elliptical response characterized by a center frequency between two transmission zeros. The ring topology also has shown its effectiveness in terms of frequency agility while main-taining good performance in adaptation.

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Adiba El fadl was born in Meknes, Morocco on January 24, 1979.She received the master in telecommunication from university of Fes, Morocco, in 2000. She received the DESA (diploma of deep-ened studies) in System of Telecommunication from University of Tetouan, Morocco, in 2004. She is currently working toward the Ph.D degree in Laboratory Systems and Telecommunications Engineering Decision, at Kenetra University. Her current research interests in-clude digital signal processing and adaptive antenna problems. Lekbir Belrhiti was born in Erfoud. He received his Master degree in Telecommunications at the ENSAF in 2012.

Hamid Bouyghf was born in Errachidia, in Morocco in 1982. He received the Engineer’s degree in Micro-electronics and telecommu-nications systems from FST- Fez in 2007. He is currently working towards the Ph.D. degree in electronic engineering in collaboration between FS-Meknes and IEMN-Lille. He interested in new applica-tions of microwaves and communic tions systems. He is now a pro-fessor at Electrical Engineering Department in High Scool of Tech-nology, Moulay Ismail University, Meknes-Morocco. Seddik Bri was born in Errachidia - Morocco. He received his PHD degree in Microwave radiometry by correlation and Application in biomedical Engineering at the Automatic Laboratory and Micro-waves, Université Ibn Tofail, Sciences Faculty, Kenitra and the HDR at the Mohmadia Scholl Engineering – Rabat in 2004. He is now a professor at in the Department of Electrical Engineering and Head of the Electrical Engineering at the High Scoll of Technology (Ecole Supérieure de Technologie de Meknes: ESTM - Moulay Ismail Uni-versity, Meknes – Morocco. His research activity focuses on Micro-wave applications in the design of communication systems in collab-oration with IEMN-Lille-France. Author of forty publications in interna-tional journals and more than sixty papers in refereed conferences. Reviewer in the international journal: JEWA, IJSAT and PIERS.