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Thesis work for the degree of Licentiate of Technology Sundsvall 2012
Analysis and Implementation of Switch Mode Power Supplies in MHz Frequency Region
Abdul Majid
Supervisors: Associate Professor Kent Bertilsson Professor Bengt Oelmann
Electronics Design Division, in the Department of Information Technology and Media
Mid Sweden University, SE-851 70 Sundsvall, Sweden
ISSN 1652-8948 Mid Sweden University Licentiate Thesis 80
ISBN 978-91-87103-13-1
ii
Akademisk avhandling som med tillstånd av Mittuniversitetet i Sundsvall framläggs till offentlig granskning för avläggande av teknologie Licentiate examen i elektronik onsdagen den 14 Juni , klockan 10:15 i sal O111 , Mittuniversitetet Sundsvall. Seminariet kommer att hållas på engelska.
Analysis and Implementation of Switch Mode Power Supplies in MHz Frequency Region
Abdul Majid © Abdul Majid, 2012 Electronics Design Division, in the
Department of Information Technology and Media
Mid Sweden University, SE-851 70 Sundsvall
Sweden
Telephone: +46 (0)60 148705
Printed by Kopieringen Mittuniversitetet, Sundsvall, Sweden, 2012
iii
ABSTRACT
A power supply is an essential part of almost every electronic device and the
current trend is towards the miniaturization of these devices. It is thus desirable to
also attempt to reduce the size of the power supply and it is possible to achieve this
objective by increasing the power density which is attainable by decreasing the size of
the passive/energy storage components such as the inductors, capacitors and the
transformer. The size of these components can also be decreased by increasing the
switching frequencies.
Linear power supplies use bulky line frequency transformers and heat sinks and
are thus not capable of providing a significant opportunity to reduce their size and
weight. Switch Mode Power Supplies (SMPS) use higher switching frequencies,
which replaces the bulky line frequency magnetics by smaller high frequency
magnetics which are then able to offer significant size and weight reductions. The
efficiency and size of the SMPS depends on a suitable switching frequency.
Previously, the SMPS were implemented using bipolar power devices and their
switching frequency range was limited to a range of a few kHz. With the availability
of modern and efficient power MOSFETs, it is possible to switch the SMPS from
several kHz to a MHz range. In addition, core based transformers were previously
used in SMPS. These transformers have hysteresis and eddy current losses. Their
switching frequency was limited to several hundreds of kHz. Recent research has
produced energy efficient multilayered PCB transformers which can be implemented
in SMPS for power and signal transfer applications, in the MHz frequency range.
Thus, with the emerging power devices in GaN and SiC technology and the
development of high frequency multilayered PCB power transformers, it is now
possible to design high frequency and power efficient isolated converters.
The focus of the research is to design, implement and evaluate energy efficient
AC-DC and DC-DC isolated converter topologies. These converters are designed by
using the latest power electronic devices and PCB transformers. They are switched in
the MHz frequency range.
In this thesis, two DC-DC converters, half bridge and full bridge, are designed,
implemented and evaluated. These converters are switched in the MHz frequency
range. The energy efficiency of the converter is measured and analysed by varying
different circuit parameters. Feedback analysis is made in the case of the Half Bridge
converter. The opto-coupler and auxiliary feedback techniques are implemented,
measured and analysed in a high frequency half bridge converter using a PCB power
transformer. The feasibility of the feedback signal, using the auxiliary winding of a
PCB power transformer, is discussed.
The multilayered PCB transformers used in the converter circuits have provided a
major contribution with regards to both the energy efficiency and size compactness.
This research work is a initial step in the design, implementation and analysis of
SMPS operating in the MHz frequency region, using PCB transformers.
v
ACKNOWLEDGEMENTS
Countless thanks to ALLAH Almighty, the Creator of all of us, worthy of all
praise, Who guides us in darkness and helps us in difficulties; who enabled me to
work on this thesis and complete it.
I am deeply indebted to my supervisors Dr. Kent Bertilsson and Prof. Bengt
Oelmann for their kind support and valuable guidance. The help, encouragement and
motivation provided by Dr. Kent Bertilsson is gratefully acknowledged. His
enthusiasm and immense knowledge helped me in all times of research and writing of
this thesis.
I express my gratitude to my fellow research group members, Jawad Saleem, Hari
Babu Kotte and Radhika Ambadipudi for their help motivation and expert opinions.
I am very grateful to my friends Naeem Ahmad, Khurram Shahzad, Khursheed,
Abdul Waheed Malik, Muhammad Imran, Mazhar Hussain, Muhammad Amir Yousaf
and Anzar Alam, for their help and motivation.
I acknowledge my colleagues Stefan Haller, Fanny Burman, Lotta Söderström,
Krister Alden, Benny Thörnberg, Class Mattsson, Cheng Peng, Najeem Lawal, and
Sebestian Bader.
I acknowledge the Higher Education Commission and Mid Sweden University for
their financial support.
My sincere thanks go to my parents, brothers and sisters who have encouraged and
prayed for me during my studies. Thanks to my younger sister Amina Jabeen, who
visited me to take care of my children. With her help I was able to spend more time on
my research work.
Finally, I am very thankful to my wife Dr. Shazia Jamil for her unconditional co-
operation and untiring support throughout the years of my studies. I am thankful to my
delightful and cute children, Muhammad Talha, Muhammad Zubair and Madeeha
Ftima for taking away all my day-time worries with their innocent smiles.
Sundsvall, May 2012
vii
TABLE OF CONTENTS
ABSTRACT ....................................................................................................................... III
ACKNOWLEDGEMENTS ................................................................................................ V
ABBREVIATIONS AND ACRONYMS ......................................................................... IX
LIST OF FIGURES .......................................................................................................... XI
LIST OF TABLES ......................................................................................................... XIII
LIST OF PAPERS ............................................................................................................ XV
THESIS OUTLINE ....................................................................................................... XVII
1 INTRODUCTION ....................................................................................................... 1
1.1 LINEAR VERSUS SWITCH MODE POWER SUPPLIES ................................................... 2 1.2 MOTIVATION FOR USING HIGHER SWITCHING FREQUENCIES ................................... 3
1.2.1 High frequency power electronic devices .................................................... 3 1.2.2 High frequency power transformers: ........................................................... 4 1.2.3 EMI filter size reduction .............................................................................. 4
1.3 CHALLENGES IN HIGH FREQUENCY CONVERTER DESIGN ........................................ 4 1.4 MAIN FOCUS OF THE THESIS ................................................................................... 5
2 HALF BRIDGE CONVERTER ................................................................................. 7
2.1 TRANSFORMER DESIGN FOR HALF BRIDGE CONVERTER .......................................... 9 2.2 MULTILAYERD PCB TRANSFORMERS ...................................................................... 9
2.2.1 Parameters of PCB transformer ................................................................. 10 2.2.2 Energy efficiency of the transformer ......................................................... 11
2.3 SIMULATION MODEL OF HALF BRIDGE CONVERTER ............................................. 11 2.4 HARDWARE IMPLEMENTATION............................................................................. 12 2.5 MEASUREMENT RESULTS OF THE CIRCUIT ............................................................ 14 2.6 LOSS ESTIMATION OF HALF BRIDGE CONVERTER .................................................. 16
3 FEED-BACK TECHNIQUES IMPLEMENTED IN HALF BRIDGE
CONVERTER ............................................................................................................ 19
3.1 DESCRIPTION OF FEEDBACK CIRCUIT.................................................................... 19 3.1.1 Opto-coupler feedback ............................................................................... 19 3.1.2 Auxiliary feedback ..................................................................................... 21
4 FULL BRIDGE CONVERTER DESIGN ............................................................... 25
4.1 SIMULATION MODEL ............................................................................................ 26 4.2 HARDWARE IMPLEMENTATION AND MEASUREMENT RESULTS ............................. 26 4.3 PARAMETERS OF PCB TRANSFORMER USED IN FULL BRIDGE CONVERTER ............. 27 4.4 DESIGN LIMITATIONS ........................................................................................... 28
5 THESIS SUMMARY AND CONCLUSIONS ......................................................... 29
5.1 FUTURE WORK ..................................................................................................... 30
viii
6 PAPER SUMMARY ................................................................................................. 31
6.1 HIGH FREQUENCY HALF-BRIDGE CONVERTER USING MULTILAYERED
CORELESS PRINTED CIRCUIT BOARD STEP-DOWN POWER TRANSFORMER .......................... 31 6.2 ANALYSIS OF FEEDBACK IN CONVERTER USING CORELESS PRINTED CIRCUIT
BOARD TRANSFORMER ..................................................................................................... 31 6.3 HIGH FREQUENCY FULL BRIDGE CONVERTER USING MULTILAYER CORELESS
PRINTED CIRCUIT BOARD STEP UP POWER TRANSFORMER PLATFORM ............................... 31 6.4 AUTHORS’ CONTRIBUTIONS ................................................................................ 33
REFERENCES .................................................................................................................. 34
PAPER I .............................................................. ERROR! BOOKMARK NOT DEFINED.
HIGH FREQUENCY HALF-BRIDGE CONVERTER USING MULTI-LAYERED CORELESS
PRINTED CIRCUIT BOARD STEP-DOWN POWER TRANSFORMERERROR! BOOKMARK NOT DEFINED.
PAPER II ............................................................. ERROR! BOOKMARK NOT DEFINED.
ANALYSIS OF FEEDBACK IN CONVERTER USING CORELESS PRINTED CIRCUIT BOARD
TRANSFORMER ........................................................... ERROR! BOOKMARK NOT DEFINED.
PAPER III ........................................................... ERROR! BOOKMARK NOT DEFINED.
HIGH FREQUENCY FULL BRIDGE CONVERTER USING MULTILAYER CORELESS PRINTED
CIRCUIT BOARD STEP UP POWER TRANSFORMER ......... ERROR! BOOKMARK NOT DEFINED.
ix
ABBREVIATIONS AND ACRONYMS
AC ............. Alternating Current
ADC ............. Analogue to Digital Converter
CTR ............. Current Transfer Ratio
FR4 ............. Flame Retardent 4
DC ............. Direct Current
EMC ............. Electro Magnetic Compatibility
EMI ............. Electro Magnetic Interference
HEMT ............. High Electron Mobility Transistors
IR ............. Infra-Red
GaN ............. Gallium Nitride
LED ............. Light Emitting Diode
MOSFET ............. Metal Oxide Semiconductor Field Effect Transistor
PCB ............. Printed Circuit Board
PWM ............. Pulse Width Modulation
SiC ............. Silicon Carbide
SMPS ............. Switch Mode Power Supplies
ZVS ............. Zero Voltage Switching
xi
LIST OF FIGURES
Figure 1 Linear Power Supply Block diagram .............................................................. 1 Figure 2 Block diagram of Switch Mode Power Supply ............................................... 2 Figure 3 Half Bridge Converter [5] ............................................................................... 7 Figure 4 Coreless PCB Transformer ........................................................................... 10 Figure 5 Energy efficiency of power transformer used in the converter circuit [21] .. 11 Figure 6 Circuit diagram of half bridge converter ....................................................... 12 Figure 7 Simulated Gate Signals for high side and low side MOSFETs .................... 12 Figure 8 Measured Gate signals for MOSFETs .......................................................... 13 Figure 9 Measured Gate signals with two MOSFET drivers ...................................... 14 Figure 10 Energy efficiency as a function of output power ........................................ 15 Figure 11 Energy efficiency as a function of switching frequency ............................. 16 Figure 12 Thermal profile of MOSFETS .................................................................... 18 Figure 13 Thermal profile of rectifier diode................................................................ 18 Figure 14 Opto-coupler diagram ................................................................................. 20 Figure 15 Schematic diagram of half bridge converter using opto-coupler feedback . 21 Figure 16 Variation of Vopto with variation of Vout for different loads .................... 21 Figure 17 Schematic diagram of half bridge converter using auxiliary feedback ....... 22 Figure 18 Variation of Vaux with Vout for different loads ......................................... 23 Figure 19 Variation of input current with Vout for different loads ............................... 24 Figure 20 Measured and calculated values of Vout ..................................................... 24 Figure 21 Full Bridge converter[5] ............................................................................. 25 Figure 22 Schematic diagram of full bridge converter ................................................ 26 Figure 23 Variation of energy efficiency with the variation of switching frequency .. 27 Figure 24 Energy efficiency of the coreless PCB power transformer ......................... 28 Figure 25 Thermal profile of MOSFETs ..................................................................... 28
xiii
LIST OF TABLES
Table 1 Comparison of different power supply technologies [2] .................................. 3 Table 13-1.Authors’ Contributions ............................................................................. 33
xv
LIST OF PAPERS
This thesis is mainly based on the following three papers, herein referred to by
their Roman numerals:
Paper I High Frequency Half-Bridge Converter using Multilayered
Coreless Printed Circuit Board Step-Down Power
Transformer
A.Majid, H.B.Kotte, J.Saleem, R.Ambatipudi, S.Haller,
K.Bertilsson
Proceedings of 8th Internal Conference on Power Electronics-ICPE 2011 at
Jeju South Korea (May 28-June 03 2011)
Paper II Analysis of Feedback in Converter using Coreless Printed
Circuit Board Transformer
A.Majid, J.Saleem, H.B.Kotte, R.Ambatipudi, S.Haller,
K.Bertilsson
Proceedings of International Aegean Conference on Electrical Machines and
Power Electronics & Electromotion-ACEMP 2011 at Turkey Istambul
(September 08-10, 2011)
Paper III High Frequency Full Bridge Converter using Multilayer
Coreless Printed Circuit Board Stepup Power Transformer
J.Saleem, A. Majid, R.Ambatipudi, H. B. Kotte,
K. Bertilsson
Proceedings of 20th European Conference on Circuit Theory and Design
Linköping Sweden August 29- 31,2011
Papers not included in this thesis
Paper IV Study the influence of the alignment and nearby metallic
objects on the Hall sensor system for current measurement
in Resistance Spot Welding Machine.
J.Saleem, A. Majid, K.Bertilsson, Linda Schuberg
Proceedings of International Conference on Environment and Electrical
Engineering 2011Rome Italy 8th May—11th May 2011
Paper V A Study of IGBT Rupture Phenomenon In Medium
Frequency Spot Welding Machine.
J.Saleem, A. Majid, S. Haller, K. Bertilsson
Proceedings of International Aegean Conference on Electrical Machines and
Power Electronics & Electromotion-ACEMP 2011 at Turkey Istambul
(September 08-10, 2011)
xvii
THESIS OUTLINE
Chapter 1: This chapter provides brief introduction of linear and switch mode power
supplies. Design issues of high frequency SMPS are discussed.
Chapter 2: In this chapter the design of half bridge converter is discussed. The
simulation results are discussed. The energy efficiency of the converter is computed
by varying various circuit parameters.
Chapter 3: This chapter is mainly focused on the feedback control techniques used in
half bridge converter. Two feedback techniques are analysed. In the first technique the
opto-coupler is used on the secondary side of the converter and in the second
technique the auxiliary winding of the transformer is used for the feedback signals.
Chapter 4: The design of high frequency full bridge DC-DC converter using coreless
PCB step-up transformer is discussed in this chapter
Chapter 5: presents a papers summary.
Papers which are basis for this research work are listed at the end.
1
1 INTRODUCTION
A power supply is an essential part of every electronic device. It converts an AC
power line voltage to a steady state DC output voltage as required by all electronic circuits.
Power supplies can be categorized into two types: linear and Switch Mode.
The block diagram of a linear regulated power supply is shown in Figure 1. It consists
of a transformer, rectifier, and filtering and regulator circuits. The rectifiers are used to
convert 50/60Hz AC voltage to a pulsating DC. The filters are used to convert the pulsating
DC to a smooth voltage. Finally, the voltage regulators are used to produce a constant
output voltage, irrespective of the variations in the ac line voltage or by the circuit loadings.
The output voltage is sampled by an error amplifier block which compares it with a
reference signal and generates an error signal on the basis of comparison. The error signal
is applied to a series pass element which alters its resistance accordingly and thus regulates
the output voltage. Hence, the series pass element absorbs any changes in the input voltage
and any dynamic changes in the output voltage due to the load changes, within its designed
tolerance band [1]. The size of the components, such as the transformers and output filters,
are very bulky in these supplies. Linear power supplies are not suitable for smaller modern
electronics system because of their high power loss, low power density and bulky size.
Figure 1 Linear Power Supply Block diagram
Like linear power supplies, the SMPS converts the available unregulated AC or DC
input to a regulated DC output. The input is taken from the AC mains and then rectified,
filtered and fed to a high frequency DC-DC converter. SMPS can be operated within the
kHz to MHz frequency range. The increased switching frequencies cause a decrease in the
size of the energy storage elements, such as the capacitors, inductors and transformers, in
an almost linear manner. SMPS is further divided into two types: isolated and non-isolated.
2
Common examples of non-isolated switch mode power supply topologies are the
buck, boost and the buck-boost converters. This type is a common building block for many
complex power supply topologies. This type is the simplest and requires less components in
comparison to those for isolated supplies.
In an isolated type, a high frequency transformer is used to isolate the output from the
input, scale the output voltage magnitude with turn ratio and to produce the multiple DC
outputs. Large step-up or step down ratios can be achieved in converters by using a
transformer. The voltage/current stresses on transistors and diodes can be minimized by
means of the correct choice of transformer turn ratio.
The block diagram of an isolated SMPS circuit is shown in Figure 2.
Figure 2 Block diagram of Switch Mode Power Supply
1.1 LINEAR VERSUS SWITCH MODE POWER SUPPLIES
The linear power supplies are bulky and less efficient in comparison to those of
SMPS. In high current equipment, the size of linear power supplies becomes larger because
of the Mains-frequency transformer and the large sized heat sink. Each linear regulator has
an average energy efficiency of 35% to 50% and the losses are dissipated as heat [2].
However, the linear regulated supplies have better performance when compared with SMPS
in terms of line and load regulations, output ripple, EMI, transient recovery and hold up
time requirements [3].
The SMPS are much more efficient and flexible than the linear regulators and it is
possible to achieve a high energy density by using them. They are portable and are
commonly used in aircrafts products, laptop adaptors, off line applications and small
instruments. Due to their smaller size they require less heat-sinking for the same output
power as compared to that for linear power supplies. The PWM SMPS are switched at a
very high frequency and at these frequencies, especially at high voltages, the switching
losses increase and thus limit the performance of the supply. To overcome this problem,
different soft switching techniques are often used and resonant and quasi resonant
switching power supplies are emerging. Recent research has focused on the design of low
profile, energy efficient and compact switch mode power supplies.
3
Although both linear and switch mode power supplies perform the same function of
converting an unregulated input voltage to a regulated output voltage but they have
significant differences in their properties and performance. The important differences are
mentioned below [4][5]:
• Linear regulators exhibit energy efficiency of 20 to 60% while SMPS have energy
efficiency of 75-95%.
• The SMPS transformers are subjected to the non-sinusoidal input/output currents
and voltages, whereas the transformers in linear supplies are subjected to
sinusoidal voltages and currents.
A comparison of different power supplies is given in Table 1.
Table 1 Comparison of different power supply technologies [2]
Linear
Regulators
PWM switching
regulators
Resonant switching
regulators
Quasi-Resonant
switching regulators
Cost Low High High High
Mass High Low-medium Low-medium Low-medium
RF Noise None High Medium Medium
Efficiency 35-50% 70-85% 78-92% 78-92%
Multiple
outputs
No Yes Yes Yes
Complexity Less Medium High High
1.2 MOTIVATION FOR USING HIGHER SWITCHING FREQUENCIES
New devices and topologies are the driving force behind the development of power
electronics. By increasing the switching frequency of the converter, it becomes possible to
reduce the size of the SMPS which is a desirable feature. The main objective of this trend
is to increase the power density and to improve the dynamic performance of the converter
by decreasing the size of the passive components such as inductors, capacitors and
transformer [6]. The motivating factors behind the design of high frequency SMPS are
based on the fabrication of high frequency power electronic devices and PCB power
transformers.
1.2.1 High frequency power electronic devices
The CoolMOS Power MOSFETs offer significant advantages at high power levels.
They can be operated with the lowest control power, cheapest drive circuitry and the
highest switching frequencies. These are suitable for applications including efficient power
supplies and low power applications such as for a battery charger, line adapter and auxiliary
supplies [7].
A great achievement from the development in power semiconductors is the wide
band-gap technology and the related power devices. Silicon Carbide (SiC) is an emerging
4
semiconductor material and offers a higher breakdown voltage, thermal conductivity,
saturated electronic drift velocity and higher junction temperature operation [8].
Gallium Nitride (GaN) is another emerging semiconductor material and GaN-based
semiconductors devices have the capability of high-power switching with sub-microsecond
and nano-second times. They are suitable for high power and high temperature applications.
These devices have high electron mobility and saturation velocity, high sheet carrier
concentration at hetero-junction interfaces, high breakdown voltages, and low thermal
impedance [9].
1.2.2 High frequency power transformers:
Multilayered PCB power transformers are highly energy efficient and can be used in
SMPS for power transfer applications in MHz frequency range. Recent research has shown
that high frequency and high energy efficiency of the step down power transformers is
possible [10].
Hence, with the fabrication of efficient high frequency electronic devices and with the
design of high frequency multilayered PCB power transformers it is now possible to design
high frequency and power efficient isolated converters.
1.2.3 EMI filter size reduction
The increased switching frequency in combination with sudden changes in current
(di/dt) or voltage(dv/dt) levels generates higher order harmonics. It produces EMI and
affects the Electro Magnetic Compatibility (EMC) of the SMPS. It is extremely important
to take EMC into consideration during the design phase of the SMPS. The line filter are
used for suppressing conducted EMI and in general, the size of the filter is expected to
decrease with increasing switching frequency of the SMPS.
1.3 CHALLENGES IN HIGH FREQUENCY CONVERTER DESIGN
Designing the high frequency power converter with a better energy efficiency and
reliability is a challenging task as there are more effects caused by the switching frequency
on power-supply system than merely on its size and efficiency. The designers encounter
severe challenges in relation to the best approach for minimizing the influential factors and
optimizing the types and values of the components.
There is always a trade-off between size, cost, weight and energy efficiency of power
supplies. If one parameter is solved, then the problem shifts to other parameters. For
instance, the improvement in power density of SMPS is achieved by reducing the size of
passive components, but this reduction in size demands a higher switching frequency which
in turn results in switching losses. The eventual effect is reduced energy efficiency.
Although, there are some topologies, e.g. resonant converters, which are used to reduce the
switching losses at higher switching frequencies the design of these topologies becomes
complicated for large line and load variations. Hence, it becomes difficult to control and
stabilize the output voltage. At megahertz frequencies, the semiconductor switch
experiences high stress which results in a switching loss.
5
Although GaN HEMT device structure exhibits better characteristics when compared
to silicon-based devices, but the gate drive design for GaN is complex because the internal
parasitic capacitances of the devices must be comprehensively considered. The Miller
capacitance (CGD) is a nonlinear function of voltage and it causes the total dynamic input
capacitance to become greater than the sum of the static capacitances. It slows the turn-on
and turn-off transition time of the transistor. It is a source of instability and dv/dt turn on
problems [11]. There are driving constraints due to a smaller CGD in GaN MOSFETs. The
presence of fast dv/dt together with an unfavorable ratio between the CGS and CGD increases
the risk of a Miller turn-on and a short-through in half bridge converters [12]. The
proportional relationship between the switching frequency and switching loss in
semiconductor devices is another factor to consider. The increase in switching frequency
will also increase the power loss. Thus, it becomes more challenging to design a high
frequency converter [14].
The influence of parasitic elements, thermal analysis and on-board PCB layout design
are important in order to achieve compact, higher power density and high efficient
converters.
1.4 MAIN FOCUS OF THE THESIS
The main objective of this thesis is to study the feasibility of energy efficient
DC-DC converters in the switching frequency range of MHz. At a higher switching
frequency the design of a transformer becomes a challenge. Recently multi-layered PCB
transformers have been designed at Mid Sweden University. The geometrical and electrical
parameters of different multi-layered PCB transformers are discussed in [10],[13],[20] &
[21]. These transformers are much smaller in size as compared to the core based
transformers. The energy efficiency of these transformers is more than 90% in the MHz
frequency region. Based on their smaller size, high energy density and better energy
efficiency, they can be implemented in SMPS for power and signal transfer applications, in
the MHz frequency range.
These multi-layered coreless PCB transformers are used in different converter circuits
and have offered major contribution in terms of energy efficiency and size compactness of
the converters [3].
The focus of this thesis is to use these multi-layered PCB transformers in the
implementation of converter circuits. These converters are switched in the MHz frequency
range. The transformers are used in step-down and step-up configurations. The energy
efficiency of the converters is computed by varying different circuit parameters. Different
feedback schemes are studied and implemented in the converters.
In the design process, the converter is analyzed by simulating the model using a
circuit simulation tool. The energy efficiency of the converter is computed by varying the
circuit parameters. The circuit components are optimized and after a successful evaluation
of the converters the prototype is built with integrated PCB power and signal transformers.
6
7
2 HALF BRIDGE CONVERTER
Half bridge converter circuits can be used in various applications such as power
supplies and motor drives and they are popular choice of converters upto several hundred
watts. A half bridge converter has two power switches that operate alternately to produce a
square wave AC at the input of the power transformer and are subjected to a voltage stress
equal to the input voltage. Two equal sized capacitors are used on the primary side to
provide a DC bias in order to balance the volt-second integrals of the two switching
intervals. Because of these capacitors the transformer sees a positive and negative voltage
during switching which results in twice the desired peak flux value of the core. Two
rectifier diodes are used on secondary side, which provide the rectification when one of the
power switches is on and act as freewheeling diodes during the time that both power
switches are off.
The transformer core is operated in the first and third quadrants of the B-H loop and
experiences twice the flux excursion of a similar forward converter core. A half bridge
converter utilizes the full available flux swing in both quadrants of the B-H loop, therefore
the primary transformer winding has half the turns for the same input voltage and power
[15].
Although, the structure of the half bridge circuit is complicated because isolated gate
drive signals are required for a high side MOSFET, there are certain advantages to using
this topology in DC-DC converter circuits. The advantage of the half bridge converter is
that the effective duty cycle, as seen by the input and output filters, is twice that of the
individual switch duty cycle. In addition, the effective frequency is twice that of the
individual switch frequencies. Both high efficiency and high power density are achievable
by using the half bridge converter topology [15]. The problem which occurs in this type of
circuits is shoot-through. Shoot-through occurs when both the transistors turn on at the
same time, thus creating a short across the main bus. This would be a disaster as it would
destroy both the transistors. This situation can occur at turn on, line or load transient, or
instability within the closed loop system [16]. The schematic diagram of the half bridge
converter is given in Figure 3.
Figure 3 Half Bridge Converter [5]
8
The operation of the half bridge converter is explained in the following section.
When the power switch S1 is on and S2 is off, the primary current Ip quickly ramps up
across the leakage inductance and delivers the secondary current. The rectifier diode D1 is
forward biased and D2 is reverse biased. The energy stored in the primary winding is
transferred to the load through the rectifier diode D1.
When S1 is turned off, the junction capacitance of switch S2 is charged by the current
flowing through the primary winding of the power transformer. After the voltage across
the drain-to-source of S2 drops to zero, the body diode of S2 conducts. It provides the path
for the energy stored in the leakage inductance of the primary winding. During the body
diode conduction period, S2 may be turned on at zero-voltage switching. The magnetizing
current that results in the interval during which S1 is on, flows through the secondary side.
The rectifier diode D2 is forced to conduct while D1 is already in a conduction state. Hence
both D1 and D2 conduct during this time and act as freewheeling diodes. They maintain the
current flow in the inductor. The magnetizing current is concurrently transferred to the
secondary side. Since both D1 and D2 are in parallel, therefore under ideal conditions, an
equal amount of current will flow through them and each diode will supply half of the load
current. In a practical scenario, the I-V characteristics of both the diodes are not exactly the
same and thus the flow of current is slightly different through two rectifier diodes. The
inductor current is the sum of the current flowing through both the diodes
When S2 is turned on and S1 is off D2 conducts and D1 is reversed biased. The output
voltage Vo across the load resistor is given as [5]:
( )DVinN
NVo ×=
1
2 2.1
where:
N1= Number of turns of primary winding
N2= Number of turns of secondary winding
D= Duty cycle of Switching waveform
The output power Pout is computed according to equation 2.2 below:
Loout IVP ×= 2.2
The energy efficiency of the converter is given as:
100×=in
out
P
Pη 2.3
9
2.1 TRANSFORMER DESIGN FOR HALF BRIDGE CONVERTER
In isolated SMPS the transformer is used to transfer the power efficiently and
instantaneously from an external electrical source to an external load. The choice of circuit
topology has a great impact on the transformer design. Full bridge and half-bridge
topologies offer the best transformer efficiencies because their cores and windings are fully
utilized.
There are two main functions of a forward mode transformer:
• It provides galvanic isolation between the input and output.
• It can either step-up or step down the primary voltage.
Ideally, a transformer does not store any energy however in practice, all transformers
store some undesired energy:
• Leakage inductance: It represents the energy stored in the non-magnetic regions
between the windings, caused by an imperfect flux coupling. In the equivalent
electrical circuit, the leakage inductance is in series with the windings.
• Mutual inductance: It represents the energy stored in the finite permeability of
the magnetic core and in the small gaps where the core halves come together. In
the equivalent circuit, mutual inductance appears in parallel with the windings.
The energy stored is a function of the volt-seconds per turn applied to the
windings and is independent of the load current.
2.2 MULTILAYERD PCB TRANSFORMERS
The high frequency magnetics are the backbone of modern SMPS. In core based
transformers, the magnetic cores are used to provide a high degree of magnetic coupling
and to reduce the leakage inductance. However these transformers cannot be used for the
converters operating in the frequency region of several MHz because of their increased core
losses, winding losses and temperature rise. The skin and proximity effects are the major
problem in high frequency transformer design. Leakage inductance and unbalanced
magnetic flux distribution are two further obstacles for the development of high frequency
transformers [17]. PCB transformers have the advantages of low costs, very high power
density, no limitation due to magnetic cores, no magnetic loss and ease of manufacturing.
They have the potential to be developed in microcircuits [18].
It has been found that coreless PCB transformers can be used for high-frequency
operations from about 200 kHz and upwards. The elimination of magnetic cores and the use
of printed windings open a new means of manufacturing low-profile transformers with a
high power density [19].
The PCB transformers can be used in converters operating in the MHz frequency
range. These PCB transformers possess very good high frequency characteristics, and they
eliminate the manual winding process. There has been significant research in the design of
coreless PCB transformers. These multi-layered coreless PCB transformers are highly
10
energy efficient [20] and have the potential to be used in both DC-DC and AC-DC
converters switching in the MHz frequency region [3] .
2.2.1 Parameters of PCB transformer
The 4:1 PCB centre-tapped power transformer, designed for a half bridge converter, is
used in the converter. It can be operated in the frequency range of 1 to 6 MHz. The
transformer is designed in four layers of PCB and it has two primary windings in the first
and fourth layers while there are two secondary windings in the second and third layers.
The primary and secondary copper windings of the transformers are etched on the FR4
PCB laminate and, by this means a better coupling between primary and secondary
windings is achieved. The primary windings are connected in series and the secondary
windings are connected in parallel so as to achieve the desired inductance. The number of
turns in the primary winding is 24 and in secondary winding, 6. The electrical parameters
of the transformer were measured using an RLC meter at 1 MHz and are:
• The self-inductances of the primary and secondary winding are 7.73µH and
2.33µH respectively.
• The leakage inductances of the primary and secondary winding are 1.38µH and
0.417µH respectively.
• The AC resistance of the primary and secondary windings of the transformer are
2.2Ω and 0.7Ω respectively.
• The inter-winding capacitance is approximately 50pF.
The 4:1 multi-layer coreless PCB power transformer and gate drive transformer used in
the circuit are shown in Figure 4.
Figure 4 Coreless PCB Transformer
Gate drive transformer
Power transformer Spiral inductor
11
2.2.2 Energy efficiency of the transformer
The maximum gain frequency of a PCB transformer is 8MHz and the maximum
impedance frequency is 3.5 MHz with an input impedance of 135Ω. The maximum energy
efficiency of the power transformer is approximately 96% at a frequency of 2.6MHz. The
power density of the transformer is 16W/cm2 [21]. The energy efficiency of this step down
power transformer for a load resistance of 10Ω and a sinusoidal excitation up to the output
power of 50W is illustrated in Figure 5.
Figure 5 Energy efficiency of power transformer used in the converter circuit [21]
2.3 SIMULATION MODEL OF HALF BRIDGE CONVERTER
The parameters are optimized by simulating the half bridge converter. CoolMOS
SPU02N60S5 power MOSFET SPICE model and the power transformer’s high frequency
model are used in this simulation. The maximum energy efficiency of the converter is
computed by variation of the circuit parameters. At best achieved conditions 85 percent
energy efficiency is observed in simulations The schematic diagram of the converter is
given in Figure 6.
High side and low side MOSFETs are switched alternately. The lower MOSFET is
switched directly by a pulsed waveform source while isolation transformer is used for the
high side MOSFET. The switching waveforms for driving high and low side MOSFETs are
shown in Figure 7.
12
PCB transformer
MicrocontrollerMOSFET
Driver
PW
M s
ignals
Isolation
Transformer
Vin
RL
M2
L_lks
L_lks
Lms
Lms
Lmp
Cps
L_lkpC1
C2
M1
D2
L
C
D2
Figure 6 Circuit diagram of half bridge converter
Figure 7 Simulated Gate Signals for high side and low side MOSFETs
2.4 HARDWARE IMPLEMENTATION
A high frequency half bridge converter circuit is implemented on a PCB. The
DSPIC33fJ016GS502 microcontroller is used in the converter so as to generate PWM
signals that are then fed to a MOSFET driver. It has the following important features and is
suitable for medium power converter topologies such as push pull and half bridge converts
topologies [22]:
• Individual time base and duty cycle for each of the six outputs of PWM generators
• Duty cycle, dead time, phase shift and frequency resolution of 1.04ns at 40MIPS
• Various PWM modes such as true independent, complementary and fixed off time.
VGS
High
VGS
Low
13
After passing through the driver, the signals are then fed to the gate of the lower
MOSFET. To drive the high side MOSFET, a multilayered coreless PCB isolation
transformer [20] is used after the gate driver. By using this gate drive transformer, a low
power consumption passive gate drive circuitry [3] is built up and utilized to drive the high
side MOSFET. Gate signals for the high and low side MOSFETS are shown in Figure 8
High Side Low Side
Figure 8 Measured Gate signals for MOSFETs
The rise and fall time of the gate signals for a high side MOSFET is more than that of
the low side MOSFET. In order to equalize the rise and fall times of both the high and low
side MOSFETs a driver is added into the circuit after the isolation transformer and before
the high side MOSFET. The Gate signals for both the high and low side MOSFETs while
using two MOSFET drivers, are shown in Figure 9.
The CoolMOS SPU02N60S5 Power MOSFETs M1 and M2 are used in the design.
These devices offer significant advantages at high power levels. They can be operated with
the lowest control power, the cheapest drive circuitry and the highest switching frequencies
[7]. The Drain-Source voltage (VDS) of the MOSFETs is 600 V, RDS(on) is 3 Ω and the
continuous drain current (ID) is 1.8 A. The output capacitance (Coss) of the MOSFET is 77
pF and the total gate charge Qg when VDD 350 V, ID 1.8 A and Vgs 1-10V is 9.5 nC [23].
On the secondary side, the SR 1660 Schottky barrier rectifier diode was used. It has a
maximum forward voltage of 0.7 V, an average forward rectified current capability of 16 A
and a maximum DC blocking voltage of 60 V [24].
14
Figure 9 Measured Gate signals with two MOSFET drivers
2.5 MEASUREMENT RESULTS OF THE CIRCUIT
The energy efficiency of the unregulated half bridge converter circuit is computed for
the input DC voltage of up-to 170 V and the switching frequency is in the range of
2 to 3 MHz, with different loads and a fixed duty cycle of 30%. The output power of the
converter at the maximum input voltage of 170 V is approximately 40 W. When computing
the efficiency, the power dissipation of the gate drive circuit is not included. The energy
efficiency (ƞ) is computed according to the following expression:
100100
2
××
=×=inin
L
out
in
out
IV
RV
P
Pη
2.4
The energy efficiency of the converter at a constant 2.5 MHz switching frequency,
30% duty cycle, different loads (10 Ω, 20 Ω and 40 Ω) and for different input voltages is
plotted in Figure 10. The converter is tested up-to a maximum output power level of 40 W.
At this power level the input voltage is 170V, switching frequency is 2.5 MHz, duty cycle
is 30% and the load resistance is 10 Ω.
High Side Low Side
15
Figure 10 Energy efficiency as a function of output power
The maximum energy efficiency of approximately 82% is obtained at the output
power level of 26 W. At this instant the input voltage is 170 V, switching frequency is
2.5 MHz, duty cycle is 30% and the load resistance is 20 Ω. From the measurement results
it is observed that there is a degradation in the energy efficiency of the converter as the
output power is increased beyond 26 W. At a higher output power and higher switching
frequencies, the losses in the semiconductor devices increase and the ZVS conditions of the
converter are lost, thus causing a reduction in the energy efficiency.
At higher switching frequencies the stress on the semiconductor devices increases.
This results in increased switching losses for the MOSFETs and the rectifier diode and thus
causes a reduction in the energy efficiency of the converter.
It is obvious from Figure 11 that as the switching frequency increases beyond
2.8 MHz the energy efficiency significantly reduces. The maximum energy efficiency of
the converter is obtained between 2.4 MHz and 2.8 MHz. Below 2.4 MHz the losses in the
transformer are dominant, while above 2.8 MHz the switching losses in the MOSFETs are
dominant.
60 80 100 120 140 160 18065
70
75
80
VIn
(Volts)
Convert
er
Effic
iency,η
(%
)
RL=10 Ohm
RL=20 Ohm
RL =40 Ohm
Pout=40W
Pout=26W
16
2.2 2.4 2.6 2.8 3 3.265
70
75
80
Frequency (MHz)
Co
nvert
er
Effic
iency,η
(%
)
Figure 11 Energy efficiency as a function of switching frequency
2.6 LOSS ESTIMATION OF HALF BRIDGE CONVERTER
From the measurement results presented in section 2.5, it can be observed that as the
switching frequency and output power of the converter increases, the energy efficiency
degrades due to power losses. The losses are estimated at a 120 V input voltage, 2.5 MHz
switching frequency, 30% duty cycle and 10 Ω load resistance. Under these conditions, the
input power of the converter is 27 W while, the output power is 22W and thus the total
power loss of the converter is about 5 W. These losses in the converter are shared by the
MOSFETs, gate driver, power transformer and rectifier diode. The total estimated loss of
the power transformer is approximately 2W [25]. The remaining 3W power is lost due to
the MOSFETs and the rectifier diode.
MOSFET Losses: There are two categories of losses in the MOSFETs i.e. the
conduction loss and the switching loss. The conduction losses in the power MOSFET can
be calculated using the on-state resistance (RDSon) of the MOSFET. The required parameters
for the conduction loss are RDSon rms current, which flows through the device, and the duty
factor. These values are not difficult to quantify and thus the instantaneous value of the
MOSFET conduction losses can be computed according to equation 2.5 [26].
)(2tIRIVP DDSonDDSconduction ×=×= 2.5
17
As shown in equation 2.6 [26], the average value of the MOSFET conduction losses
can be computed by the integration of the instantaneous power losses over the switching
cycle.
22
00
_ ))((1
)(1
DrmsDSonD
T
DSon
SW
T
Con
SW
avgcon IRdttIRT
dttPT
PSWSW
×=×== ∫∫ 2.6
The RDSon for CoolMOS SPU02N60S5 is 3 Ω and IDrms is 0.3 A.
According to equation 2.6 the average conduction loss of each MOSFET is 0.4 W.
Switching losses occur as a result of a simultaneous exposure of a MOSFET to high
voltage and current during a transition between the open and closed states and is usually
difficult to predict. It is difficult to model the complex switching behavior and switching
loss of a power MOSFET analytically due to the non-linear characteristics of the MOSFET
parasitic capacitances. At higher switching frequencies, the exact estimation of switching is
difficult. The impact of the parasitic capacitance and stray inductance also influences the
switching losses of the converter.
Since the converter is not fully operated under ZVS conditions, it thus becomes
inevitable that there will be a switching. The switching loss of the MOSFETs is estimated
to be approximately 1W according to equation 2.7 [27].
SWoffonDSDsw fttVIP ⋅+⋅= )(2
1 2.7
where:
ID is the drain current
VDS is the Drain-Source voltage
fsw is the switching frequency
ton is the turn-on switching time
toff is the turn-off switching time
The remaining 1.2W loss is shared by the rectifier diode and other circuit components.
From the loss estimations, it is obvious that the MOSFETs are the prominent source of loss
within the converter. The temperature profile of the MOSFETs measured using an IR
thermal camera is given in Figure 12
18
Figure 12 Thermal profile of MOSFETS
Figure 13 Thermal profile of rectifier diode
Diode Lossess: The output rectifier also contributes to the losses of the converter.
Although the Schottky diodes have the lowest forward voltage drop, when this is compared
to a normal P-N junction diode, its forward voltage increases more quickly with a higher
current. An analysis of the switching loss of the rectifier diode is complicated. The thermal
profile of SR-1660 Schottky diode is shown in Figure 13.
19
3 FEED-BACK TECHNIQUES IMPLEMENTED IN HALF
BRIDGE CONVERTER
Almost all power supplies use a feedback loop to achieve a constant DC output
voltage. The feedback voltage is applied to the controller, which compares the output
voltage with a reference voltage and executes a control algorithm. This control algorithm is
used to generate the required PWM signals to drive the MOSFETs. The design of the
feedback network is one of the most important aspects of a converter’s design. The
feedback design is slightly more sophisticated mathematically than other aspects of the
power supply design [28]. To design an isolated feedback stage is more challenging than
designing an isolated power stage. In a feedback loop, the galvanic isolation is required
between the input and output. Various feedback techniques are used in the implementation
of an isolated feedback control.
Standard phototransistor opto-couplers are one of the various techniques used to
implement an isolated feedback [29], [30]. Although it provides the required isolation, there
are several disadvantages when implementing an opto-coupler feedback, such as a variable
loop gain due to opto-coupler tolerance and sensitivity to temperature, as well as its
relatively high cost.
The other alternatives to that of an opto-coupler feedback are isolated magnetic
feedback techniques. The magnetic feedback has good temperature stability and is
insensitive to component tolerance [31]. The magnetic feedback implementation on the
secondary side of the converter is described in [28]. It provides true high voltage isolation
and can be made insensitive to temperature variations.
The auxiliary voltage feedback technique is another alternative to the opto-coupler
feedback. It provides a technique to control the output voltage without the need of a
feedback circuit on the secondary side of the power supply. The auxiliary voltage feedback
has certain advantages as compared to the opto-coupler feedback, such as a higher power
density, lower cost and a lower standby power [32].
3.1 DESCRIPTION OF FEEDBACK CIRCUIT
Two feedback techniques are analyzed in the high frequency half bridge converter
circuit design. In the first technique an opto-coupler is used on the secondary side and the
isolated feed-back signals are analyzed. The variation of the opto-coupler feedback voltage
with the variation in the output voltage is measured for different loads.
In the second technique, the feasibility of a feedback signal using the auxiliary
winding of a coreless PCB power transformer is discussed. The variation of the auxiliary
voltage with the variation of the output voltage for different loads is analyzed.
3.1.1 Opto-coupler feedback
The SMPS require an isolated output voltage control method. The opto-couplers are
commonly used to provide an isolated feedback voltage in SMPS [29]. They consist of a
20
photo-diode that radiates energy proportional to the current flowing through it, and a
phototransistor that uses this energy to cause a linearly related collector current. Opto-
couplers also offer a high degree of noise rejection or isolation combined with their
insulation characteristics [33]. The important parameter for opto-couplers is their transfer
efficiency, usually measured in terms of their current transfer ratio (CTR). This is the ratio
between a current change in the output transistor and the current change in the input LED
which produced it. A diagram of an opto-coupler is shown in Figure 14.
Figure 14 Opto-coupler diagram
The CTR, which is the ratio of the phototransistor collector current to the LED
forward current is measured according to equation 3.1.
F
C
I
ICTR = 3.1
The opto-coupler feedback technique is implemented in a high frequency half
bridge converter. An IL207A opto-coupler is used to provide an isolated feedback
voltage Vopto. The IL207A opto-couplers are optically coupled pairs with a Gallium
Arsenide infrared LED and a silicon NPN phototransistor. They are capable of
transmitting signal information, including a DC level, by maintaining a high
degree of isolation between the input and output [34]. The isolated feedback
voltage Vopto is measured at the ADC input of the microcontroller. The variation
output voltage Vout affects the feedback voltage Vopto from the secondary side of the
converter. A schematic diagram of the converter with an opto-coupler feedback is
shown in Figure 15.
IF IC
21
ADC Module
Opto-coupler
RL
M2
L_lks
L_lks
Lms
MicrocontrollerMOSFET
DriverP
WM
sig
na
ls
Lms
Lmp
Cps
L_lkpC1
C2
M1
D1 L
C
IsolationTransformer
VinD2
Vopto
Figure 15 Schematic diagram of half bridge converter using opto-coupler feedback
It is observed that the feedback voltage taken from the opto-coupler varies in
proportion to the variation of the output voltage. For a constant output voltage, there is no
deviation with the variation of the load resistor. The opto-coupler feedback voltage varies
linearly with the variation output voltage as shown in Figure 16.
15 16 17 18 19 20 21 221
1.2
1.4
1.6
1.8
2
2.2
VOut
(Volts)
Opto
-Couple
r fe
edback (V
olts)
RL=20 Ohm
RL=40 Ohm
RL=90 Ohm
No Load
Figure 16 Variation of Vopto with variation of Vout for different loads
3.1.2 Auxiliary feedback
The auxiliary feedback eliminates the need for an opto-coupler on the secondary side
and also reduces the device count and the size of the power supply. In strongly coupled
transformers the auxiliary voltage is proportional to the output voltage. The variation of the
output voltage with varying loads will result in a proportional variation in the voltage
22
produced by the auxiliary winding [35]. This technique can be used to regulate the output
voltage without requiring a secondary feedback in the converter.
The PCB power transformer used in the half bridge converter consists of two number
of turns of auxiliary windings in two layers. The inductance of the auxiliary winding is
0.193uH with a DC resistance of 0.42 ohms. The auxiliary winding of the power
transformer is strongly coupled with its primary winding.
The voltage taken across the auxiliary winding of transformer is rectified, filtered and
applied to a voltage divider in order to obtain the desired value. The schematic diagram of a
half bridge converter with auxiliary feedback is shown in Figure 17. The resistors R1 and
R2 are used to scale down the auxiliary voltage to the voltage levels of the controller ADC
input. The voltage across R2 is measured as an auxiliary feedback voltage.
The shunt resistor Rshunt is used in series with the lower MOSFET so as to measure the
input current. The input current in combination with the auxiliary feedback voltage is
required to accurately predict the output voltage of the converter.
ADC Module
RC Filter
I_in
D2
Laux
Vin
Isolation
Transformer
C
L
D2
M1
C2
C1 L_lkp
Cps
Lmp
Lms
PW
M s
ign
als
Vaux
MOSFET
DriverMicrocontroller
Lms
L_lks
L_lks
M2
RL
Rshunt
D
R1
R2
C
R
C
Figure 17 Schematic diagram of half bridge converter using auxiliary feedback
In the auxiliary feedback design, the variation of feed-back voltage Vaux is measured
with the variation of the following circuit parameters:
• Fixed value of output voltage Vout and different values of load resistors.
• Variation of output voltage Vout and constant load resister value.
23
It is observed that at a constant output voltage Vout, the auxiliary voltage Vaux varies
almost linearly with the variation of the load resistors. It also varies with the variation
of the output voltage for constant load resistors. The variation of the auxiliary feed-
back voltage with the variation of the output voltage for different values of load
resistors is shown in Figure 18.
16 17 18 19 20 21 221
1.5
2
2.5
3
VOut
(Volts)
Au
x V
oltage
(Vo
lts)
RL=20 Ohm
RL=40 Ohm
RL=90 Ohm
NO LOAD
Figure 18 Variation of Vaux with Vout for different loads
At a constant output voltage the variation of Vaux with the variation in output load is
due to a change of primary current through the power transformer. The primary winding of
transformer is coupled with the auxiliary winding. Therefore any change in the load resistor
will change the input current through the primary winding and hence the auxiliary voltage
varies. In Figure 18 it is obvious that the auxiliary feedback voltage changes around 25%
for a given output voltage at different loading conditions. With this variation, it is not
possible to sense the output voltage correctly and this thus creates uncertainty in a closed
loop system.
In order to correctly sense the output voltage, the input current must be measured. As
shown in Figure 17 a shunt resistor Rshunt is used in series with the lower MOSFET to
measure the input current. The variation of input current is measured under the following
conditions.
• Fixed value of output voltage Vout and different values of load resistors.
• Variation of output voltage Vout and constant load resister value.
The measurement results are plotted in Figure 19. It is inferred that at constant output
voltage, the input current varies almost linearly with the variation of the load resistors. For
a fixed Vout = 20V, the value of the input current is different for different load resistors
(20Ω, 40Ω and 90Ω). For a constant value of the load resistor, the variation of the input
current is linear with the variation of output voltage.
24
16 18 20 220
0.05
0.1
0.15
0.2
0.25
0.3
VOut
(Volts)
Input C
urr
ent (
Am
pe
re)
RL=20 Ohm
RL=40 Ohm
RL=90 Ohm
NO LOAD
Figure 19 Variation of input current with Vout for different loads
From the measurement results of Figure 18 and Figure 19 it can be observed that in
order to correctly sense the output voltage, a linear expression can be established between
the input current and the auxiliary feedback voltage. The linear expression is given in
Equation 3.2.
inauxout IBVAV ×+×= 3.2
where A and B are constants and are selected to provide a constant value at a
specified output voltage.
The output voltage Vout is calculated according to equation 3.2 and the results are
plotted in Figure 20. The dotted line in Figure 20 shows the ideally calculated value of Vout
as a function of the actually measured value of Vout. However, it can be observed that with
the assistance of Equation 3.2, the calculated output voltage is nearly independent of the
load resistors at a specific output voltage (20V in this case). The equation gives the worst
result under no load conditions, but this is still less than a 4% error.
14 16 18 20 22 2415
16
17
18
19
20
21
22
23
Measured VOut (Volts)
Ca
lcu
late
d V
Ou
t (V
olts)
RL=20 Ohm
RL=40 Ohm
RL=90 Ohm
NO LOAD
Ideal calculated Vout
Figure 20 Measured and calculated values of Vout
25
4 FULL BRIDGE CONVERTER DESIGN
Full-bridge converters are mainly used in high-power applications such as motor
drives, uninterruptible power supplies and switch mode transformer isolated power
supplies. The schematic diagram of a full bridge converter is shown in Figure 21.
During the first half cycle, transistors S1 and S4 simultaneously turn on and off while
in the second half cycle, transistors S2 and S3 simultaneously turn on and off. If both
switches of one side (i.e S1 and S2) are simultaneously switched off then an output current
will continuously flow, which will create non-linearity in the relationship between the
control voltage and the average output voltage [36]. A short-through occurs when both the
transistors of one leg (S1 and S2) are switched on simultaneously.
The voltage stress on the power switches is limited to the input voltage source value
and the full input voltage appears across the primary winding. On the secondary side of the
transformer, the voltage is stepped-up or stepped-down, rectified and filtered in order to
produce a DC output voltage.
Figure 21 Full Bridge converter[5]
The operation of a full bridge is similar to that of the half bridge under steady-state
conditions [5]. The output voltage Vout is directly related to the on state time Ton of the
transistor. The relation is derived by integrating the inductor voltage over a time period Ts.
The average value of Vout is given in Equation 4.2 [5].
−+−== ∫∫∫
+ ss
s
on DTT
T
out
T
in
p
s
s
Ts
out
s
out dtVdtVoutVN
N
TdttV
TV
2
200
)(2
)(1
4.1
26
DV
N
NV in
p
sout 2=
4.2
where
s
on
T
TD =
Since the full bridge converter consists of four power switches, four drivers are thus
required to drive the power switches. The design of a full bridge is more complex than that
of a half bridge converter.
4.1 SIMULATION MODEL
The simulation design of the high frequency full bridge DC to DC converter with the
high frequency circuit model of the coreless PCB step-up transformer is shown in Figure
22. The transformer primary windings are connected to the full bridge converter output,
while the center tapped secondary windings of the transformer are connected to a full wave
rectifier circuit. The converter circuit is simulated in the frequency range from 1MHz to
4MHz, and the energy efficiency of the open loop converter is calculated. At 2.5 MHz
switching frequency, 9V input voltage 35% duty cycle and 160 Ω load resistor, the energy
efficiency obtained in simulation is approximately 82%.
Figure 22 Schematic diagram of full bridge converter
4.2 HARDWARE IMPLEMENTATION AND MEASUREMENT RESULTS
A full bridge step-up converter circuit is implemented on a PCB. A
DSPIC33fJ016GS502 microcontroller is used in the converter in order to generate PWM
signals that are fed to a MOSFET driver. Since the MOSFETs are operated at low voltage,
no isolation is thus used for the high side MOSFETs and the gate signals are directly fed to
all MOSFETs. The FDS8958 dual N- and P-Channel enhancement mode power FETs are
used in the design. They have a 30V Drain-Source Voltage and a 2Watt Power dissipation
27
for dual operation [37]. On the secondary side of the power transformer, a Schottky
rectifier diode “20CWT10FN” is used. It has a reverse voltage blocking capability of 100V
and a forward rectified current capability of 10A.
The energy efficiency of the unregulated full bridge converter circuit is computed for
the input DC voltage of up-to 9 V, 35% duty cycle and switching frequency in the range of
1 to 4 MHz, with the load resistances of 160 Ω. The maximum power achieved in this
design is 5.39W. The variation of energy efficiency with the variation switching frequency
is shown in Figure 23. It can be observed that as the switching frequency increases beyond
2.4 MHz, the switching losses increase and the energy efficiency degrades.
Figure 23 Variation of energy efficiency with the variation of switching frequency
4.3 PARAMETERS OF PCB TRANSFORMER USED IN FULL BRIDGE CONVERTER
A coreless PCB center-tapped step-up power transformer with a turns ratio of 1:4
designed at Mid Sweden University is used for the full bridge converter. It can be operated
in the 1 to 4 MHz frequency region with energy efficiency greater than 90%. The outer
diameter of the PCB power transformer is 36mm and the distance between the two
consecutive layers of the PCB is 0.4mm. The transformer structure has two secondary
windings in the first and the fourth layers while there are two primary windings in the
second and third layers for achieving a better coupling between the primary and secondary
windings. The secondary windings are connected in series and the primary windings are
connected in parallel to achieve the desired inductance. The parameters of the step-up
transformer used in the full bridge converter are as follows:
• The self-inductances of the primary and secondary winding are 1.275µH and
10.45µH respectively.
• The leakage inductances of the primary and secondary winding are 0.326µH and
2.86µH respectively.
28
• The AC resistance of primary and secondary winding of the transformer are
0.27Ω and 2.3Ω respectively.
• The inter-winding capacitance is approximately 101pF.
The energy efficiency with a sinusoidal excitation of the transformer is shown in Figure 24
Figure 24 Energy efficiency of the coreless PCB power transformer
4.4 DESIGN LIMITATIONS
The total losses of the full bridge converter are shared by the MOSFETs, transformer
and diodes. The coreless PCB transformer in the converter design has shown major
contribution in attaining both a high energy efficiency and compactness. At higher
switching frequencies, the switching loss in the power MOSFET becomes significant. This
places a limit in attaining high power levels and a high efficiency in the converter design.
The temperature profile of the power MOSFETs at a power level of 5W, recorded using an
IR thermal camera, is shown in Figure 25.
Figure 25 Thermal profile of MOSFETs
29
5 THESIS SUMMARY AND CONCLUSIONS
Chapter 1: The basic introduction of linear and switch mode power supplies is given. The
advantages and disadvantages of both the power supplies are discussed. The motivating
factors for a high frequency SMPS design, such as the high frequency power MOSFETs,
PCB transformers and smaller EMI filters are discussed. It can stated that, with the
improvement in power MOSFET technology and the development of high frequency
multilayered PCB power transformers, it is now possible to design high frequency and
power efficient isolated converters. Although the size of SMPS can be decreased by
increasing the switching frequencies but there are certain challenges in the design of high
frequency switching converters. These challenges are discussed in this chapter.
Chapter 2: In this chapter the design of a half bridge DC-DC converter is discussed. The
theoretical background and the basic operating principle of a half bridge converter circuit
and its transformer design are discussed. A simulation model and the hardware
implementation of a high frequency half bridge converter are also presented. The PCB step
down power transformer is used in the converter. A PCB isolation transformer is also used
for providing the isolated gate signals for the high side MOSFETs. The parameters and
energy efficiency of both transformers are given. The maximum energy efficiency of the
PCB step down power transformer used in the design is 96% at a switching frequency of
2.6 MHz. The energy efficiency of the converter of the converter is measured up to a 40W
output power. The converter is switched up to 3 MHz switching frequencies. The maximum
energy efficiency is approximately 82%. This work is a preliminary step in the design of
forward converters in the MHz frequency region and the losses in a converter are estimated.
Chapter 3: Two feedback techniques are implemented in the half bridge converter which
was presented in chapter 2. Measurement results for both the feedback techniques are
analysed. It is observed that, although the output voltage can be predicted with the use of an
opto-coupler feedback, this opto-coupler feedback is temperature sensitive and is relatively
expensive. The feasibility of a feedback signal, using the auxiliary winding of a coreless
PCB power transformer, is discussed. It is observed that in order to accurately predict the
output voltage with auxiliary feedback, the primary current must be measured in
combination with the auxiliary voltage. Both the auxiliary feedback voltage and the input
current vary linearly with the variation of the output voltage. Therefore, a linear
mathematical expression is established between the auxiliary feedback voltage and the
input current in order to calculate the output voltage based on the variation of the input
current and the auxiliary voltage.
Chapter 4: In this chapter the design of a full bridge DC-DC converter is discussed. A
brief theoretical background and the basic operating principle of a full bridge converter
circuit and its transformer design are discussed. A simulation model and the hardware
implementation of the converter are described. The measurement and simulation results of
the converter are presented. The maximum energy efficiency of 82% has been achieved in
30
the simulation circuit while, in the actual hardware circuit a maximum energy efficiency of
72% has been achieved. A coreless PCB step up power transformer designed at Mid
Sweden University is used in the implementation of the converter circuit. In switching the
frequency range of 1-4 MHz, a maximum energy efficiency of 90% has been achieved for
the power transformer. The limitations of the converter are discussed.
5.1 FUTURE WORK
The EMI is a very important issue for almost all electronic equipments. The increased
switching frequency, in combination with sudden changes in the current (di/dt) or
voltage(dv/dt) levels generates higher order harmonics, which causes EMI and affects the
Electro Magnetic Compatibility (EMC) of the SMPS.
The EMC design is a very important requirement for SMPS. Therefore it is essential
to limit the EMI so that the product is developed according to EMC standards. To achieve
this, there are many approches such as filtering, shielding and isolation etc. Future work
will be focused on the design of one or two stage EMI filters for DM and CM noise
attenuation in high frequency power supplies. The filters will be characterized and their
signal behaviour will be studied.
31
6 PAPER SUMMARY
6.1 HIGH FREQUENCY HALF-BRIDGE CONVERTER USING MULTILAYERED
CORELESS PRINTED CIRCUIT BOARD STEP-DOWN POWER TRANSFORMER
A compact and high density half bridge DC-DC converter is designed and tested by the
implementation of the circuit on a printed circuit board. The converter is operated in the
MHz frequency region with a maximum output power of 40 W. The maximum energy of
the unregulated converter was approximately 82 %. The maximum energy efficiency as a
function of input voltage, duty cycle and switching frequency was computed. Multilayered
coreless PCB transformers used in the converter circuit have provided a major contribution
in both the energy efficiency and its compactness. It is observed that in the future, it will be
possible to design very small and compact switch mode power supplies once better
switching devices and techniques have been developed.
6.2 ANALYSIS OF FEEDBACK IN CONVERTER USING CORELESS PRINTED
CIRCUIT BOARD TRANSFORMER
In this paper, the feasibility of a feedback signal, using the auxiliary winding of a
coreless PCB power transformer, is discussed. Both opto-coupler and auxiliary feedback
voltages are implemented, measured and analyzed. Although the opto-coupler feedback
voltage remains constant for a given output voltage for different loads, the implementation
of opto-coupler feedback remains temperature sensitive and has a relatively high cost.
To accurately sense the output voltage with auxiliary feedback, the primary current is
also measured in combination with the auxiliary voltage. A mathematical expression is
established to calculate the output voltage with the variation of the input current and the
auxiliary voltage. This expression provides good results for different loads, while, under no
load conditions, it offers an error less of than 4%, which is quite acceptable in many cases.
It is observed from the analysis that the auxiliary feed-back can be used as a feed-back
control loop for a high frequency half-bridge DC-DC converter as a better alternate than the
opto-coupler feedback because it is simple to implement, insensitive to temperature
variations and is cost effective.
6.3 HIGH FREQUENCY FULL BRIDGE CONVERTER USING MULTILAYER
CORELESS PRINTED CIRCUIT BOARD STEP UP POWER TRANSFORMER
PLATFORM
In this paper the design of a full bridge DC to DC converter using a multilayer coreless
PCB step-up power transformer is discussed. The design of the converter is tested up to 5 to
6Watts. The measured energy efficiency of the unregulated converter is 75% at 2.4MHz
frequency. The major loss contribution in the design was due to the switches. This places a
limit on attaining high power levels and high efficiency in such a converter design. The
coreless PCB transformer in the converter design has provided a major contribution in
attaining high energy efficiency and compactness. Better switching devices with high
32
power dissipation level in the required voltage/current range can improve both the power
level and the efficiency of the converter.
33
6.4 AUTHORS’ CONTRIBUTIONS
The exact contributions of the authors of the three central papers in this thesis are
summarized in Table 6-1.
Table 6-1.Authors’ Contributions
Paper # Main author
Co-authors Contributions
I AM
HK
JS
RA
SH
KB
AM : Literature study, design, implementation and
detailed analysis
HK : Design of gate drive circuitry, discussion and
review of paper.
JS : Help in implementation, discussion and review.
RA : Power and signal transformer design,
secondary inductor design and parameters extraction
and transformer characterisation.
KB: Supervision of the implementation work and
final review of paper
II AM
JS
HK
RA
SH
KB
AM: Literature study, design, implementation and
detailed analysis
JS: Help in measurements, discussion and review.
HK: Discussion and review of paper.
RA: Transformer design, discussion and review of
paper.
SH: Help in design of auxiliary Feedback Circuit
KB: Supervision of the implementation work and
final review of paper
III JS
AM
RA
HK
KB
JS : Literature study, Design, implementation and
detailed analysis.
AM: Testing , analysis, discussion and review.
RA: Power transformer design, secondary inductor
design and parameters extraction.
HK: Discussion and review of paper
KB: Supervision of the implementation work and
final review of paper
1. Abdul Majid (AM)
2. Kent Bertilsson (KB)
3. Jawad Saleem (JS)
4. Hari Babu Kotte (HK)
5. Radhika Ambatipudi (RA)
6. Stefan Haller (SH)
34
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