Bandpass Filtering Doherty Power Amplifier With Enhanced ...static.tongtianta.site/paper_pdf/4847536e-7aee-11e... · systems have become another important issue. Therefore, the efﬁciency

IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—I: REGULAR PAPERS, VOL. 63, NO. 3, MARCH 2016 337

Bandpass Filtering Doherty Power Amplifier WithEnhanced Efficiency and Wideband

Harmonic SuppressionShao Yong Zheng, Member, IEEE, Zhao Wu Liu, Yong Mei Pan, Member, IEEE,

Yongle Wu, Wing Shing Chan, Member, IEEE, and Yuanan Liu

Abstract—There is an ever increasing demand for both functionintegration and improved performance in wireless communicationdevices. Therefore, the integration of efficient linearizing ampli-fying and bandpass filtering functions is proposed for the firsttime in this paper. A new Doherty amplifier configuration withattractive behaviors including bandpass filtering, wideband har-monic suppression, and enhanced efficiency, is presented. A com-ponent crucial to the successful implementation of the proposedconfiguration is the bandpass quadrature coupler with arbitrarycoupling coefficient. Owing to the flexibility of a circular patchin locating the frequencies of different operational modes, it isthus possible to implement good bandpass characteristics. Withthe introduction of this new component, the Main and Auxiliaryamplifiers can be driven more efficiently, resulting in both lin-earity and efficiency improvements. To verify the validity of theproposed configuration, a Doherty amplifier has been designedand fabricated. Its measured performance is compared with theconventional design and found to exhibit a bandpass filteringcharacteristics with a wide suppression band up to the thirdharmonic frequency. Meanwhile, the proposed amplifier achievesa measured power-added-efficiency improvement of 10% and anACLR improvement better than 5.8 dB at the higher outputpower region.

Index Terms—Arbitrary coupling coefficient, bandpassfiltering, circular patch, Doherty amplifier, efficiency, harmonicsuppression.

Manuscript received July 27, 2015; revised November 29, 2015; acceptedDecember 21, 2015. Date of publication March 22, 2016; date of currentversion April 5, 2016. The work described in this paper was supportedby the National Natural Science Foundation of China (No. 61401523 andNo. 61422103), Foundation for Distinguished Young Talents in Higher Edu-cation of Guangdong, China (No. 2014KQNCX002) and National Key BasicResearch Program of China (973 Program) (No. 2014CB339900). This paperwas recommended by Associate Editor R. Gomez-Garcia.

S. Y. Zheng and Z. W. Liu are with the Department of Electronics and Com-munication Engineering, Sun Yat-sen University, Guangzhou 100044, China,and also with the SYSU-CMU Shunde International Joint Research Institute,Shunde 528300, China (e-mail: [email protected]).

Y. M. Pan is with the School of Electronic and Information Engineering,South China University of Technology, Guangzhou 510641, China.

Y. Wu and Y. Liu are with the School of Electronic Engineering, BeijingUniversity of Posts and Telecommunications, Beijing 100876, China.

W. S. Chan is with the Department of Electronic Engineering, City Universityof Hong Kong, Kowloon, Hong Kong.

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TCSI.2016.2515419

I. INTRODUCTION

W ITH the need to conserve energy and reduce environ-mental pollution, green communications has become a

major requirement in the development of industry. As the mainsource of energy consumption in a communication system,power amplifiers (PAs) contributes 40%–60% of the energyusage in the base station and improvements in its efficiencywill help reduce its carbon footprint immensely. In addition, theminiaturization of key components in wireless communicationssystems have become another important issue. Therefore, theefficiency improvement and size reduction are important topicsin the design of PAs.

In order to improve the efficiency of PAs, different tech-niques have been proposed, such as envelope tracking, dynamicload modulation and so on [1], [2]. Moreover, the coexistence ofmultiple standards requires future front ends to simultaneouslyhandle signals with different modulation formats, which canlead to a combined waveform with high peak-to-average powerratio (PAPR). Thus a Doherty power amplifier (DPA), whichwas originally proposed in [3], has been rediscovered andwidely adopted for high-PAPR signal amplification.

DPAs are simple, easy to implement and relatively widebandwhen compared to other efficiency enhancement techniques.Two amplifiers are used in the basic DPA topology, compris-ing the Main and Auxiliary PA. With deliberately control ofthe Auxiliary PA, the combined output load is dynamicallymodulated to keep the Main PA working at peak efficiencyover a wide range of output back-off (OBO). However, thelimited bandwidth observed in practical DPA runs counterto the requirement for a broader operating bandwidth. Tosolve this problem, several techniques have been proposedfor bandwidth enhancement [4], dual band operation [5], andtunable operating frequency [6], [7]. However, the optimumperformance cannot be achieved due to several factors thatinclude low gain of the Auxiliary PA and parasitic effects.Several design techniques such as adaptive bias [8], switchedDoherty [9], and digital predistortion [10] have been proposedfor the improvement in performance of the DPAs. However,these methods employ extra complicated circuitry, which addsto their size and cost. To overcome this problem, an unevenpower divider is employed at the input of the DPA for betterload modulation [11]. As an extension, the adaptive input-power distribution can simultaneously provide the enhancement

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338 IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—I: REGULAR PAPERS, VOL. 63, NO. 3, MARCH 2016

Fig. 1. Topology for the PA and bandpass filter in the RF front end.

in linearity and efficiency [12]. Alternatively, a transformercan be utilized for output combining to improve the efficiency[13]. The complex combining load was proposed to extend thehigh-efficiency range of the DPA [14]. The miniaturization ofDPAs can be achieved either by compressing the size of theindividual components, or by integrating different functionswithin a single component. For a differential DPA, the numberof transistors can be reduced to three for a compact size[15]. In [16], the impedance inverting network and offset linesrequired in the conventional topology can be eliminated forsize reduction. Alternatively, if the related functionalities can beimplemented within a single configuration, the integration levelwill increase significantly. In order to minimize interferencefrom adjacent bands and unwanted noise, a bandpass filter(BPF) is usually placed before or after the power amplifiers(PAs). This is usually implemented based on the cascadetopology shown in Fig. 1. This will however increase theinsertion loss, resulting in the deterioration in overall systemperformance. To overcome this problem, a co-design approachwas proposed to match the filter’s input port directly to the tran-sistor with output matching network entirely eliminated [17].Alternatively, the functionality of the BPF can be integratedtogether with output matching network. A microstrip line BPFwas utilized as the quarter wave length impedance transformerto achieve the desired maximum power point and bandpassfiltering characteristics [18]. The microstrip line BPF can alsobe integrated with the output matching network of a class Famplifier for the reduction of insertion loss and circuit size[19]. Out of band interference up to harmonic frequenciesshould also be suppressed in a multi-standard wireless com-munication system, which has been implemented for differentcomponents [20]–[22]. In addition, the control of harmonicat the input port can be utilized to achieve the enhancementin linearity [23]. Therefore, it is also necessary to combineharmonic suppression functionality within the DPA. How-ever, these approaches cannot be applied directly to the DPAfor the desired function integration and further performanceenhancement.

In this paper, a novel Doherty amplifier configuration whichexhibits enhanced efficiency, bandpass filtering, and widebandharmonic suppression is proposed. It is realized with the useof a bandpass quadrature coupler with arbitrary coupling co-efficient. This paper is organized as follows. Section II givesthe theoretical study in the performance enhancement of theDPA. In Section III, the snowflake shaped patch is proposedto implement a quadrature coupler with different couplingcoefficients and excellent filtering function integrated. Forexperimental validation, the measured performance of DPAprototypes (conventional and proposed) is given in Section IV.Finally, a conclusion is drawn in Section V.

Fig. 2. The block diagram of conventional DPA.

Fig. 3. The schematic of the Doherty amplifier.

II. THE CONVENTIONAL DOHERTY AMPLIFIER

The conventional DPA is comprised of Main and Auxiliaryamplifiers, power splitter, input and output matching networks,offset lines, impedance inverter, and an additional phase com-pensation network, as shown in Fig. 2. The Main amplifier isbiased in class AB or B, while the Auxiliary amplifier is biasedin class C. The impedance inverter is usually implementedusing a quarter-wave transmission line. Therefore, an additionaldelay line should be introduced after the input power splitter forphase compensation. The operational diagram used to analyzethe DPA is shown in Fig. 3. Two current sources which areassumed to be linearly proportional to the input voltage signalare connected in parallel to the load impedance ZL.

When the DPA operates in the low-power region, theAuxiliary amplifier remains in the cut-off state. Through theimpedance inverter modulation, the Main amplifier load im-pedance is two times that of the optimum load for maximumoutput power. When the input power increases, the Dohertypower amplifier efficiency will reach its first maxima. Then,the Auxiliary amplifier will turn on. When the Auxiliary am-plifier output current IA increases to the value of the Mainamplifier output current IM , the Main amplifier load impedancewill decrease to 2ZL, and the efficiency will reach its secondmaxima. However, the bias voltage of the Auxiliary amplifieris lower than that of the Main amplifier, IA,max does not equalsto IM,max if an equal power divider is utilized at the input. Theload impedance of both amplifiers may not be fully modulatedto the value of the optimum impedance, and therefore results inperformance degradation.

Based on Fig. 3, the relationship between the voltage VL, thecurrents IM and IA, and load impedance ZL can be obtainedusing Kirchhoff law:

VL = ZL(IM + IA). (1)

ZHENG et al.: BANDPASS FILTERING DOHERTY POWER AMPLIFIER WITH ENHANCED EFFICIENCY AND WIDEBAND HARMONIC SUPPRESSION 339

Fig. 4. The drain currents of MW6S004NT1 under Class AB and Class C.

When the Auxiliary amplifier turns on, the load impedances ofthe two amplifiers are obtained as:

ZM =Z2T

ZL

(1 + IA

IM

) (2)

ZA =ZL

(1 +

IMIA

). (3)

It can be found that the Main amplifier load impedance ZM

is modulated by the Auxiliary amplifier output current IA. Asthe gate voltage of the Auxiliary amplifier is lower than that ofthe Main amplifier, if the input power is equally split betweentwo amplifiers, IA,max should be smaller than IM,max.

For verification, the drain current of Freescale MOSFETMW6S004NT1 under different modes is extracted from themodel. The corresponding current value with varied inputpower levels are shown in Fig. 4. The quiescent bias currentof the Main amplifier (Class AB) is 50 mA, while the Auxiliaryamplifier (Class C) is biased at cut-off. It can be observed thatthe desired equal currents cannot be achieved using an equalpower divider, which verifies the previous investigation.

According to the previous analysis, an unequal power dividershould be used in the DPA to increase the Auxiliary amplifiermaximum output current IA,max for better load modulation.Therefore, the best power division ratio should be determinedto realize the optimum active load modulation. Based on thedefinition of the transconductance (gm), when the DPA reachesits maximum output power level, the gate-source voltages of theMain and Auxiliary amplifiers are obtained as:

Vgs,Main =IM,max − IM,DC

gm,Main(4)

Vgs,Aux =IA,max − IA,DC

gm,Aux. (5)

The above formula is based on an active device with constanttransconductance, where IM,DC and IA,DC are the quiescentbias currents of the Main and Auxiliary amplifiers. According tothe operating principle of DPAs, the Auxiliary amplifier shouldbe biased in class C, thus the corresponding quiescent biascurrent IA,DC is zero. gm,Main and gm,Aux are the transcon-ductances of the Main and Auxiliary amplifiers, respectively.

Fig. 5. The block diagram of DPA cascaded with BPFs.

When the DPA reaches its maximum output power level, thedriving power of the Main and Auxiliary amplifiers are given by:

Pin,Main =1

2

(Vgs,Main)2

Rin,Main(6)

Pin,Aux =1

2

(Vgs,Aux)2

Rin,Aux(7)

where Rin,Main and Rin,Aux are the input resistances of theMain and Auxiliary amplifiers, respectively.

Thus the power division ratio can be determined as:

Pin,Aux : Pin,Main =1

2

(Vgs,Main)2

Rin,Main:1

2

(Vgs,Aux)2

Rin,Aux. (8)

Since the unequal bias voltages of the Main and Auxiliaryamplifiers result in the difference in the maximum current, inputimpedance, and transconductance, different input power levelsare required for these two amplifiers to optimize the active loadimpedance modulation. Thus an unequal power divider can beused for performance enhancement in output power, efficiencyand linearity. Moreover, to reject unwanted signal injecting intothe DPA, additional BPFs should be employed before the inputpower divider or the two amplifiers, as described in Fig. 5. How-ever, this will cause mismatching and additional loss, whichresults in significant performance degradation. To solve theseproblems, a device which can simultaneously provide arbitrarypower division ratio, quadrature phase difference, bandpassfiltering and harmonic suppression characteristics, is necessary.However, there is no configuration reported that can realizesuch a high level integration.

III. BANDPASS QUADRATURE COUPLER WITH ARBITRARY

COUPLING COEFFICIENT AND HARMONIC SUPPRESSION

A. Bandpass Quadrature Coupler Configuration

A quadrature coupler is an important component that canprovide signals with a phase difference of 90◦ at two isolatedports and is used in many wireless communication applications.Other than applications in DPA, the quadrature coupler is com-monly used in mixers, phase shifters, beam forming networks,and antenna arrays. The conventional approach to achieve filter-ing is to insert bandpass filters at the coupler ports, but this re-sults in excessively large size and high insertion loss. Recently,several designs have been proposed to combine the coupler andfilter functions into one configuration. The electromagneticallycoupled resonators have been incorporated within the circuitto achieve this integration [24]. High-impedance transmissionlines have been used to suppress the second harmonic withthe aid of interdigitated shunt capacitors [25]. Lumped elementbandstop resonators have also been employed to implement


Fig. 6. Configuration of the snowflake shaped patch.

a compact coupler with harmonic suppression [26]. However,this approach cannot be directly extended for high frequencyapplications. In addition, unequal power dividing/combiningis required for the implementation of DPA and beamformingnetwork of antenna arrays [27], [28]. However, to the author’sknowledge, there is no quadrature coupler configuration re-ported that simultaneously provides bandpass characteristics,arbitrary coupling coefficient and harmonic suppression, whichis necessary for the proposed configuration.

Fig. 6 shows the structure of a snowflake shaped patchelement, which is a patch loaded with four slots. The slots withwidth of Ws and lengths (Ls1 and Ls2) are loaded on the patchacross the horizontal and vertical planes. The radii and anglesof these circular sectors are given by

R1 =R3

R2 =R4

θ1 = θ2 = θ3 = θ4 = 90◦.

To explain the harmonic suppression characteristics of theproposed patch element, the related analysis that provides es-sential information are shown below.

The resonant frequency of a basic circular patch (R1 = R2 =R3 = R4 = R) operating at TMnm mode can be obtainedusing [29]:

f0 =χmn · v

2πReff

√εeff(R)

. (9)

The constant v is the speed of light in free space, χmn is themth zero of the derivative of Bessel’s function of order n. n isthe angular mode number, and m is the radial number. Theyrepresent the magnetic field variations in the two directions,indicating the specified TM mode propagating in the cavityformed by the patch and ground plane. Reff is the effectiveradius of the patch, which is given by

Reff = R ·[1 +

2h

πRεr

(ln

{πR

2h

}+ 1.7726

)] 12

(10)

and εeff is the effective dielectric constant ratio of the substrate,which is given by

εeff(R) =εr + 1

2+

εr − 1

2

[1 + 12

h

R

]− 12

(11)

where εr is the relative dielectric constant of the substrate.All circuits described in this paper are realized using sub-

strate Rogers RO4003C with a dielectric constant ratio of 3.38and thickness of 0.813 mm.

Fig. 7. Current distributions within in the circular patch. (a) TM11 mode(b) TM21 mode.

To investigate the effects of slots on the operation of the snow-flake shaped patch, the current distributions within the patch un-der different operating modes are obtained using Ansoft HFSS.For ease of analysis, the geometrical parameters for the snow-flake shaped patch are predetermined to be: R1=R2=18 mm.

Four slots with slot width (Ws = 2 mm) are loaded orthogo-nally onto the circular patch (R1 = R2 = 18 mm) as describedin Fig. 6. The current distribution within the circular patchunder different modes is depicted in Fig. 7. For the TM11 mode,the current distribution forms a pattern within the whole circularpatch. For the higher order TM21 mode, the current distributionis repeated in each quadrant of the patch. The slots effectivelyincrease the TM11 and TM21 mode current path lengths,subsequently reducing the resonant frequencies of TM11 andTM21 mode. The resonant frequencies under different modesare calculated for varying slot lengths (Ls), as described inFig. 8. When the slot width (Ls) increases from 4 mm to13 mm, the resonant frequency for TM11 mode decreases from2.5 GHz to 1.76 GHz, and the resonant frequency for TM21

mode decreases from 3.87 GHz to 1.99 GHz. Therefore, minia-turization is achieved for both TM11 and TM21 mode operationof the snowflake shaped patch. More importantly, it can beobserved that the resonant frequency for TM21 mode decreasesfaster than that for TM11 mode. This can be easily explained byobserving the current distributions of the two operation modes.The slots are placed perpendicular to the current lines withinthe patch under TM21 mode, therefore they affect more on theresonant frequency of TM21 mode than that of TM11 mode.This unique property can therefore be employed to implementdifferent devices for size reduction and harmonic suppression,which will be presented in the following section.


Fig. 8. Simulated resonant frequencies for TM11 and TM21 mode of asnowflake shaped patch with different slot length Ls.

Fig. 9. Configuration of the bandpass quadrature patch coupler with widesuppression band.

Fig. 10. Equivalent coupled-resonator network of patch quadrature couplerbased on snowflake shaped patch.

As shown in Fig. 9, the snowflake shaped patch configura-tion is proposed to realize a quadrature coupler structure withfiltering characteristic. Each port is separated by 90◦, whilethe inset feed approach is used at the input/output ports forharmonic suppression. The patch coupler can be transformedinto a coupled-resonator network, as described in Fig. 10.

The relation between adjacent ports can be represented bytwo resonators with coupling coefficient k. The related couplingcoefficients are given by

k12 = k34 =1

Qe

√1− C2

(12)

k23 = k14 =C

Qe

√1− C2

(13)

Fig. 11. Frequency responses of the quadrature couplers based on circularsector patch and snowflake shaped patch. R1 = 18 mm, R2 = 19.2 mm,Ws = 2 mm, and Ls = 14 mm.

where C is the power division ratio |S21| of the proposed band-pass quadrature coupler, Qe is the external quality factor whichequals to the inverse of its operational bandwidth. Therefore,the coupling coefficient (k12) can be determined for a desiredoperational bandwidth and power division ratio. Owing to theflexibility of the proposed patch, the ratio of radii (R1/R2)can be used to control the coupling coefficient k12, as wellas k23.

To investigate the functionality of the proposed snowflakeshaped patch, a comparison between couplers based on circularsector patch and snowflake shaped patch is conducted. Therelated frequency responses for couplers with same dimensionalparameters are shown in Fig. 11. The operating frequency ofthe coupler based on a snowflake shaped patch is found to be1.8 GHz. Its operating frequency is lower than the configurationbased on a circular sector patch [24], indicating a compactsize. Moreover, the coupler based on a snowflake shaped patchcan suppress the second harmonic, which can be found in theresponses of the coupler based on the circular sector patch.This can be verified by observing that the slot length Ls isable to adjust the frequency for TM21 mode approaching tothat for TM11 mode, as shown in Fig. 8. In order to further


Fig. 12. Frequency responses of proposed quadrature coupler based on snow-flake shaped patch without and with inset fed approach. R1 = 18 mm, R2 =19.2 mm, Wc = 0.3 mm, Lc = 8 mm, Ws = 2 mm, and Ls = 14 mm.

suppress the third harmonic, the inset fed approach is in-vestigated. A comparison between couplers with and withoutinset fed approach was also conducted. It can be observed inFig. 12 that the introduction of an inset fed mechanism has abetter harmonic suppression. However, a spur can be observedat around 3.4 GHz within the suppression band. This can beeliminated by introducing different slot lengths (Ls1 and Ls2),which is utilized in the following designs.

To determine the relationship between the coupling coeffi-cient and ratio of R1 and R2 for the proposed snowflake shapedpatch coupler, the related parametric study is conducted. Thecoupling coefficient |S31| versus the ratio is plotted in Fig. 13.It can be observed that, the coupling coefficient |S31| decreasesfrom 3 dB to 10 dB when the ratio is varied from 1.07 to 1.24.The arbitrary coupling coefficient functionality can be realizedwith the proposed configuration by adjusting the ratio withoutcomplex analysis, and results in a simple design procedure.Alternatively, the different slot lengths (Ls1 and Ls2) can beemployed to control the coupling coefficient of the coupler.Therefore, the high flexibility of snowflake shaped patch usedto implement quadrature coupler with integrated functionalitieshas for the first time been found.

Fig. 13. The coupling coefficient |S31| of the bandpass patch quadraturecoupler with different radius ratios (R2/R1).

Fig. 14. Photographs of the designed bandpass patch quadrature couplers.(a) 3 dB. (b) 6 dB.

B. Design Examples and Experimental Results

For demonstration purposes, the design and experimentalresults of two bandpass quadrature couplers with different cou-pling coefficients centered at 1.8 GHz, will be presented in thissection. The circuits are fabricated using Rogers RO4003C witha dielectric constant ratio of 3.38 and thickness of 0.813 mm.Fig. 14 shows photographs of the designed bandpass filteringquadrature couplers. First of all, the snowflake shaped patch isused to implement the miniaturized patch quadrature couplerwith filtering characteristics. Dimensions of the circuit areR1 = R2 = 19 mm, Wc = 0.3 mm, Lc = 8 mm, Ws = 2 mm,Ls1 = 14.7 mm, and Ls2 = 13 mm.

Fig. 15 shows the simulated and measured frequency re-sponses of the designed 3 dB quadrature patch coupler. Mea-sured responses show good agreements with simulated results.The design circuit exhibits quadrature equal power divisionwhile maintaining a bandpass response. Moreover, the widerange of suppression up to the third harmonic achieves a highrejection level of at least 32 dB. The measured coupling coeffi-cient S21 and S31 follow each other closely with amplitude im-balance smaller than 1 dB throughout the measured frequencyrange from 1.64 GHz to 1.9 GHz. The phase difference betweenoutput ports is 90◦ ± 5◦ within a relative bandwidth of 14.7%centered at 1.77 GHz.


Fig. 15. Simulated and measured responses of designed 3 dB coupler withfiltering characteristics.

Owing to the flexibility of the snowflake shaped patch, thecoupler topology is able to provide a 6 dB coupling coefficient,90◦ phase difference, good in band matching and out ofband rejection simultaneously. According to the analysis givenpreviously, the resulting dimensions of the circuit are: R1 =R2 = 19 mm, Wc = 0.35 mm, Lc = 7.7 mm, Ws = 2 mm,Ls1 = 15.6 mm, and Ls2 = 12.7 mm. It can be found that the6 dB coupler occupies the same circuit size as the previous 3 dBcoupler. The variation in coupling coefficient is primarily real-ized by varying the slot lengths (Ls1 and Ls2).

Fig. 16 shows the simulated and measured frequency re-sponses in terms of S-parameters for the designed 6 dBbandpass quadrature patch coupler. Good agreement betweensimulation and measurement can be observed. The couplingcoefficient |S31| is 6 ± 0.6 dB across the 1.58–1.89 GHz band.The measured phase difference between the two output ports is90◦± 5◦ within the frequency band from 1.66 GHz to 1.88 GHz.Moreover, a high rejection level can be found within the sup-pression band up to the third harmonic. It can be concluded thatthe coupling coefficient specified by certain applications can beeasily implemented based on the proposed snowflake shaped

Fig. 16. Simulated and measured responses of designed 6 dB coupler withfiltering characteristics.

TABLE ICOMPARISON BETWEEN PROPOSED PATCH QUADRATURE

COUPLER AND PREVIOUS WORKS

patch. Besides, the bandpass filtering characteristics such as therejection levels attained in the lower attenuated band can befurther improved by introducing different input/output feedingmechanisms on the proposed patch element.

Table I summarizes the comparison between the proposedpatch coupler with other structures that can be found in theliterature. The proposed configuration is found to occupy asmaller size than a conventional patch based quadrature coupler[27]. Moreover, this structure realizes filtering and harmonicsuppression simultaneously, which cannot be implemented in[25]–[27]. Nevertheless, the patch based configuration makesit more suitable for high frequency applications than previ-ous works found in the literature [25], [26], which were im-plemented using either microstrip lines or lumped elements.In addition, the proposed quadrature coupler exhibits highestsimplicity in structure compared to existing structures foundin the literature. Finally, the arbitrary coupling coefficient


Fig. 17. The block diagram of proposed DPA.

characteristic which cannot be implemented in previous works[25], [26], demonstrates the flexibility of proposed snowflakeshaped patch.

IV. PROPOSED BANDPASS DOHERTY AMPLIFIER

WITH HARMONIC SUPPRESSION

A. Proposed DPA Configuration

To minimize interference from adjacent bands, the use ofBPFs is the most straightforward approach. But these BPFsusually introduce insertion loss and additional circuit size,making it less desirable for use in DPA. Since DPA requiresextra power splitter and delay lines, which already result inlarge insertion loss and circuit size, anything additional wouldbe undesirable. The out of band suppression characteristics canbe implemented by increasing the number of filter stages, butthis would result in additional insertion loss and circuit size.Therefore, the integration of bandpass filtering and harmonicsuppression functionalities would be desirable for the enhance-ment of DPA performance. To achieve function integrationand performance enhancement simultaneously, the bandpassquadrature coupler is used as the input power divider in a DPA.The detailed block diagram for the amplifier is shown in Fig. 17.As the quadrature coupler can provide 90◦ phase differencebetween output ports, the delay line for phase compensationnetwork in Fig. 5 can be eliminated for compact size. Moreover,the BPFs in Fig. 5 have also been eliminated.

B. Experimental Results

For experimental validation, two Doherty amplifiers basedon the proposed and conventional configurations are designed,fabricated and measured. Both of them are implemented ona FR4 substrate with a relative dielectric constant ratio of4.4, thickness of 0.8 mm. The transistors used are MOSFETMW6S004NT1 by Freescale. The Main amplifiers for bothcases are biased in Class AB with a drain quiescent currentof approximately 50 mA. According to the analysis given inSections II, the power division ratio for the quadrature couplerrequired to achieve optimum load modulation is 1 : 1.3, indicat-ing a 1.15 dB difference between two output ports. Based onthe coupler structure described in Section III, dimensions of thecoupler are obtained as: R1 = 16.4 mm, R2 = 17.4 mm, Wc =0.25 mm, Lc = 6 mm, Ws = 2.7 mm, and Ls = 11.8 mm.Fig. 18 shows the photographs of conventional DAP and pro-posed bandpass filtering DPA. The proposed DPA occupies acircuit size of 140.1× 68.4 mm2, indicating a comparable sizecompared to the conventional one.

Fig. 18. The photographs of the conventional and proposed DPAs. (a) Conven-tional. (b) Proposed.

Fig. 19. Measured S-parameters of proposed DPA.

Fig. 19 shows the measured small signal S-parameters of thedesigned DPA. Owing to the integration of bandpass filteringfunctionality within the quadrature coupler, the designed DPAexhibits a bandpass filtering responses with harmonic sup-pression up to the third harmonic. Moreover, the rejectionlevel is found to be larger than 27.9 dB and 40.7 dB for thelower and higher suppression band respectively, as indicated inFig. 19. The maximum small signal gain of 14.3 dB is mea-sured at 1.8 GHz. The frequency range for 1-dB bandwidthis 1.72–1.85 GHz. Therefore, additional functionalities includ-ing bandpass filtering and wideband harmonic suppressionhave been integrated within the DPA without enlarging thecircuit size.

Fig. 20 depicts the measured power added efficiency (PAE)and drain efficiency (DE) for the designed amplifiers. TheDPAs have a comparable performance for the region withoutput power lower than 35 dBm. Operation in the intended


Fig. 20. Measured power added efficiencies (PAE) and drain efficiency (DE)of the proposed and conventional DPAs.

Fig. 21. Measured ACLRs of the DPAs for LTE signals.

back-off can be observed when the input power is larger than30 dBm. For the high power region, the proposed DPA is moreefficient due to its better load modulation. The highest outputpower is extended from 38.1 dBm to 40 dBm. More impor-tantly, the corresponding PAE is increased dramatically from31.8% to 41.8%.

To evaluate the linearity of the amplifiers, they were drivenwith a 10-MHz LTE signal at 1800 MHz. ACLR measurementsat 20 MHz offset from the carrier frequency were performed asshown in Fig. 21. It is observed that the ACLR of the proposedDPA is at least 12 dB better than the conventional DPA for theregion with output power larger than 25 dBm. The proposedDPA achieves a maximum ACLR level of −40.7 dBc, which isaround 5.8 dB better than that of the conventional one.

Table II summarizes the performance of the proposed DPAwith the conventional DPA and other PAs presented in theliterature [12], [14], [18], [19]. The proposed DPA exhibitsbandpass filtering with a wide suppression band up to the 3rdharmonic, which cannot be observed in previous works. Theintroduction of BPFs in the PAs [18], [19] mainly focuses onthe integration of bandpass filtering. Different from this, theproposed DPA can provide an efficiency improvement of 10%,which is larger than those for adaptive input-power distribution

TABLE IICOMPARISON BETWEEN PROPOSED DPA AND PREVIOUS WORKS

approach [12] and complex combing load [14]. In addition,the proposed configuration also demonstrated good linearitycompared to other works. Therefore, an enhanced Dohertyoperation is achieved, with bandpass characteristics and im-proved efficiency and linearity performance compared to theconventional DPA.

V. CONCLUSION

The proposed DPA uses a patch quadrature coupler with arbi-trary coupling coefficient as the input power divider. The patchcoupler is designed to provide bandpass filtering, wideband har-monic suppression, and arbitrary power division. Besides thebandpass filtering functionality, an enhanced Doherty operationis achieved, with an improved efficiency and linearity perfor-mance compared to a conventional DPA. The proposed Dohertyamplifier demonstrates a measured ACLR improvement betterthan 5.8 dB within the high output power region, as well as anincreased power-added-efficiency of 10% at 1.8 GHz.

REFERENCES

[1] F. H. Raab, P. Asbeck, S. Cripps, P. B. Kenington, Z. B. Popovic,N. Pothecary, J. F. Sevic, and N. O. Sokal, “Power amplifiers and transmit-ters for RF and microwave,” IEEE Trans. Microw. Theory Tech., vol. 50,no. 3, pp. 814–826, Mar. 2002.

[2] F. H. Raab, “High-efficiency linear amplification by dynamic load mod-ulation,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, vol. 3,pp. 1717–1720.

[3] W. H. Doherty, “A new high efficiency power amplifier for modulatedwaves,” Proc. IRE, vol. 24, pp. 1163–1182, Sep. 1936.

[4] D. Kang, D. Kim, Y. Cho, B. Park, J. Kim, and B. Kim, “Design ofbandwidth-enhanced Doherty power amplifiers for handset applications,”IEEE Trans. Microw. Theory Tech., vol. 59, no. 12, pp. 3474–3483,Dec. 2011.

[5] W. Chen, S. Zhang, Y. Liu, Y. Liu, and F. M. Ghannouchi, “A concur-rent dual-band uneven Doherty power amplifier with frequency-dependentinput power division,” IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 61,no. 2, pp. 552–561, Feb. 2014.

[6] A. M. Mohamed, S. Boumaiza, and R. Mansour, “ReconfigurableDoherty power amplifier for multifrequency wireless radio systems,” IEEETrans. Microw. Theory Tech., vol. 61, no. 4, pp. 1588–1598, Apr. 2013.

[7] A. M. Mohamed, S. Boumaiza, and R. Mansour, “Electronically tun-able Doherty power amplifier for multi-mode multi-band base stations,”IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 61, no. 4, pp. 1229–1240,Apr. 2014.

[8] J. Nam and B. Kim, “The Doherty power amplifier with on-chip dynamicbias control circuit for handset application,” IEEE Trans. Microw. TheoryTech., vol. 55, no. 4, pp. 633–642, Apr. 2007.

[9] T. Apel, Y.-L. Tang, and O. Berger, “Switched Doherty power amplifiersfor CDMA and WCDMA,” in Proc. IEEE Radio Freq. Integr. Circuits(RFIC) Symp., Jun. 2007, pp. 259–262.

[10] J. Sirois, S. Boumaiza, M. Helaoui, G. Brassard, and F. M. Ghannouchi,“A robust modeling and design approach for dynamically loaded and dig-itally linearized Doherty amplifiers,” IEEE Trans. Microw. Theory Tech.,vol. 53, no. 9, pp. 2875–2883, Sep. 2005.

[11] J. Kim, J. Cha, I. Kim, and B. Kim, “Optimum operation of asymmetrical—Cells-based linear Doherty power amplifiers—Uneven power drive andpower matching,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 5,pp. 1802–1809, May 2005.


[12] M. Nick and A. Mortazawi, “Adaptive input-power distribution in Dohertypower amplifiers for linearity and efficiency enhancement,” IEEE Trans.Microw. Theory Tech., vol. 58, no. 1, pp. 2764–2771, Oct. 2010.

[13] E. Kaymaksut, B. Francois, and P. Reynaert, “Analysis and optimiza-tion of transformer-based power combining for back-off efficiency en-hancement,” IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 60, no. 4,pp. 825–835, Apr. 2013.

[14] X. H. Fang and K.-K. M. Cheng, “Extension of high-efficiency rangeof Doherty amplifier by using complex combining load,” IEEE Trans.Microw. Theory Tech., vol. 62, no. 9, pp. 2038–2047, Sep. 2014.

[15] S. Chen and Q. Xue, “Compact triple-transistor Doherty amplifier de-signs: Differential/power combining,” IEEE Trans. Microw. Theory Tech.,vol. 61, no. 5, pp. 1957–1963, May 2013.

[16] S. Watanabe, Y. Takayama, R. Ishikawa, and K. Honjo, “A miniaturebroadband Doherty power amplifier with a series-connected load,” IEEETrans. Microw. Theory Tech., vol. 63, no. 2, pp. 572–579, Feb. 2015.

[17] K. Chen, J. Lee, W. J. Chappell, and D. Pertoulis, “Co-design of highlyefficient power amplifier and high-Q output bandpass filter,” IEEE Trans.Microw. Theory Tech., vol. 61, no. 11, pp. 3940–3950, Nov. 2013.

[18] Y. C. Li, K. C. Wu, and Q. Xue, “Power amplifier integrated with band-pass filter for long term evolution application,” IEEE Microw. WirelessCompon. Lett., vol. 23, no. 8, pp. 424–426, Aug. 2013.

[19] L. Gao, X. Y. Zhang, and Q. Xue, “Compact power amplifier with band-pass response and high efficiency,” IEEE Microw. Wireless Compon. Lett.,vol. 24, no. 10, pp. 707–709, Oct. 2014.

[20] T. Mury and V. F. Fusco, “Inverse class-E amplifier with transmission lineharmonic suppression,” IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 54,no. 7, pp. 1555–1561, Jul. 2007.

[21] Z. K. Ma and G. A. E. Vandenbosch, “Wideband harmonic rejectionfiltenna for wireless power transfer,” IEEE Trans. Antennas Propag.,vol. 62, no. 1, pp. 371–377, Jan. 2014.

[22] R. Mirzavand and M. M. Honari, “Compact microstrip Wilkinsonpower dividers with harmonic suppression and arbitrary power divisionratios,” IEEE Trans. Microw. Theory Tech., vol. 61, no. 1, pp. 61–68,May 2013.

[23] C. S. Aitchison, M. Mbabele, M. R. Moazzam, D. Budimir, and F. Ali,“Improvement of third-order intermodulation product of RF and mi-crowave amplifiers by injection,” IEEE Trans. Microw. Theory Tech.,vol. 49, pp. 1148–1154, Jun. 2001.

[24] H. Uchida, N. Yoneda, Y. Konishi, and S. Makino, “Bandpass directionalcouplers with electromagnetically-coupled resonators,” in IEEE MTT-SInt. Microw. Symp. Dig., 2006, pp. 1563–1566.

[25] K.-Y. Tsai, H.-S. Yang, J.-H. Chen, and Y.-J. E. Chen, “A miniaturized3 dB branch-line hybrid coupler with harmonics suppression,” IEEEMicrow. Wireless Compon. Lett., vol. 21, no. 10, pp. 537–539, Oct. 2011.

[26] J.-A. Hou and Y.-H. Wang, “Design of compact 90◦ and 180◦ couplerswith harmonic suppression using lumped-element bandstop resonators,”IEEE Trans. Microw. Theory Tech., vol. 58, no. 11, pp. 2932–2939,Nov. 2010.

[27] S. Y. Zheng, J. H. Deng, Y. M. Pan, and W. S. Chan, “Circular sectorpatch hybrid coupler with arbitrary coupling coefficient and phase differ-ence,” IEEE Trans. Microw. Theory Tech., vol. 61, no. 5, pp. 1781–1792,May 2013.

[28] H. Zhang, B. Hu, and X. Zhang, “Compact equal and unequal dualfrequency power dividers based on composite right/left-handed transmis-sion lines,” IEEE Trans. Ind. Electron., vol. 59, no. 9, pp. 3464–3472,Sep. 2012.

[29] C. A. Balanis, Modern Antenna Handbook. New York, NY, USA: Wiley,2008.

Shao Yong Zheng (S’07–M’11) was born in FujianProvince, China. He received the B.S. degree in elec-tronic engineering from Xiamen University, Fujian,China, in 2003, the M.Sc., M. Phil., and Ph.D. de-grees in electronic engineering from City Universityof Hong Kong, Kowloon, Hong Kong, in 2006, 2008,and 2011, respectively.

From 2011 to 2012, he was a Research Fellowwith Department of Electronic Engineering, CityUniversity of Hong Kong. He is currently an Asso-ciate Professor with Department of Electronics and

Communication Engineering, Sun Yat-sen University, Guangzhou, China. Hisresearch interests include microwave/millimeter wave circuits and evolutionaryalgorithms.

Zhao Wu Liu was born in Hunan Province, China.He received the B.S. degree in electronic engineeringfrom Hunan University of Science and Engineering,Hunan, China, in 2014, and is currently workingtoward the MEng. degree at Sun Yat-sen Univer-sity, Guangzhou, China. His research interests in-clude high efficiency power amplifiers and linearizedpower amplifiers.

Yong Mei Pan (M’11) was born in Huangshan,Anhui Province, China, in 1982. She received theB.Sc. and Ph.D. degrees in electrical engineeringfrom the University of Science and Technology ofChina (USTC), in 2004 and 2009, respectively.

She is currently an Professor with School of Elec-tronic and Information Engineering, South ChinaUniversity of Technology, Guangzhou, China. Herresearch interests include dielectric resonator anten-nas, leaky wave antennas, and metamaterials.

Yongle Wu received the B.Eng. degree in commu-nication engineering and the Ph.D. degree in elec-tronic engineering from Beijing University of Postsand Telecommunications (BUPT), Beijing, China, in2006 and 2011, respectively.

During April to October in 2010, he was a Re-search Assistant at the City University of Hong Kong(CityU), Kowloon, Hong Kong. In 2011, he joinedthe BUPT and became an Associate Professor inthe School of Electronic Engineering in BUPT. Hisresearch interests include generalized Smith charts,

generalized transmission lines, and microwave components design.

Wing Shing Chan (M’94) received the B.Sc.(Eng.)degree in electronic engineering from Queen MaryCollege, University of London, U.K., in 1982 and thePh.D. degree from City University of Hong Kong,in 1995.

From 1982 to 1984, he worked for PlesseyRADAR in the Solid-State Techniques Departmentas an Engineer as part of a team that producedthe world’s first solid-state RADAR transmitter ins-band. From 1984 to 1988 he worked for MicrowaveEngineering Designs Limited as a Senior Design

Engineer in RF/Microwave amplifiers. In 1988 he joined the Department ofElectronic Engineering at the City University of Hong Kong as a Lecturer.He is now an Associate Professor in the same department.

Dr. Chan is a Chartered Engineer of the Engineering Council, U.K., and amember (MIEE) of the IEE since 1991. He was the past Chairman of the IEEEAP/MTT Chapter, HK Section. He has previously served as a member of theRadio Spectrum Advisory Committee (RSAC) in the Office of the Telecommu-nications Authority.

Yuanan Liu received the B.E., M.Eng., and Ph.D.degrees in electrical engineering from Universityof Electronic Science and Technology of China,Chengdu, in 1984, 1989, and 1992, respectively.

In 1984, he joined the 26th institute of ElectronicMinistry of China to develop the inertia navigatingsystem. In 1992, he began his first post-doctor po-sition in EMC lab of Beijing University of Postsand Telecommunications (BUPT), Beijing, China. In1995, he started his second post-doctor in broadbandmobile lab of Department of System and Computer

Engineering, Carleton University, Ottawa, Canada. From July 1997, as Profes-sor, he is with wireless communication center of College of TelecommunicationEngineering, BUPT, where he is involved in the development of next-generationcellular system, wireless LAN, Bluetooth application for data transmission,EMC design strategies for high speed digital system, and EMI and EMSmeasuring sites with low cost and high performance.

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