EVS28 International Electric Vehicle Symposium and Exhibition 1

EVS28

KINTEX, Korea, May 3-6, 2015

Boost Converter Selection and analysis for Automotive

Applications

Moataz Elsied1, 2, Amrane Oukaour1, Hamid Gualous1, Youssef salmanie1, Amr Amin2,

Radwan Hassan2 1 Lusac laboratoire –, BP78 Rue Louis Aragon (Cherbourg-France)

Université of Caen Basse- Normandie, Caen, France

Phone:0033 986 812685, e-mail: amrane.oukaour@unicaen.fr

2Electrical Power and Machines Department

Helwan University, Faculty of Engineering

Helwan, Cairo, Egypt

Phone: 0033 689 326 447, e-mail: moataz.elsied@unicaen.fr

Abstract

During the past decade, power electronics research has focused on the development of multi-phase dc/dc

power converters for electric vehicles (EVs) and DC micro-grids applications, and it is estimated that there

is still a huge potential of this field during the coming years. In this paper, Four-Phase Interleaved Boost

Converter (FP-IBC) that interfaces different power sources with the powertrain of hybrid electric vehicles

(HEVs) is introduced, discussed and analyzed. The proposed converter is compared with different topologies

such as conventional boost converter (BC), Multi-device boost converter (MDBC), and Two-phase

interleaved boost converter (TP-IBC) and Multi-device interleaved boost converter (MD-IBC). The

comparison is performed at different switching frequencies and power ratings to show the effect of these

variables on the converters losses. The comparison indicated that the FP-IBC is able to the size of passive

components with high efficiency compared with the other topologies of boost converters. The simulation

results and analysis proved that FP-IBC is more powerful than other dc/dc converter topologies in achieving

high performance and reliability for high-power rating dc/dc converters.

Keywords: electric vehicles (EVs), DC micro-grids, Four-Phase Interleaved Boost Converter (FP-IBC), small

signal model (SSM).

1 Introduction The growing number of human population yields

increased energy consumption and depletion of

finite resources such as, oil and gas. Therefore,

Electrical Vehicles (EVs) are seriously considered

as an alternative to fossil-fueled vehicles [1]-[2].

In EVs, fuel cells (FCs), super-capacitors (SCs),

and batteries are usually used as energy storage

devices. Combining such energy sources leads to a

FC/SC/battery hybrid power system (HPS) [3]-[4],

as it is shown in Figure. 1. Unlike single-sourced

systems, HPS has the potential to provide the load

with high quality, more reliable, and efficient

power. In these systems, FCs are emerging as a

promising supplementary power sources due to

their merits of cleanness, high efficiency, and high

reliability. Because of FC’s drawbacks such as long

startup time and slow dynamics, mismatch power

EVS28 International Electric Vehicle Symposium and Exhibition 2

between the load and FC should be managed by

an Energy Storage System (ESS) such as batteries

and SCs [5]-[6].

On the other hand, batteries are usually used as

storage devices for smoothing output power,

enhancing the peak power capacity, and

improving the system dynamic characteristics and

startup transitions. But, batteries suffer from some

serious drawbacks such as low life cycle, low

power density and long recharging time. To

overcome these disadvantages, SCs are used in

transition periods for their high power density to

deliver the required energy. However, their energy

density is low which limits their contribution

during extended transient states. Therefore, a

combination of batteries and SCs seems to be a

better choice as ESS working with FC as a

supplementary source, which not only provides a

higher power density but also increases the energy

storage capability of EVs [7].

In EVs, a dc/dc boost converter is a key element

to interface HPSs to the EV’s dc-bus. Various

dc/dc boost converters topologies have been

studied and analyzed for EV applications in [8]-

[11]. It is illustrated in [12] that Multi-Device

Interleaved Boost Converter (MD-IBC) is more

powerful and more efficient than other boost

converter topologies such as conventional Boost

Converter (BC), Two-Phase Interleaved Boost

Converter (TP-IBC), and Multi-Device Boost

Converter (MDBC).

In this paper, the proposed FP-IBC outperforms

MD-IBC and other topologies of boost converters

with its higher reliability, higher efficiency,

higher power and voltage ratings. The FP-IBC and

the other converters are modeled and simulated

using Matlab/Simulink. Simulation results are

analyzed and discussed. Moreover, the

converters’ power losses are evaluated

numerically to benchmark their efficient

performance. Simulation results along with

numeric comparison for efficiency have

demonstrated that the proposed converter is very

promising for EVs applications.

Figure 1: Block diagram for the FC/SC/battery hybrid

power system

2 Structure of the FP-IBC The structure of the FP-IBC is depicted in Figure.

2a. The proposed converter consists of four dc/dc

boost converter modules connected in parallel.

Figure. 2b shows the switching device gate signals

at D=0.5 where D is the duty cycle. The gate signals

are successively phase shifted by Ts /(n*m), where

Ts is the switching period, n is the number of phases,

and m is the number of parallel switches per phase.

For FP-IBC, m=1 and n=4. As such, the current

delivered by the electric source is shared equally

between each phase and has a ripple content of

period Ts/4. Similarly, the frequency of the output

voltage and the input current is n times higher than

the switching frequency fsw. As result, the size of

passive devices such as capacitor and inductor will

be reduced by n times compared to the conventional

BC. In addition, the system reliability and converter

power rating will be increased by using paralleling

phases. Moreover, the current sharing equally

between each phase will provide tight sizing of

power semiconductors, distribution of losses

between modules and size’s optimization of the

converter. These advantages are behind the use of

FP-IBC as a dc/dc converter for EV power systems

in particular for high power applications as opposed

to other converter topologies. The structures of

conventional BC, MDBC, TP-IBC, and MD-IBC

are described and discussed in details in [12].

(a)

(b)

Figure 2: (a) FP-IBC structure (b) The switching pattern

of FP-IBC

EVS28 International Electric Vehicle Symposium and Exhibition 3

3 FP-IBC Modeling The boost converter’s double pole and right half

plane (RHP) zero whose location in the s-plane is

dependent on the input voltage, output voltage,

load resistance, inductance, and output

capacitance. Therefore, appropriate selection of

the transfer function parameters and control law

are important for the system stability criteria to

achieve proper operation.

In this section, a generalized small signal model

(SSM) which is derived in [13]-[15] is applied to

the proposed converter to develop a transfer

function relating the duty cycle with the inductor

current and the output voltage. This model can

also be applied for all topologies of boost

converters which are discussed in this paper. The

transfer function between inductor current and

duty cycle (1) is a second order, two poles in the

left-half plane (LHP) and a LHP zero. The LHP

zero is written in (2).

𝐻𝑖(𝑠) =𝐼𝐿(𝑠)

𝑑(𝑠)=

𝑉𝑜 (𝑚+𝜎)

𝜎 𝑅𝑙+𝑛 𝑅𝑜(1−𝑚𝐷)2

(1+ 𝑠 𝜔𝑧𝑖⁄ )

(𝑠2

𝜔02+

2𝜉𝑠

𝜔0+1)

(1)

𝜔𝑧𝑖 = 1

𝐶( 𝑅𝑐+𝜎 𝑅𝑜

𝑚+𝜎) (2)

𝐻𝑣(𝑠) =𝑣𝑜(𝑠)

𝑑(𝑠)=

𝑉𝑜 [𝑛 𝑅𝑜(1−𝑚𝐷)2−𝑚 𝑅𝑙]

(1−𝐷) [𝑛 𝑅𝑜(1−𝑚𝐷)2+𝜎 𝑅𝑙] (1+ 𝑠 𝜔𝑧𝑣1⁄ )(1− 𝑠 𝜔𝑧𝑣2⁄ )

(𝑠2

𝜔02+

2𝜉𝑠

𝜔0+1)

(3)

𝜔𝑧𝑣1 = 1

𝐶 𝑅𝑐 (4)

𝜔𝑧𝑣2 = 𝑛 𝑅𝑜(1−𝑚𝐷)2−𝑚 𝑅𝑙

𝑚 𝐿 (5)

Formulation (3) consists of double pole, RHP zero

and LHP zero and this equation describes the

relation between output voltage and duty cycle.

The zero in the LHP is introduced in (4), and the

zero in RHP is given in (5).

𝜔0 = √𝜎 𝑅𝑙+ 𝑛 𝑅𝑜(1−𝑚𝐷)2

𝜎 𝐿 𝐶 (𝑅𝑜+𝑅𝑐) (6)

The double pole frequency ω0 presented in (6)

depends on the input voltage (𝑣𝑖𝑛) and the output

voltage (𝑣𝑜) as well as inductance (𝐿) and output

capacitance (𝐶). It is also important to note that

ω0 depends on the load resistance ( 𝑅𝑜 ), the

internal resistance of the inductor (𝑅𝑙 ) and the

internal resistance of the capacitor ( 𝑅𝑐 ). The

system damped ratio for both transfer functions is

given by,

𝜉 = 𝜎 𝐿+𝐶 [𝜎 𝑅𝑙 (𝑅𝑜+𝑅𝑐)+𝑛 𝑅𝑐 𝑅𝑜(1−𝑚𝐷)2]

2 √𝜎 𝐿 𝐶 (𝑅𝑜+𝑅𝑐)[𝜎 𝑅𝑙+ 𝑛 𝑅𝑜(1−𝑚𝐷)2] (7)

Where 𝐷 is the nominal duty ratio and its

expression is described by,

𝐷 = 𝑣𝑜−𝑣𝑖𝑛

𝑚 𝑣𝑜 𝜎 =

(1−𝑚 𝐷)

(1−𝐷) (8)

4 Closed Loop Control Design The boost converter feedback control is a nonlinear

function of the duty cycle, which makes the

controller design is more challenging from the

viewpoint of stability and bandwidth [16]-[17]. The

control design of different topologies of boost

converter consists of two control loops, inner and

outer loop, the inner loop is used for current control,

which is much faster than the voltage outer control

loop as it is shown in Figure .3. Where Cv(s) is the

voltage compensator and Ci(s) is current

compensator that assures cancellation of the static

error and high bandwidth. The resultant control

signal is (m1), while the duty cycle signal is (d).

The transfer function of the current and voltage

controllers are introduced in [15] and it is used in

this work to regulate the converter’s output voltage.

Compensators used in the inner and outer loops

introduce two poles and a zero. A pole at the origin

is considered as an integral action and provides a

very high gain at low frequencies. Moreover, the

pole-zero pairs (𝑝𝑖 ,𝑧𝑖) for current controller and

(𝑝𝑣 ,𝑧𝑣) for voltage controller aim to reduce the

phase shift between the frequency of the two plant

zeros and the frequency of two plant poles.

The transfer function (9) and (10) are used to design

the current and voltage controller and can be written

as the following:

𝐶𝑖(𝑠) = 𝑘𝑐𝑖 (𝑠+ 𝑧𝑖)

𝑠(𝑠+ 𝑝𝑖) (9)

𝐶𝑣(𝑠) = 𝑘𝑐𝑣 (𝑠+ 𝑧𝑣)

𝑠(𝑠+ 𝑝𝑣) (10)

𝑧𝑖 = 𝑤𝑐𝑛𝑖 √1−sin ∅𝑚𝑑𝑖

1+sin ∅𝑚𝑑𝑖 , 𝑝𝑖 =

𝑧𝑖

√1−sin ∅𝑚𝑑𝑖1+sin ∅𝑚𝑑𝑖

(11)

𝑧𝑣 = 𝑤𝑐𝑛𝑣 √1−sin ∅𝑚𝑑𝑣

1+sin ∅𝑚𝑑𝑣 , 𝑝𝑣 =

𝑧𝑣

√1−sin ∅𝑚𝑑𝑣1+sin ∅𝑚𝑑𝑣

(12)

Where 𝑤𝑐𝑛𝑖 and 𝑤𝑐𝑛𝑣 are the new crossover

frequency for the current loop and the voltage outer

loop, respectively. The compensator phase margin

of the current and the voltage controller are

abbreviated as ∅𝑚𝑑𝑖 and ∅𝑚𝑑𝑣, respectively. 𝑘𝑐𝑖 is

the current controller gain where 𝑘𝑣𝑖 is the voltage

EVS28 International Electric Vehicle Symposium and Exhibition 4

controller gain which can be calculated as the

following:

𝑘𝑐𝑖 = |1

Ti(S)| at 𝑤𝑐 = 𝑤𝑐𝑛𝑖 (13)

𝑘𝑣𝑖 = |1

Tv(S)| at 𝑤𝑐 = 𝑤𝑐𝑛𝑣 (14)

Where 𝑇𝑖(𝑠), 𝑇𝑣(𝑠) are the open loop transfer

functions for the inner and outer loop respectively.

𝑇𝑖(𝑠) = 𝐶𝑖(𝑠)𝐻𝑖(𝑠) for the inner loop

(15)

Tv(s) =Cv(s)Ci(s)Hv(s)

1+Ci(s)Hi(s) for the outer loop

(16)

Figure 3: Control design loop configuration

5 Power-Losses calculations

model This section aims to provide a mathematical tool

for the calculation of total power losses in boost

converter. The total power losses (𝑝𝑙𝑜𝑠𝑠_𝑡𝑜𝑡) in any

boost converter can be considered as the sum of

the inductor copper losses (𝑝𝑙), the inductor core

losses ( 𝑝𝑐𝑜𝑟𝑒 ), the capacitor losses ( 𝑝𝑐) , the

switching (𝑝𝑠) and the conduction (𝑝𝑐𝑜𝑛) losses of

the switching devices [18].

In this study, the Metal Oxide Semiconductor

Field Effect Transistor (MOSFET) and diode

represent the switching devices. The switching

losses of MOSFET ( 𝑝𝑠𝑀 ) and diode ( 𝑝𝑠𝐷 )

calculations are based on the MOSFET data sheet

parameters and can be obtained from the

following equations:

𝑝𝑠 = 𝑝𝑠_𝑀+𝑝𝑠_𝐷 (17)

𝑝𝑠_𝑀 = (𝐸𝑜𝑛_𝑀 + 𝐸𝑜𝑓𝑓_𝑀) 𝑓𝑠𝑤 (18)

𝑝𝑠_𝐷 = (𝐸𝑜𝑛_𝐷 + 𝐸𝑜𝑓𝑓_𝐷) 𝑓𝑠𝑤 ≈ 𝐸𝑜𝑛_𝐷 𝑓𝑠𝑤

(19)

𝐸𝑜𝑛_𝑀 = 𝑈𝑑𝑑 𝐼𝐷_𝑜𝑛 𝑡𝑟𝑖+𝑡𝑓𝑢

2 + 𝑄𝑟𝑟 (20)

𝐸𝑜𝑓𝑓_𝑀 = 𝑈𝑑𝑑 𝐼𝐷_𝑜𝑓𝑓 𝑡𝑟𝑢+𝑡𝑓𝑖

2 (21)

𝐸𝑜𝑛_𝐷 = 1

4𝑄𝑟𝑟 𝑈𝑑𝑟𝑟 (22)

Where 𝐸𝑜𝑛_𝑀 , 𝐸𝑜𝑛_𝐷 are the turn-on energy losses

of the power MOSFET and diode respectively,

𝐸𝑜𝑓𝑓_𝑀 , 𝐸𝑜𝑓𝑓_𝐷 are the turn-off energy loss, 𝑓𝑠𝑤 is

the switching frequency, 𝑈𝑑𝑑 is the open-circuit

voltage on the switching device, 𝑈𝑑𝑟𝑟 is the voltage

across the diode during reverse recovery, 𝑄𝑟𝑟 is the

reverse recovery charge, 𝑡𝑟𝑖 is switch rated rising

time, and 𝑡𝑓𝑖 is the rated falling time. 𝐼𝐷_𝑜𝑛 ,𝐼𝐷_𝑜𝑓𝑓,

𝑡𝑓𝑢 and 𝑡𝑟𝑢 analysis are listed in [19].

On the other hand, the conduction losses for

switching devices are calculated using the following

equations:

𝑝𝑐𝑜𝑛 = 𝑝𝑐_𝑀 + 𝑝𝑐_𝐷 (23)

𝑝𝑐_𝑀= 𝑅𝐷𝑆𝑜𝑛 𝐼𝑀_𝑟𝑚𝑠2 (24)

𝑝𝑐_𝐷= 𝑈𝐷𝑜 𝐼𝐷_𝑎𝑣 + 𝑅𝐷 𝐼𝐷_𝑟𝑚𝑠2 (25)

Where 𝑝𝑐_𝑀 , 𝑝𝑐_𝐷 are the conduction power losses

for the power MOSFET and diode as well as

𝐼𝑀_𝑟𝑚𝑠 , 𝐼𝐷_𝑟𝑚𝑠 are the root-mean square (RMS)

value of the drain current and diode current

respectively. 𝑅𝐷𝑆_𝑜𝑛, 𝑅𝐷 are the resistance of the

MOSFET and diode during conduction, 𝐼D_𝑎𝑣 is the

average value for the diode current and 𝑈𝐷𝑜 is the

diode on-state zero-current voltage.

The copper losses of the passive components are

approximately given by:

𝑝𝑐 + 𝑝𝑙 = 𝐼𝑐_𝑟𝑚𝑠2 𝑅𝑐 + 𝐼𝐿_𝑟𝑚𝑠

2 𝑅𝑙 (26)

Traditionally, inductor core loss has been divided

up into two components: hysteresis loss and eddy

current loss. According to the Steinmetz equation,

estimation of core losses based on the charts given

by the manufacturer (METGLAS, POWERLITE®

C-Cores) are calculated as follow:

𝑝𝑐𝑜𝑟𝑒 (W) = 𝑎. 𝑓𝑠𝑤𝑏. 𝐵𝑐 . 𝑤 (27)

𝐵 = 0.4𝜋.𝑁.𝛥𝐼 .10−4

𝐿𝑔 (28)

Where 𝑓𝑠𝑤 is the switching frequency in kHz, 𝐵 is

flux density in Tesla, while 𝑎 , 𝑏 , and 𝑐 are the

coefficients, which depend on the lamination

material, thickness, conductivity, as well as other

factors. It is assumed in this work as listed in

manufacture datasheet [20] that a=6.5, b= 1.5, and

c=1.74. 𝑤 is the weight of the core (kg), N is the

number of turns per coil, 𝛥𝐼 is the current ripple,

and 𝐿𝑔 is the length of the air-gab (cm).

Finally, the output power ( 𝑝𝑜 ) and converter

efficiency (𝜂) are shown the next equations:

EVS28 International Electric Vehicle Symposium and Exhibition 5

𝑝𝑜 = 𝑣𝑜 𝐼𝑜_𝑎𝑣𝑟 (29)

𝑝𝑙𝑜𝑠𝑠__𝑡𝑜𝑡 = 𝑝𝑐 + 𝑝𝑙 + 𝑝𝑐𝑜𝑟𝑒 + 𝑝𝑠 + 𝑝𝑐𝑜𝑛 (30)

𝜂 =𝑝𝑜

𝑝0+𝑝𝑝+ 𝑝𝑠+𝑝𝑐𝑜𝑛 (31)

Where 𝐼𝑜_𝑎𝑣𝑟 is the average output current and

𝐼𝑐_𝑟𝑚𝑠 , 𝐼𝐿_𝑟𝑚𝑠 are the RMS value of capacitor and

inductor current.

6 Simulation results and

discussion A comparative study is carried-out to evaluate the

proposed converter’s efficiency with regards to

other boost converter topologies. The system is

implemented using Matlab®/Simulink to

investigate the dynamic performance of the FP-

IBC compared to different topologies of boost

converter. Moreover, the efficiency of all studied

converters is investigated at different power rating

and switching frequencies. The converters’

parameters are summarized in Table. 1.

Simulation runs are carried-out using a battery

whose voltage is set to 100V as the converters

input voltage and resistive load (defined as

common load). The battery module comprises a

package with Ns cells that are connected in series

and Np batteries that are connected in parallel.

The parameters and characteristic of the battery

are introduced in Table. 2.

The output voltage reference is set to 200V. Using

(17)-(31), the switching and conduction losses are

calculated for each switch. The analytical

calculation is based on the datasheet of the

MOSFET module SKM121AR.

Figures.4, 5, 6, 7 and 8, respectively, show the

dynamic response of BC, MDBC, TP-IBC, MD-

IBC and FP-IBC during the step load variation. It

can be noticed from the figures that the input

current ripples frequency of FP-IBC is multiplied

by four compared to conventional BC and two

times compared to TP-IBC and MDBC. As a

result, the size of the passive components

(inductor, output capacitor, and EMI filter) is

reduced by 75% compared with conventional BC

and 50% compared with TP-IBC topology at the

same switching frequency. Moreover, the

dynamic response of the FP-IBC is faster than

other topologies and it has very small voltage

ringing at load step, because the interleaved

control between the power switches provides a

higher system bandwidth. It can be also noticed

from the figures that MD-IBC has approximately

the same dynamic performance as FP-IBC. For

this purpose, the current study should be

completed by the comparative efficiency study to

provide the efficient model for EVs application.

Table 1: dc/dc converter parameters

Items 𝐑𝐥(𝑚𝛺) 𝐑𝐜(𝑚𝛺) 𝐋(𝜇𝐻) 𝐂(𝜇𝐹) 𝒏 𝒎

Con-BC 64 0.50 600 800 1 1

MDBC 32 1.2 300 400 1 2

TP-IBC 32 1.2 300 400 2 1

MD-IBC 14 2 150 200 2 2

FP- IBC 14 2 150 200 4 1

Table 2: Battery module parameters

Capacity 65Ah

Initial SOC% 80%

Nominal voltage 19.2V

No of cell in series, Ns 5

No of parallel modules, Np 3

Figure 4: dynamic response of BC

EVS28 International Electric Vehicle Symposium and Exhibition 6

Figure 5: dynamic response of MDBC

Figure 6: dynamic response of TP-IBC

Figure 7: dynamic response of MD-IBC

Figure 8: dynamic response of FP-IBC

Figures. 9, 10, and 11 show the converters

efficiency versus the switching frequency range

from 5 to 40 kHz at power ratings of 5kW, 10kW,

and 20kW. It is worthy to note that for the switching

frequency range [10–20 kHz] and for the case of

power equal to 20kW, the efficiency decreases by

0.157 % for the FP-IBC and by 0.259 % for the

conventional BC (see Figure. 11).

For the change in the operating power rating range

from [10kW-20kW] at the same switching

frequency [20 kHz], the efficiency decreases by

0.1% for the FP-IBC and 9.69% for the

conventional BC (see Figures.10, 11).

Figure 9: Converters efficiency versus switching

frequency at 5kw.

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Figure 10: Converters efficiency versus switching

frequency at 10kw.

Figure 11: Converters efficiency versus switching

frequency at 20kw.

As it is shown in Figures. 9, 10, and 11, the

proposed FP-IBC converter has the highest

efficiency and hence, the lowest total power loss

compared to other boost converter topologies, in

particular at high power and high switching

frequencies.

Figure.12 presents the passive elements and

switching devices power losses of the FP-IBC and

other boost converter topologies at fsw =20kHz

and 20kW. This case is chosen to investigate the

worst case scenario in terms of power losses. On

the other hand, Figure. 13 shows the distribution

of all losses for each converter at the same

operating conditions.

As it is illustrated in Figures. 12 and 13, the

proposed converter is able to reduce the total

losses and especially the passive elements losses.

This comes from the reduction of its passive

components size by four times compared to

conventional BC and two times compared to TP-

IBC and MDBC. It is noteworthy that the

proposed converter (FP-IBC) efficiency

characteristics make it a good candidate for EVs,

particularly in high-power applications.

Figure 12: Converters MOSFET and Passive elements

power losses at 20kW.

Figure 13: Distributed power losses in (W) for each

converter

7 Conclusion In this research, an efficient FP-IBC has been

proposed for EVs applications. The performance of

the FP-IBC is compared against other boost

converter topologies and evaluated numerically

with power losses calculations. Simulation results

show that FP-IBC is the most efficient converter

among different boost converter topologies. FP-

IBC achieves this ranking because of the reduction

of its passive components size by four times

compared to conventional BC and two times

compared to TP-IBC and MDBC. Moreover, the

current and voltage ripples are also reduced by four

times compared to conventional BC and two times

compared to TP-IBC and MDBC. Simulation

results along with numeric comparison for

efficiency have demonstrated that the proposed

converter is very promising for EVs applications.

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Moataz Elsied was born in 1982. He

received the B.Sc. and M.Sc. in

Electrical Engineering from Helwan

University, Helwan, Egypt, in 2004,

2011. He is currently a PhD

candidate at LUSAC laboratory-

University of Caen Basse-Normadie,

Cherbourg, France. His research

interests include Energy

management systems, Power

electronic converters, and real time

control systems of smart grids.

A. Oukaour was born in Algeria

1963. He received Ph.D degree in

electrical engineering from Pierre &

Marie-Curie University (Paris IV),

France in 1993. From 1995 to 2003

he was an Associate Professor at the

Antilles and French Guyana

University. From 2003 to 2011 he

was an Associate Professor at the

Caen Basse Normandie University.

His current research interests are in

the field of Ageing State Diagnosis

of Power Sources (Batteries –

Supercapacitors – Full Cell …).

Hamid Gualous received the Ph.D.

degree in electronics from the

university Paris XI Orsay,France, in

1994. From 1996 to 2009, he was an

Associate Professor at the university

of Franche-Comte in FEMTO-ST

laboratory, France. His main

research activities are concerning

energy storage device

(SuperCapcitors and battery), hybrid

power sources, and energy

EVS28 International Electric Vehicle Symposium and Exhibition 9

management for vehicle

applications.

Radwan Hassan was born in Egypt,

1947. He received the B.Sc. and

M.Sc. in Electrical Engineering from

Helwan University, Helwan, Egypt,

in 1970, 1978. He received Ph.D

degree in electrical engineering in

1982 from (UMIST)- U.K. He is

currently a full professor in the

university of Helwan, Egypt. His

research interests include

Energy Conversion, Drives,

Control, Interfacing, and Industrial

applications

Amr Amin was born in Egypt, 1955.

He received the B.Sc. in Electrical

Engineering from Helwan

University, Helwan, Egypt and the

M.Sc. from New Mexico State

University, in 1978, 1982. He

received Ph.D degree in electrical

engineering from New Mexico State

University, Las Cruces, New

Mexico, U.S.A., 1986... He is

currently a full professor in the

University of Helwan, Egypt. His

research interests include Power

Electronics, Renewable Energy,

Electric Drives, Power Systems,

Computer-Based Controllers, and

Artificial Intelligence.