Broad band Transmission over Residential Power Lines

International Journal of Science and Modern Engineering (IJISME)

ISSN: 2319-6386, Volume-1 Issue-12, November, 2013

44

Published By:

Blue Eyes Intelligence Engineering

& Sciences Publication

Retrieval Number: L05291111213/2013©BEIESP

Abstract: Bridging and Transmission of VDSL2 broadband over

power lines has received considerable attention recently to cater

to broadband distribution within the premises of a residence.

Power lines are fundamentally different from telephone lines

both in topology and load impedance. Power lines have a thicker

gauge and shorter straight lengths, apart from a large number of

bridge taps (BT) with inductive load terminations, which are not

matched to line impedances. In this paper ABCD parameters of

the individual sections are used to analyze the power line

channel of upto 10 bridge taps over a 600 meter length. The

noise profiles considered include periodic impulse noise which is

predominant over power line sections, apart from AWGN.

Impulse noise PSD has been computed.Tone loading profiles

have been obtained using Discrete Multitone Transmission

(DMT) as in VDSL2 over a bandwidth of 30 MHz. This analysis

points to the fact that lower Transmit PSD would suffice to

match the rates achievable by traditional VDSL2 when bridge

taps are open. However with inductive loads in the BTs as is

typical in residences, we recommend a two-step approach of (a)

equipping existing VDSL2 modem front end hybrids with

settable impedances that would approach a conjugate match of

the loaded line along with (b) capability to nominally increase

the Transmit PSD and added subbands to achieve the desired

rates in a seamless manner as in VDSL2.

Index Terms: channel modelling, discrete multitone, Power

line communication

I. INTRODUCTION

Traditional twisted pair copper already laid for plain old

telephone system supports last mile wired access with the

evolution of ADSL2 and VDSL2 standards. Distribution of

broadband over the last mile (local area) and last feet (in

premises) has received considerable attention recently and is

becoming as indispensable as access to electrical power.

There has been a growing interest in the possibility of

exploiting the power grid to provide broadband internet

access to residential customers [1]. The attractive feature is

the presence of a vast infrastructure in place for power

distribution.

Manuscript Received on November 2013.

Mrs.Usha Rani .K .R, Associate Professor, Dept. of ECE, R.V.College of

Engineering, Bangalore, Karnataka, India

Dr. S Ravi Shankar, Professor, Dept. of ECE, R.V.College of Engineering,

Bangalore, Karnataka, India

H.M.Mahesh, Professor, Department of Electronics, Bangalore University,

Bangalore, Karnataka, India

Nandan Nayak, Student, R.V.College of Engg.,Bangalore, Karnataka,India

Vijay Singh, NRB, R.V.College of Engg.,Bangalore, Karnataka,India.

Since 2006 power line standards have begun to evolve that

address the capability of power line network to distribute

broadband within a substation area and within the premises

of a house. The advantages are (a) economy of house cabling

(b) broadband access at all power points in a house and (c)

guaranteed rates unlike in WLAN where there is a loss of rate

due to shadowing effects, externally induced RF

interferences and concerns about long term effects of

Electromagnetic radiation.

The power line poses unique challenges in channel modeling

that have only been partially addressed in literature [5-11].

Apart from a thicker gauge (typically 14 AWG) the power

lines are fundamentally shorter in length (as compared to

telephone lines) but have a large number of parallel bridge

taps with predominantly inductive load terminations that are

often switched in and out of circuit rendering the channel to

be slow time varying frequency dependant. Further unlike

telephone lines that are terminated by modem analog front

ends that meet low, medium and long line length

impedances, power lines are never terminated with anything

that is even close to its characteristic impedance. Top-down

models for PLC channel transfer function employ a time

domain approach as proposed by Zimmermann and Dostert

[8], as well as by Philipps [9]. Bottom-up PLC channel

models use matrix representation for transfer functions

derived from transmission line theory as obtained by S. Galli

[1], H. Meng [7] and Huangqiang Li, Yunlian Sun [10].

However existing literature [20 - 22] on the data rates

achievable for typical in house PLC do not include the

presence of the dominant impulse noise, effect of loads being

switched in and out and consider a well defined Transmit

PSD. Further no recommendations to improve the data rates

are provided. In this paper we address the issue of data rates

achievable over Power line Carrier (PLC) for Discrete

Multitone (DMT) based line codes in the presence of impulse

noise and with inductive unmatched loads at BT

terminations. In an effort to reuse the signal processing tasks

of a DSL modem, we examine the Transmit PSD masks

employed in VDSL2 for Non-echo cancelled Frequency

division multiplex over typical PLC channel models. We

present the received signal to noise ratio (SNR) profiles of

typical PLC considering AWGN and impulse noise which is

predominant over power lines [17]. Rate adaptive tone

loading profiles have been presented with and without

impulse noise cases for loops upto 600 meters with upto ten

bridge taps when they are

shorted, open and terminated

with typical inductive loads.

When the bridge taps are open

Broad band Transmission over Residential

Power Lines Employing VDSL2: The Channel

Capacity Analysis

Usha Rani K R , Ravishankar S , H.M.Mahesh , Nandan Nayak, Vijay Singh

Broad band Transmission over Residential Power Lines Employing VDSL2: The Channel Capacity Analysis

45

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or short there is enough usable SNR available to meet rates of

80 Mbps or so the Transmit PSD can be reduced in such

cases. However, with inductive impedances present at the

bridge tap terminations there is a need to provide

approximate conjugate impedance to the line impedance seen

by the modem, so as to meet the required rates. It is

recommended that the modem be equipped with switchable

impedances in its hybrid arm to reasonably approach the

actual conjugate impedance as seen by the modem. The

constantly changing transfer function of the PLC due to the

load impedances getting switched in and out of BTs may be

managed by monitoring the SNRs in the sync frame for a

sudden drop, and initiate a seamless change in the rate like

the Quick rate adaptation scheme of VDSL2 by altering the

hybrid impedance on the fly. The paper is organized as

follows - In section II the transfer function of PLC models

have been developed based on the two-port network theory.

The channel capacities for different network topologies, with

impulse noise, unmatched load conditions employing DMT

are analyzed in section III, and Simulation results for

different topologies are presented in section IV.

II. POWER LINE MODELLING

Power lines were originally designed for transmission of

power at 50Hz. Unshielded power line has a very hostile

channel characteristic for high frequency signal propagation

up to 30 MHz needed for broadband distribution within a

house. There are two methods of modeling viz; the top-down

method and bottom-up method. In the Top-down method,

model parameters are obtained from measurements [5].

Accuracy of this model is affected by measurement

equipment and measurement methods. In this paper we use

the bottom-up approach to analyze the indoor power line

theoretically [2, 6]. In the bottom-up method the indoor

power line is modeled as a cascade of two-port networks with

„n‟ distributed elements, each one with system characteristics

described by either transmission or scattering matrices [7].

The main advantage of matrix representation is that it

intrinsically considers all the impedance discontinuities,

regardless of the network complexity [10]. In the following

subsections we describe the propagation parameters followed

by the analysis.

A. Power line parameters:

The live and neutral cables can be used as a PLC two-wire

transmission channel and regarded as a distributed

parameter network. Hence, it can be described by circuit

parameters that are distributed over its length viz; inductance,

capacitance, resistance and conductance given by [14]

f

aR

2

(1)

(2)

a

dC

1cosh

(3)

tan2 fCG (4)

Where „a‟ and „d‟ are the diameter and separation distance of

the power lines respectively. Here „µ‟ is permeability in free

space, „‟ its permittivity in free space, „σ‟ the conductivity

of conductor, and „δ‟ is the depth factor. Based on

transmission line theory, the propagation constant „γ‟ is [14]

22

2

22

2

81

81

2 L

RLCj

L

R

L

CR

(5)

Here „ω‟ is the angular frequency. Attenuation is the Real

part „α‟ of the propagation constant and the imaginary part

„β‟ is the phase constant.

A uniform transmission line can be modeled as a two- port

network. ABCD parameters are useful for characterizing

two-port networks shown in Fig 1.

Figure 1: Two-port model of power line

The transfer function of the network is given by [16]

BAZ

Z

v

vfH

L

L

1

2

(6)

The parameters A and B are elements of the transfer

matrix T described in the following sub-section.

B. Channel model:

Indoor power line network consists of a main (straight)

propagation path and multiple distribution paths (bridge

taps). The transfer matrix of main propagation Ts (straight

path) is [13]

f

R

a

dL

2cosh 1



46

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)cosh()sinh(

1

)sinh()cosh( o

linelineo

lineline

llZ

γlZγl

Ts

(7)

Where „lline‟ is the length of the line and γ is its propagation

constant.

The transfer matrix of distribution branch Td is [13]

dZ

TDC

BA

tapin

1

101

_ (8)

For a BT line with an open end the

)coth(_ tapZZ l

otapin

Where „ltap‟ is the length of the tap.

And for a BT line terminated with an inductive load the tap

input impedance is

)1(coshsinh

)1(sinhcosh

_

oR

oR

o

tap

o

tap

oR

oRtaptap

tapin

ZZ

ZZ

Z

l

Z

l

ZZ

ZZll

Z

C. Model analysis:

PLC employs thicker cables, AWG12 and AWG14 that

have far less attenuation compared to telephone cables.

However because of a significant number of the bridge taps

inside the house, there is an additional attenuation in power

cables. The topologies of indoor power lines considered for

analysis are shown in Figure 2. We compute the frequency

dependent transfer functions using the method described in

section B above. Specifically

1. Compute the ABCD parameters Ts and Td using

equations (7) & (8) respectively for 14AWG Power

cable for both cases with open bridge taps i.e ZL =

Infinity.

2. Repeat the same as in step 1 with an inductive load of

600mH that is typical in Fans and machines inside a

house.

3. Obtain the transfer function H(f) for cases 1 and 2 .

Figure 2: Indoor power line network topologies

The frequency dependent normalized transfer functions

20log are shown in Figures 3 to 6 for the loops shown

in Figure2. Note the presence of a dip in Figures 4 - 6 is due

to the presence of a bridge taps in loops 2 – 4, that act as

tuned LC circuit.

0 0.5 1 1.5 2 2.5 3

x 107

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

frequency

Tran

sfer

func

tion

H(f)

DB

simulated channel frequency response

Figure 3: Frequency response of loop 1]

0 0.5 1 1.5 2 2.5 3

x 107

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

Tra

nsfe

r fu

nctio

n H

(f)D

B

frequency

Figure 4: Frequency response of loop 2

0 0.5 1 1.5 2 2.5 3 3.5

x 107

-1000

-800

-600

-400

-200

0

200

Tran

sfer

func

tion

H(f)D

B

frequency


Figure 5: Frequency response of loop 3 with BT open.


47

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0 0.5 1 1.5 2 2.5 3 3.5

x 107

-3000

-2500

-2000

-1500

-1000

-500

0

Tra

nsfe

r fu

nction H

(f)D

B w

ith inductive load

frequency


Figure 6: Frequency response of loop 3 with

an inductive loaded BT.

III. ANALYSIS OF POWER LINE

CHANNEL CAPACITY IN THE

PRESENCE OF PERIODIC IMPULSIVE

When computing the channel capacity, the transmit

Power spectral density up to 30 MHz as per the VDSL2

standard G993.2 [12] is employed. Energy and bits in every

tone are allocated adaptively according to the channel

characteristics. The classification and characterization of

impulse noise over PLC networks has been reported by

Zimmermann [19] and Degardin et al in [20]. Periodic

impulse noise is predominant in power lines due to inductive

loads being switching in and out at the bridge tap

terminations. Measurements made on fifty kinds of

appliances shows that 95% of these appliances generate

periodic impulse noise [17]. The switching operations of

these appliances generate high di/dt and dv/dt and creates

conducted and radiated disturbance. This noise takes the

form of impulse train. The delay between two successive

impulses is in the range of 5μs to 100 μs. The pulse is

approximated by the sum of damped sinusoids [17] shown in

equation 9 below. The parameters of the damped sinusoids

are estimated from the measurement using genetic

algorithms [23] for 14 gauge is provided in the table below

(9)

Where A1,A2,:Amplitude, ω1,ω2: pulsation, α1 α2,:Damping

ratio.

A1 0.058

A2 0.01

ω1 2π*11*10^6 rad/s

ω2 2π*26*10 ^6 rad/s

α1 5*10^6

α2 2*10^6.

The Power spectrum of periodic impulse noise may now be

obtained and is shown in Fig 7. This PSD used as the noise

component in further computations along with AWGN

0 5 10 15 20 25-18

-16

-14

-12

-10

-8

-6

-4

Frequency (MHz)

Pow

er/

frequency (

dB

/Hz)

Power Spectral Density

Figure 7: Impulse noise PSD

In [24] the effect of impulse noise was analyzed on the

channel capacity by corrupting few tones in a frame. In this

paper impulse noise PSD has been obtained.

The SNR at the receiver is computed from

2))(())((

)()( fH

AWGNfsepowerimpulseNoi

fwerTxSignalpofSNR

(10)

SNR from equation 10 is obtained for the loops described in

Fig 2 using the transfer function H(f) given in the equation

(6). Upstream (US) SNR plots along with transmitted signal

PSD for both cases; without and with impulse noise are

shown in the fig.10 and figure11 for the two loops.

Downstream (DS) SNR plots without and with impulse noise

for the two loops are shown in figure.12 and 13. The tone

loading is obtained from these SNRs using a modified

version of Shannon‟s theorem

]1[log)( 2

i

iiSNR

roundbroundb

(11)

Here SNRi is the SNR in the ith tone whose center

frequency is i*4.3125 KHz, and the gap factor

τ=9.8+6=14.8db. The value of 9.8 assures that a bit error rate

(BER) of 10^-7 would be met in the channel along with a 6db

degradation margin [16].Water filling of energy across all

the tones ensures that the total energy does not exceed the

standards specified limit of +21dbm across all usable tones.

Fine gains across all tones ensure that the surplus energies

are redistributed among the tones.

With the DMT symbol rate is kept at 4000 symbols/sec as

for DSL the total channel capacity can now be obtained by

summing the bits loaded in each sub-channel considering the

usable tones in the up-stream and down-stream transmitted

signal PSD. Channel capacity is given by

bpsbC

i

i )4000)(( (12)



48

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IV. RESULTS AND DISCUSSIONS

Modems supporting VDSL2 for telephone lines may be

reused with changes in the analog hybrid section only.

Simulations were performed to compute the line rates for the

various loops using the Transmit spectral density of VDSL2

standard [12] for US and DS under different load conditions

described below.

(a) Loops with single open BT, five open BTs and ten

open BTs.

(b) More realistic cases of loops with BTs terminated by

typical inductive loads of 600mH.

(c) SNR computations and bit loading pattern for the

loops shown in figure 2 were performed using

equations 9 to 11.

The results for various cases are presented in the

following order

1. As a reference, the US and DS capacities for a plain

loop (fig 2a) are shown in Table 1.

2. For loop2 with open bridge tap, the SNR profiles are

shown in figures 10 and 12 for US and DS.

3. The SNR profiles in the US and DS bands for loop 3

with all BTs open are shown in figures 11 and 13.

The resonance effect of the bridge tap degrades the

SNR, which gets further reduced due to the addition

of impulse noise.

4. Using the bit loading profiles the channel capacities

were computed as above, and shown in table 1 for all

the three loops with open ended terminations at the

bridge taps. An observation is that, SNR is high

enough to support non zero bit loading over a portion

of the stop bands corresponding to US and DS bands.

In this case the gain value for the stop band would be

set to zero to ensure no energy is transmitted in that

band.

We now examine the practical case of loops with bridge

taps that are terminated by inductive loads as found in

residences.

5. As a first example consider the BT in loop 2 (fig 2b)

to be terminated in an inductive load of 600mH and

with impulse noise. The US and DS SNR profiles

obtained are shown in figures 14 and 15.

6. Bit-loading profile for loop 2 US and DS with

inductive load and impulse noise with conjugate and

new PSD are shown in the figure 20 and 21.

7. In the case of loop 3 and 4 (figs 2c, 2d), when the BTs

are terminated in inductive loads of 600mH, the US

and DS SNRs are too low to support any positive tone

loading as shown in column 2 of Table 4. This is

primarily due to impedance mismatch since the

modem is not matched to the new line impedance.

To overcome this we need to ensure that the modems have

switchable impedances in their hybrids to closely match a

variety of line impedances with inductive loads terminated in

their BTs. The hybrid impedances could be switched in based

on a rapid SNR computation done in VDSL2 „Quick rate

adaptation‟ along with analysis to determine the next

impedance to be set in. This scheme along with a capability

to increase the Transmit PSD along with added subbands

nominally would suffice to meet the rate requirements.

8. When the SNRs are already high enough to support a

non-zero bit loading profile a nominal increase in

Transmit PSD would suffice as is evidenced for loop

2 with BT terminated in an inductive load. These

improvements are shown in columns 3 and 5 of

Table 2 & 3.

9. As an example of improved rates obtained by

conjugate matching close to the line look in

impedances, we revisit the cases of loop 3 and loop 4

with their BTs terminated in inductive loads of

600mH. The US and DS SNRs along with bit loading

profiles with impulse noise are shown in figures 16

through 19 for loop 3 and in figures 22 through 25

for loop 4. The rates are tabulated in Table 4 &5.

Note the improvement in rates for loop 3 when the

impedance is changed to a value closer to line look in

impedance of loop 3. Small rate improvements can

now be obtained by a nominal increase in transmit

PSD along with added subbands. This can be seen in

the column 3 & 5 in Table 4 and column 2 & 4 in

Table 5.

0 1000 2000 3000 4000 5000 6000 7000-150

-100

-50

0

50

100

150

Tones

SN

R &

Sig

nal P

SD

noise US: Line length(600mt) with a tap after 550mts

snr without imp.noise

snr with imp.noise

signal PSD

Figure 10: US SNR of loop 2 with and without impulse noise

0 1000 2000 3000 4000 5000 6000 7000-1000

-800

-600

-400

-200

0

200

Tones

SN

R

noise US: Line length(1000mt) with a taps after 100mts


snr with imp.noise

Figure 11: US SNR of loop 3 with and without impulse noise


49

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0 1000 2000 3000 4000 5000 6000 7000-150

-100

-50

0

50

100

150

Tones

SN

R &

Sig

nal P

SD

DS:line length(600mt) with a tap after 550mt

snr without noise

snr with noise

Figure 12: DS SNR of loop 2 with and without impulse noise

0 1000 2000 3000 4000 5000 6000 7000-1000

-800

-600

-400

-200

0

200

Tones

SN

R


snr without noise

snr with noise

Figure 13: DS SNR of loop 3 with and without impulse noise

0 1000 2000 3000 4000 5000 6000 7000-200

-150

-100

-50

0

50

Tones

SN

R &

Sig

nal P

SD

noise US: Line length(600mt) with a tap after 550mts


snr with imp.noise

signal PSD

Figure 14: US SNR of loop 2 with inductive load and with

impulse noise PSD

0 1000 2000 3000 4000 5000 6000 7000-200

-150

-100

-50

0

50

100

Tones

SN

R &

Sig

nal P

SD


snr without noise

snr with noise

singnal PSD

Figure 15: DS SNR of loop 2 with inductive load and with impulse noise

PSD

0 1000 2000 3000 4000 5000 6000 7000-1000

-800

-600

-400

-200

0

200

Tones

SN

R

noise US: Line length(1000mt) with a taps after every 100mts

Figure 16: US SNR of loop 3 with inductive load, with

impulse noise PSD and conjugate

0 1000 2000 3000 4000 5000 6000 7000-1000

-800

-600

-400

-200

0

200

Tones

SN

R


snr without noise

snr with noise

Figure 17: DS SNR of loop 3 with inductive load, with impulse noise PSD

and conjugate

0 1000 2000 3000 4000 5000 6000 70000

2

4

6

8

10

12

14

16

18

tones

bit

patt

ern

for

uplo

adin

g

bit loading pattern for uploading for vdsl

Figure 18: Bit-loading profile for loop 3 US with inductive

load and impulse noise with conjugate

0 1000 2000 3000 4000 5000 6000 70000

1

2

3

4

5

6

7

8

tones

bit

patt

ern

for

dow

nloa

ding

bit loading pattern for downloading for vdsl with inductive load

Figure 19: Bit-loading profile for loop 3 DS with inductive

load and impulse noise with conjugate



50

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0 1000 2000 3000 4000 5000 6000 70000

5

10

15

20

25

30

35

40

45

tones

bit p

att

ern

for

uplo

adin

g



load and impulse noise with Conjugate and

new PSD

0 1000 2000 3000 4000 5000 6000 70000

5

10

15

20

25

30

35

40

tones

bit

patt

ern

for

dow

nloa

ding

bit loading pattern for downloading for vdsl

Figure 21: Bit-loading profile for loop 2 DS with

inductive load and impulse noise with

Conjugate and new PSD

0 1000 2000 3000 4000 5000 6000 7000-1000

-800

-600

-400

-200

0

200

Tones

SN

R

noise US: Line length(1000mt) with a taps after every 100mts


snr with imp.noise

signal PSD

Figure 22: US SNR of loop 4 with inductive load,

with impulse noise PSD and conjugate

Figure 23: DS SNR of loop 4 with inductive load, with

impulse noise PSD and conjugate

0 1000 2000 3000 4000 5000 6000 70000

5

10

15

20

25

tones

bit

patt

ern

for

uplo

adin

g

bit loading pattern for uploading for vdsl with inductive load

Figure 24: Bit-loading profile for loop 4 US with

Inductive load and impulse noise with

conjugate

0 1000 2000 3000 4000 5000 6000 70000

2

4

6

8

10

12

14

tones

bit

patt

ern

for

dow

nloa

ding

bit loading pattern for downloading for vdsl with inductive load

Figure 25: Bit-loading profile for loop 4 DS with

Inductive load and impulse noise with conjugate

0 1000 2000 3000 4000 5000 6000 70000

5

10

15

20

25

30

35

tones

bit p

att

ern

for

uplo

adin

g



Load and impulse noise with Conjugate (Zo=1+0.1i) and New PSD

increased 10dB

0 1000 2000 3000 4000 5000 6000 70000

5

10

15

20

25

30

35

40

tones

bit

patt

ern

for

uplo

adin

g



Load and impulse noise with

Conjugate (Zo=0.6+0.1i) and New

PSD increased 10dB


51

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TABLE 1: CAPACITY ESTIMATION FOR DIFFERENT

LOOP S (WITHOUT LOAD).

TABLE 2: CAPACITY ESTIMATION FOR LOOP2

(US)WITH INDUCTIVE LOAD

TABLE 3: CAPACITY ESTIMATION FOR LOOP2

(DS)WITH INDUCTIVE LOAD

TABLE 4: CAPACITY ESTIMATION FOR LOOPS 3,4

(US)WITH CONJUGATE

TABLE 5: CAPACITY ESTIMATION FOR LOOPS 3,4 (DS) WITH

CONJUGATE

V. CONCLUSION

Recognizing that broadband distribution within a residence

is gaining importance with the maturing of telephone line

based DSL, in this paper we analyze the performance of an

indoor power line (AWG14) with upto ten BTs, using ABCD

matrix channel model based on two- port network theory.

SNR profiles and bit-loading

profiles have been computed

using VDSL2 masks that

Line

Topolo

gy

US

capacity

without

impulse

noise

US

capacit

y with

impuls

e noise

PSD

DS

capacity

without

impulse

noise

DS

capacit

y with

impuls

e noise

PSD

Loop 1 111.744

Mbps

59.469

Mbps

160.588

Mbps

87.444

Mbps

Loop 2

BT

open

102.420

Mbps

50.978

Mbps

134.528

Mbps

65.112

Mbps

Loop 3

BT

open

71.736

Mbps

24.420

Mbps

85.92

Mbps

52.776

Mbps

Line

Topolo

gy

US rates

with

inductiv

e load

US rates

with

inductive

load and

imp. noise

US

rate

with

and

conju

gate

imped

ance

US rate

with new

VDSL2

PSD and

conjugate

impedanc

e

Loop 2 2.673

Mbps

0 Mbps 78.10

4

Mbps

121.564

Mbps

Line

Topology

DS

capac

ity

with

induc

tive

load

DS

rates

with

induct

ive

load

and

imp.

noise

DS

rate

with

and

conju

gate

imped

ance

DS rate

with new

VDSL2

PSD and

conjugate

impedance

Loop 2 12.40

8

Mbps

1.116

Mbps

77.048

Mbps

168.388

Mbps

Line

Topology

US &

DS

rates

with

indu

ctive

load

only

US

rates

with

conjuga

te

Impeda

nce

Zo in

Hybrid

With

load

US rates

with

inductive

load and

imp.

Noise

PSD and

conjugat

e

Impedan

ce

Zo in

Hybrid

US rates

with new

VDSL2

PSD

increase

by 10dB

in

transmit

bands

and

conjugat

e

Impedan

ce

Zo in

Hybrid

Loop 3

Zo=1+0.1i

0

Mbps

14.184

Mbps

4.780

Mbps

18.468 Mbps

Loop 4

Zo=1+0.1i

0

Mbps

51.05

Mbps

16.577

Mbps

33.081

Mbps

Loop 3 Zo=0.6+0.1i

0

Mbps

32.676

Mbps

9.276

Mbps

43.464 Mbps

Line

Topology

DS rates

with

conjugate

Impedanc

e

Zo in

Hybrid

With load

DS rates

with

inductive

load and

imp. Noise

PSD and

conjugate

Impedanc

e

Zo in

Hybrid

DS rates

with new

VDSL2

PSD increase

by 10dB in

transmit

bands and

conjugate

Impedanc

e

Zo in

Hybrid

Loop 3

Zo=1+0.1i

9.432

Mbps

1.118

Mbps

1.176

Mbps

Loop 4

Zo=1+0.1i

52.88Mbps 19.472

Mbps

19.472

Mbps

Loop 3

Zo=0.6+0.1i

32.14

Mbps

7.224

Mbps

7.188

Mbps

Loop 3

Zo=0.1+0.1

i

81.568 Mbps

40.104

Mbps

58.820

Mbps



52

Published By:




employ DMT line code along with the presence of dominant

Impulse noise apart from AWGN. The current Transmit PSD

suffices for open ended BTs. However BTs when inductive

loaded result in severe shortfall in data rates due to mismatch

between line impedance and characteristic impedance. Data

rates are shown to be considerably improved by adopting

settable values of conjugate impedances in the hybrid of the

modem that match with the line look-in impedance. We thus

recommend a combination of settable hybrid impedances and

a capability to nominally increase Transmit PSD along with

added subbands to achieve desired rates. This method has a

distinct advantage in that it reuses the entire digital portion

of existing ADSL and VDSL2 modems.

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AUTHOR PROFILE

First Author Profile

Mrs.Usha Rani .K .R, Associate Professor,

R.V.College of Engg.Bangalore,Karnataka, India,

email: [email protected]

PPPofessor

ngalore

International conferences:

1. UshaRani.K.R,Dr.S.Ravishankar,Dr.H.M.Mahesh, "Analysis of

noise and load effects on broadband performance over residential

power lines employing VDSL2" India Conference (INDICON

2011), Annual IEEE, 16-18 Dec.2011, BITS, Hyderabad.

2. Usha Rani.K.R,Dr.S.Ravishankar, " Performance Analysis for

Broadband over residential power lines using VDSL2 Profiles" in

the proceedings IEEE International Conference Signal processing,

Communication & Computing (ICSPCC2011), 14th -16th

Sept.2011, Xi'an, China.

3. Usha Rani.K.R,Dr.S.Ravishankar,Dr.H.M.Mahesh,M.Bharathi,

"An Analysis of Broadband Capacities with Impulse Noise over

Residential Power lines" in the proceedings International

Conference on advances in Recent Technologies in Communication

and Computing 2010,( ARTCom 2011), 14th-15th Sept.2011,

Bangalore.

4. UshaRani.K.R,Dr.S.Ravishankar,Dr.H.M.Mahesh,M.Bharathi,

"Analysis of Power Line Networks for Broadband Transmission" in

the proceedings International Conference on advances in Recent

Technologies in Communication and Computing 2010,( ARTCom

2010).,Kottayam, Kerala, 15th-16th October 2010

5. M.Bharathi,Dr.S.Ravishankar, Usha Rani.K.R "Frequency

Domain Reflectometry based SELT Approach for Loop Topology

Estimation" in the proceedings International Conference on

advances in Recent Technologies in Communication and

Computing 2010 ( ARTCom 2010).,Kottayam, Kerala, 15th-16th

October 2010

6. M.Bharathi,Dr.S.Ravishankar, Usha Rani.K.R "Capacity

Estimation of ADSL Line Using Frequency domain based SELT

technique" in the proceedings International conference on advances

in computing, control and telecommunication technologies

,Organized by ACEEE , Kerala, Dated Dec 26-28 2009.

7. M.Bharathi,Dr.S.Ravishankar, Usha Rani.K.R "Frequency domain

SELT approach for line length estimation", Second International

conference on signal and image processing, Vidya Vikas Institute of

Technology, Mysore, Dated 12-14 Aug 2009.

National conferences:

1. UshaRani.K.R,Dr.S.Ravishankar,M.Bharathi 'Review of channel

modeling for power line communication', National conference on

Advances in communication and computing, Karpagam college of

Engineering, Coimbatore, Dated 18th Sep 2009.


53

Published By:




Second Author Profile

Dr. S Ravi Shankar, Professor, R.V.College of Engg.,Bangalore,

Karnataka,India,

email: [email protected]

sor

M.Tech IIT Kharagpur, Ph.D IIT Madras

Papers Published in National and International

Journals:

1. Narasimhan, M.; Ravishankar, S,”Radiation from aperture

antennas radiating in the presence of a dielectric sphere”, IEEE

Transactions on Antennas and Propagation, Volume: 30 Issue: 6

Nov 1982, Page(s): 1237- 1240

2. Narasimhan, M.; Ravishankar, S,” Probe uncompensated near-field

to far-field transformation for scanning over an arbitrary surface”,

IEEE Transactions on Antennas and Propagation, Volume: 33

Issue: 4 Apr 1985, Page(s): 467- 472

3. Narasimhan, M.; Ravishankar, S. “Multiple scattering of EM waves

by dielectric spheres located in the near field of a source of

radiation”, IEEE Transactions on Antennas and Propagation,

Volume: 35 Issue: 4 Apr 1987, Page(s): 399- 405

4. Ravishankar, S.; Biswagar, Prakash “Analysis of Dielectric Lens -

Adaptive Array Antennas for Shaped Beam Applications”, Sarnoff

Symposium, 2006 IEEE 27-28 March 2006, Page(s): 1-4 Digital

Object Identifier 10.1109/SARNOF.2006.4534746

5. S.Ravishankar & M.S.Narasimhan, “The Effect of probe

Directivity in spherical Near Field Antenna Measurements” April –

June 1982, Vol 2, No 2, Electromagnetics Journal.

6. S.Ravishankar “EM Scattering by bodies of arbitrary shape and

unknown Constitution” April 1993. EMC Test and Design, IEEE

EMC/ ESD Conference Proceedings, Denver Colorado.

International and National Conference:

1. Ravishankar, S.; Rukmini, T.S.; Kumaraswamy, H.V.; Sunit, S.;

Karthik, Y.; Vinay, P.; Ravishankar, A,”Analysis of hemispherical

Dielectric lens antennas for wireless applications”, Applied

Electromagnetics, 2007. APACE 2007. Asia-Pacific Conference on

4-6 Dec. 2007, Page(s): 1-6. Digital Object Identifier

10.1109/APACE.2007.4603917

2. Ravishankar, S.; Padmaja, K.V.; Uma, B.V, “Implementation of

Novel Bit Loading Algorithm with Power

Cut-Back in ADSL Transmitter on DSP”, International Conference

on Computational Intelligence and Multimedia Applications, 2007,

Volume: 1 13-15 Dec. 2007, Page(s): 570-574, Digital Object

Identifier .1109/ICCIMA.2007.139

3. Ravishankar, S.; Uma, B.V.; Shreeprasad, M, “Rate-reach

performance improvements in ADSL

interference environment using adaptive wavelet threshold”,

International Conference on Wavelet Analysis and Pattern

Recognition, 2008. ICWAPR '08, Volume: 2 30-31 Aug. 2008,

Page(s): 634-638, Digital Object Identifier

10.1109/ICWAPR.2008.4635856

4. Ravishankar., S; Padmaja., K.V; Sridhar, Santosh,

“Implementation of dual latency operation in VDSL2 with

downstream power back off on DSP chip”, International Conferece

on Signals and Electronic Systems, 2008. ICSES '08, 14-17 Sept.

2008 Page(s): 427-430 Digital Object Identifier 0.1109

/ICSES.2008.4673456

5. Ravishankar, S.; Uma, B. V.; Shreeprasad, M. “Application of

wavelet for improvement of rate-reach performance in ADSL

interference environment”, International Conference on

Communication

Technology, 2008. ICCT 2008. 11th IEEE 10-12 Nov. 2008,

Page(s): 565-568 Digital Object Identifier

10.1109/ICCT.2008.4716124

6. Ravishankar, S.; Uma, B.V.; Padmaja, K.V,”Realization of

Adaptive Filters for NEXT Crosstalk Mitigation in Two Wire

ADSL on DSP”,Conference on Computational Intelligence and

Multimedia Applications, 2007. International Conference on

Volume: 1 13-15 Dec. 2007, Page(s): 567-569

Digital Object Identifier 10.1109/ICCIMA.2007.136

7. Dr. Ravishankar S, Mahesh A," Spherical Mode Analysis of patch

array antenna", Conference on IEEE sponsored International

conference on Antenna & Wave propagation, Barcelona, Spain,

April 12-16 2010.

8. Dr. Ravishankar S, Mahesh A," Gain enhancement of micro strip

array antenna with dielectric lens antenna: comparative study",

Conference on IEEE-AEMC, kolkata, India, Dec-14-16 2009.

9. Dr. Ravishankar S, Mahesh A," Design of 2X2 microstrip patch

array antenna embedded in hemispherical dielectric lens for

airborne mobile communications", Conference on IEEE Sponsored

International Radar symposium IRSI 2009, India, Dec 8-11, 2009.

10. Dr. Ravishankar S, S.K.Thakur ,Mahesh A," Collimation

Properties of Micro strip Patch Fed Dielectric Lens Antenna for

Broadband Mobile Communication", Conference on IEEE

sponsored International Conference on Future Computer and

Communication, Kuala Lumpur, Malaysia, Pages 522-526 Year of

Publication: 2009 ISBN:978-0-7695-3591-3, India, Dec 8-11,

2009.

11. Dr. Ravishankar S, M.Bharathi,Usha Rani.," Capacity Estimation

Using Frequency Domain Based SELT", Conference on Advances

in Computers Control and Telecommunication Technologies (ACT

2009), 26-28 Dec 2009.

12. Dr. Ravishankar S, M.Bharathi,Usha Rani.," Frequency domain

SELT approach for line length estimation", Conference on Second

International conference on signal and image processing, Vidya

Vikas Institute of Technology, Mysore, 12-14 Aug 2009.

International Workshop:

1. Ravishankar, S, “Analysis of shaped beam dielectric lens antennas

for mobile broadband applications”, IEEE International Workshop

on Antenna Technology: Small Antennas and Novel Metamaterials,

2005. IWAT 2005, 7-9 March 2005 , Page(s): 539- 542, Digital

Object Identifier 10.1109/IWAT.2005.1461135.

International Symposium:

1. Ravishankar, S.; Biswagar, P, “Spherical modal analysis of shaped

dielectric lens antennas for mobile broadband applications (MBS)”

Antennas and Propagation Society International Symposium, 2005

IEEE Volume: 2A 3-8 July 2005, Page(s): 454- 457 vol. 2A

2. S.Ravishankar & M.S.Narasimhan, “Near Field to Near Field

Transformations - Spherical Modal Approach” IEEE International

Radar Symposium India - 83, Oct 1983.

3. S.Ravishankar, “The Effects of antenna Mounted Bodies on the

Radiation from VSAT Earth Station Antennas”, Sept 1993, The

Third IEEE/URSI International Symposium on antennas and EM

Theory, NANJING, Peoples Republic of China.

Patents:

1. US Patent 10/ 684363 filed Oct 15, 2003, S.Ravishankar and Satya

Sudhakar – Assignee Texas Instruments,“Determining distance to

echo points in a wire line medium”.

2. US Patent 10/ 757987 filed Jan 16, 2004, S.Ravishankar –

Assignee Texas Instruments

“Antennas supporting High Density of wireless users in specific

directions”.

3. Indian Product patent, 4592/416/MAS/2000 dated June 2000,

S.Ravishankar, Satcom

Lab ITI, M: N Redundancy switches for Modems and Converters

with transponder and

Polarization Hopping”.

mailto:[email protected]

Documents

Broad band Transmission over Residential Power Lines