CHAPTER 9 Operational-Amplifier andAmplifier and Data …ocw.snu.ac.kr/sites/default/files/NOTE/1678.pdf · 2018. 1. 30. · 2007 Fall: Electronic Circuits 2 CHAPTER 9 Operational-Amplifier

2007 Fall: Electronic Circuits 2

CHAPTER 9Operational-Amplifier andOperational Amplifier and Data-Converter Circuits

Deog-Kyoon Jeongdkj @ [email protected]

School of Electrical EngineeringS l N ti l U i itSeoul National University

Introduction

In this chapter, we will be covering…

The Two-Stage CMOS Op Ampe o S age C OS Op pThe Folded-Cascode CMOS Op AmpThe 741 Op-Amp CircuitThe 741 Op Amp CircuitD/A ConverterA/D ConverterA/D Converter

9/23/2007 (c) 2007 DK Jeong 2/78

9.1 The Two-Stage CMOS OP AMP

C ti it CCompensation capacitance Cc(together with Cgd6) is Miller-multiplied by the gain of the second stage

Systematic output dc offset

)/()/( LWLW

Systematic output dc offset → can be eliminated by keeping

5

7

4

6

)/()/(2

)/()/(

LWLW

LWLW

⋅=

Figure 9.1 The basic two-stage CMOS op-amp configuration.

9/23/2007 (c) 2007 DK Jeong 3/78


Input Common-Mode RangeTo Keep Q1 & Q2 in saturation

To Keep Q5 in saturation

tpOVtnSSICM VVVVV −++−≥ 3

15 SGOVDDICM

VVVV

VVVV

−−−=

−−≤

Input common-mode range

15 OVtpOVDD VVVV=

Input common mode range

513 OVOVtpDDICMtptnOVSS VVVVVVVVV −−−≤≤−++−

9/23/2007 (c) 2007 DK Jeong 4/78


Output SwingTo keep Q6 & Q7 saturated,

We need to keep the magnitude of Vo

76 OVDDOOVSS VVVVV −≤≤+−

We need to keep the magnitude of Vovas low as possible

However, counteracted by the need to have high fT for Q6

fT is proportional to Vov (in Section 6.2.3)

9/23/2007 (c) 2007 DK Jeong 5/78


Voltage Gain

Figure 9.2 Small-signal equivalent circuit for the op amp in Fig. 9.1.

)2/(2211

IIggG

Rin

=⋅

===

∞=

and,|| 442

2421

11211

VrV

rwhererrR

VVggG

AA

OVOVmmm

===

9/23/2007 (c) 2007 DK Jeong 6/78

2/ and

2/ ,|| 42421 I

rI

rwhererrR oooo


Voltage Gain (cont)

Figure 9.2 Small-signal equivalent circuit for the op amp in Fig. 9.1.

6

662

2

OV

Dmm V

IgG ⋅==

6

7

7

77

6

66762

6

,||D

A

D

Ao

D

Aooo

OV

IV

IV

randIVrwhererrR ====

9/23/2007 (c) 2007 DK Jeong 7/78

676 DDD


Voltage Gain (Cont.)

111 m RGA −= 222 m RGA −=

421

2)||( oom

V

rrg−= 7662

)||( oom

V

rrg−=

42

1

11

AA

OV

VV

V

+−=

76

6

11

AA

OV

VV

V

+−=

9/23/2007 (c) 2007 DK Jeong 8/78

42 AA VV 76 AA VV


Voltage Gain (Cont.)

The overall dc voltage gain

)||()||( 2211

21

mm

v

rrgrrgRGRG

AAA

===

Output resistance of the op amp)||()||( 766421 oomoom rrgrrg=

|| rrR

9/23/2007 (c) 2007 DK Jeong 9/78

76 || ooo rrR =


Frequency Response

Capacitance C1, C2

C C C C C C

&

1 2 2 4 4 6

2 6 7 7

gd db gd db gs

db db gd L

C C C C C C

C C C C C

= + + + +

= + + +

Pole & Zero (in Section 7.7.1)

: dominant pole: dominant pole11

2Pf

R G R C≅1

1 2 2

22

2

2

Pm C

mP

fR G R C

GfC

π

π≅

2

2

2

2m

Z

CGf

C

π

π≅

9/23/2007 (c) 2007 DK Jeong 10/78

2 CCπ


Frequency Response

To guarantee stability, unity-gain frequency ft must be

lower than fp2 & fz, so that 20log|Av| crosses 0 db at its

-20db/dec decaying section20db/dec decaying section

1m ffGfAf


Simplified Equivalent Circuit

Figure 9.3 An approximate high-frequency

equivalent circuit of the two-stage op amp.

This circuit applies for frequencies f >> fP1.

Based on the assumption that |A2| is large and a virtual ground appears at the input terminalappears at the input terminal

The second stage effectively acts as an integrator that is fed with the output current signal of the first stage : Gm1Vid

9/23/2007 (c) 2007 DK Jeong 12/78


Phase Margin Excess Phase Shift

)(tan2

12

P

tP f

f−−=φ

)(tan 1z

tz

ffff−−=φ : right half plane zero

)(tan)(tan90 12

1

z

t

P

ttotal f

fff −− ++°=φ

180PM total−°= φ

Phase Margin

)(tan)(tan90 12

1

z

t

P

t

total

ff

ff −− −−°=

φ

9/23/2007 (c) 2007 DK Jeong 13/78

Figure 9.4 Typical frequency response of the two-stage op amp.


Phase Margin

Problem: Additional phase lag by zero

Solution: Include R in series with CC. The transmission zero can be Cmoved to other less-harmful locations

9/23/2007 (c) 2007 DK Jeong 14/78


Phase Margin

0V = 1

1

0

222 VG

R

VV

imi

o

=+

=By selecting . ,

1Z

2

∞== fG

Rm

1R

)1(

1

RCs

sCR

C

=

+

2

1

mGR >By selecting

fz is at a negative real-axis: the extra phase

9/23/2007 (c) 2007 DK Jeong 15/78

)(2

RG

Cm

C − adds to the phase margin


Slew Rate

1V applied at the inputA large signal will exceed the voltage required to turn off one side of the pairg g g q pand switch the entire bias current I to the other side

tCItvo =)( CC

o )(

C

mt

OmmtOVOVt

C CGf

VIgGVVf

CISR

πωπ

2, since ,2 1

111 ======

9/23/2007 (c) 2007 DK Jeong 16/78

COVC CVC π21


Example 9.1 ( ) ( )( )1 2 4 6 6 7

2 2 21 1v m o o m o oA g r r g r r

I IV V

=

( )( )

6

6

2

2 2 21 1 2 2 2

DA A

OV OV D

A

I IV VV I V I

VV

= × × × × ×

⎛ ⎞= ⎜ ⎟⎝ ⎠

v A

2

To obtain A =4000, given V =20V,4004000 0.316V

OV

OV

V

VV

⎝ ⎠

= ⇒ =OVV

1 2

' 2 21

To obtain the required (W/L) ratios of Q and Q ,1 1100 80 0.3162 2D p OV

W WI k VL L

⎛ ⎞ ⎛ ⎞= ⇒ = × ×⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠1 1

1 2

2 225 25 and 1 1

For Q3 and Q4

p L LW m W mL m L m

μ μμ μ

⎝ ⎠ ⎝ ⎠

⎛ ⎞ ⎛ ⎞∴ = =⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠

2

3 3 4

For Q3 and Q41100 200 0.3162

W W WL L L

⎛ ⎞ ⎛ ⎞ ⎛ ⎞= × × ⇒ =⎜ ⎟ ⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠ ⎝ ⎠

2

101

1 50F Q 200 80 0 316

mm

W W m

μμμ

=

⎛ ⎞ ⎛ ⎞⎜ ⎟ ⎜ ⎟

9/23/2007 (c) 2007 DK Jeong 17/78

25

5 5

50For Q , 200 80 0.3162 1

W W mL L m

μμ

⎛ ⎞ ⎛ ⎞= × × ⇒ =⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠


Example 9.17

7 5

Since Q is required to conduct 500 A,1252.5

1W W mL L m

μμμ

⎛ ⎞ ⎛ ⎞= =⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠7 5

26

6

1For Q , 500 200 0.3162

50

WL

W

μ⎝ ⎠ ⎝ ⎠⎛ ⎞= × ×⎜ ⎟⎝ ⎠

⎛ ⎞

6

REF

50 1

Finally, let's select I =20 A, thus

W mL m

μμμ

⎛ ⎞⇒ =⎜ ⎟⎝ ⎠

8

0.1W WL L

⎛ ⎞ ⎛ ⎞=⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠5

51

mm

μμ

=

9/23/2007 (c) 2007 DK Jeong 18/78


Example 9.13 | 5| | 1|

1.33 0.52

6 | 7|

The input resistance is practically infinite,

and the output resistance is

1 20 6 7

12

200.5

20

9/23/2007 (c) 2007 DK Jeong 19/78


Example 9.1 P26

To determine f2 2 0.5 3 2 /DIG g mA V×= = = =2 6

32

2 122

3.2 /0.316

3.2 10 6372 2 0.8 10

m mOV

mP

G g mA VV

Gf MHzCπ π

−

−

= = = =

×⇒ ≅ = =

× ×

32

To move the transmission zero to s=1 1 316

3.2 10mR

G −

∞

= = = Ω×

oFor a phase margin of 75 , the phase shift due tothe ot

1

second pole at f=f , must be 15 , that is,

t 15 637 t 15 171o otf f MH− ⇒12

tan 15 637 tan15 171o ot tP

f f MHzf

= ⇒ = × =

11 1

2 100 where 0.63 /2 0.316V

mC m m

t

G AC G g mA Vf

μπ

×= = = =

3

6

6

0.63 10 0.62 171 10

2 2 171 10 0.316 340

t

C

t OV

f

C pF

SR f V V sπ

π π μ

−×⇒ = =

× ×= = × × × =

9/23/2007 (c) 2007 DK Jeong 20/78

t OV

9.2 The Folded-Cascode CMOS OP AMP

Q Q i t diff ti l iQ1, Q2: input differential pairQ3, Q4: cascode transistorsEach of Q1, Q2 is operating at aEach of Q1, Q2 is operating at abias current (I/2)The bias current of each of Q3,Q4 is (I I/2)Q4 is (IB – I/2)The cascode current mirror Q5to Q8: for high output resistance

Figure 9.8 Structure of the folded-cascode CMOS op amp.g p p

CL: the total capacitance at the output nodeThe load capacitance contributes to frequency compensation

9/23/2007 (c) 2007 DK Jeong 21/78

The load capacitance contributes to frequency compensation


Input common-mode range and the output voltage

swing Assuming that Q9 and Q10 are operated at the edge of saturationp g

tnOVDDICM VVVV +−= 9max ||

tnOVOVSSICM

tnOVDDICM

VVVVVVVVV

+++−= 111min9max ||

SGOVDDBIAS VVVV −−= 4101 ||

Figure 9.9 A more complete circuit for the folded-cascode CMOS amplifier of Fig. 9.8. tnOVOVSSo

OVOVDDo

VVVVvVVVv+++−=

−−=

57min

410max ||||

9/23/2007 (c) 2007 DK Jeong 22/78


Input common-mode range and the output voltage

swing

Figure 9.10 Small-signal equivalent circuit of the folded-

cascode CMOS amplifier. Note that this circuit is in effect

an operational transconductance amplifier (OTA)an operational transconductance amplifier (OTA).

)2/(2

1121

OVOVmmm V

IVIggG ====

)(||)]||)([(|| 8661024464 oomooomOOO rrgrrrgRRR ==

9/23/2007 (c) 2007 DK Jeong 23/78

)}(||)]||)({[( 866102441 oomooommOmV rrgrrrggRGA ⋅==


Frequency response

OL

Om

id

O

RsCRG

VV

+=

1 OLid

1The dominant pole has a frequency fP

OLP RC

fπ2

1=

The unity-gain frequency ft

mGfRGfL

mPOmt C

fRGfπ2

==

9/23/2007 (c) 2007 DK Jeong 24/78


Slew rate

When IB>I, the current that will flow into CL will be I4-I6 = IB-(IB-I) = I

I12 OVt

L

VfCISR π==

9/23/2007 (c) 2007 DK Jeong 25/78


Example 9.2

22 2 20A DiD D V II I W⎛ ⎞⎜ ⎟ ' 2, , 0.25

A DiD Dm o

iOV D D OV

g rV I I L k V

⎛ ⎞= = = = =⎜ ⎟⎝ ⎠

9/23/2007 (c) 2007 DK Jeong 26/78


Example 9.2 Note that for all transistors, 160V/V, 1.0Vm o GSg r V= =

11 1 9

1.25V 3V

m o GS

SS OV OV tn ICM DD OV tn

ICM

V V V V V V V VV

− + + + ≤ ≤ − +

⇒ − ≤ ≤

7 5 10 4

1.25V 2V

ICM

SS OV OV tn o DD OV OV

o

V V V V v V V Vv

− + + + ≤ ≤ − −

⇒ − ≤ ≤

( )( )4 4 4 2 10 16o

o m o o oR g r r r≅ = ( )0 200 80 9.1421 28

M

R g r r M

= Ω

≅ = Ω6 6 6 8

4 6

3 6

21.286.4

0 8 10 6 4 10 5120V/V

o m o o

o o o

R g r r MR R R M

A G R −

≅ = Ω

∴ = = Ω

∴ × × ×

9/23/2007 (c) 2007 DK Jeong 27/78

0.8 10 6.4 10 5120V/Vv m oA G R∴ = = × × × =


Example 9.23

12

0.8 10 25.52 2 5 10

mt m o P

L

Gf G R f MHzCπ π

−

−

×= = = =

× ×2 2 5 1025.5 5

5120

L

tp

Cf MHzf kHzA

π π

= = =

6

12

5120

200 10 40V μs5 10

vA

ISRC

−

−

×= = =

×5 10Finally, to determine the power dissipation we notethat the total current is

LC ×

500 A=0 5mA and the totalμthat the total current is 500 A 0.5mA, and the totalsupply voltage is 5V, thus

5 0 5 2 5mWP

μ

= × =

9/23/2007 (c) 2007 DK Jeong 28/78

5 0.5 2.5mWDP = × =


Increasing the input common-mode range:Rail-to-rail input operation

An NMOS and a PMOS differential pair placed in parallel would provide an inputstage with a common-mode range thatexceeds the power supply voltage in bothdirections.

Rail-to-rail input operationEach of the current increments indicated isequal to Gm(Vid/2).

This assumes that both differential pairs

OmVidOmO RGAVRGV 2,2 == Figure 9.11 A folded-cascode op amp that employs twoparallel complementary input stages to achieve rail-to-rail

input common-mode operation. Note that the two “+”

t i l t d t th d th t “ ” t i l

9/23/2007 (c) 2007 DK Jeong 29/78

will be operating simultaneously. terminals are connected together and the two “–” terminalsare connected together.


Increasing the output voltage range:The wide-swing current mirror

2t OVV V+ 2 OVVt OV OV

Figure 9.12 (a) Cascode current mirror with the voltages at all nodes indicated. Note that the minimum

voltage allowed at the output is V + V (b) A modification of the cascode mirror that results in the reduction

9/23/2007 (c) 2007 DK Jeong 30/78

voltage allowed at the output is Vt + VOV. (b) A modification of the cascode mirror that results in the reduction

of the minimum output voltage to VOV. This is the wide-swing current mirror.

9.3 The 741 OP-AMP Circuit

The IC design philosophyM tl t i tMostly transistorsRelatively few resistorsOnly one capacitorThis philosophy is dictated by the economics (silicon area, ease of fabrication, quality of realizable components) of the fabrication of active and passive components in IC form.

Two power supplies (+VCC and –VEE)Normally V V 15VNormally, VCC=VEE=15VBut the circuit also operates satisfactorily with ±5V.

With a relatively large circuit, the first step in the analysis is the identification of its recognizable parts and their functions.

9/23/2007 (c) 2007 DK Jeong 31/78


Figure 9.13 The 741 op-amp circuit. Q11, Q12, and R5 generate a reference bias current, IREF. Q10, Q9, and Q8 bias

the input stage, which is composed of Q1 to Q7. The second gain stage is composed of Q16 and Q17 with Q13B acting

as active load. The class AB output stage is formed by Q14 and Q20 with biasing devices Q13A, Q18, and Q19, and an

i t b ff Q T i t Q Q Q d Q t t t th lifi i t t t h t i it d

9/23/2007 (c) 2007 DK Jeong 32/78

input buffer Q23. Transistors Q15, Q21, Q24, and Q22 serve to protect the amplifier against output short circuits and

are normally cut off.


Bias CircuitQ Q R : I (reference bias current)Q11, Q12, R5: IREF (reference bias current)Q11, Q10, R4: Widlar current sourceQ8, Q9: Current mirror, Q12,Q13A, Q13B : Current mirrorQ18 Q19: 2VBE drops between Q14 and Q20Q18, Q19: 2VBE drops between Q14 and Q20

Short-Circuit Protection CircuitryR6, R7, Q15, Q21, Q24, R11: normally off

9/23/2007 (c) 2007 DK Jeong 33/78

R6, R7, Q15, Q21, Q24, R11: normally off


The input stage (Differential stage)p g ( g )Q1~Q7 with biasing performed by Q8~Q10 Q1, Q2 : Emitter follower, Rin high Q3, Q4 : Common base amp, Level shifter, Q1,Q2 protection

(npn: Breakdown 7V, pnp: Breakdown 50V)Q5, Q6, Q7, R1, R2, R3 :

Load circuit of input stageCurrent mirror (High resistance)

9/23/2007 (c) 2007 DK Jeong 34/78

Current mirror (High resistance),Differential >> Single ended (Q6 Collector)


The second stage (Single ended high gain stage)Intermediate stage: Q16, Q17, Q13B, R8, R9Q16 : Emitter follower(High Rin), Q17 : Common emitter ampQ16 : Emitter follower(High Rin), Q17 : Common emitter ampQ13B : Active load (High gain), CC: Frequency compensationDominant pole : 4Hz, unity gain bandwidth : 1MHz

9/23/2007 (c) 2007 DK Jeong 35/78


The output stage (Buffering stage)Class AB output stage, Low Rout, Large load currentQ14, Q20 : Complementary pairQ14, Q20 : Complementary pairQ18, Q19 are fed by Q13A and bias Q14, Q20Q23 : Emitter follower (minimizing loading effect on second stage)

9/23/2007 (c) 2007 DK Jeong 36/78


The output stage (buffering stage)Class AB output stage is utilized in 741 Op-Amp

Figure 9.14 (a) The emitter follower is a class A output stage. (b) Class B output stage. (c) The output of

a class B output stage fed with an input sinusoid. Observe the crossover distortion. (d) Class AB output stage.

9/23/2007 (c) 2007 DK Jeong 37/78


Device parameters

1410 A, 200, 125 VS AI Vβ−= = =

1410 A, 50, 50 VS AI Vβ−= = =

For NPNFor PNP

9/23/2007 (c) 2007 DK Jeong 38/78

14 140.25 10 A, 0.75 10 ASA SBI I− −= × = ×For Multi-collector PNP

9.4 DC Analysis of the 741

Reference bias current

12 1111 12

5

( ) 0.73 ( 15 , 0.7 )CC EB BE EEREF CC EE BE EBV V V VI mA V V V V V V

R− − − −

= = = = = ≅5

Input-stage bias

Widlar Current Source

11 10 10 4

10 4 10 11 10ln ( ) 19

BE BE C

REFT C S S C

V V I RIV I R I I I AI

μ

− =

= = ⇒ =10CI

9/23/2007 (c) 2007 DK Jeong 39/78

Figure 9.15 The Widlar current source.


From symmetryIf the npn β is high

21 CC II =III ≅If the npn β is high

Q3 and Q4 base current :

III BB ≅= 43

PP

IIββ

≅+1

Using the result in Eq.(6.21)Q8 and Q9 current mirror :

PC

IIβ/21

29 +=

Node X : if

Pβ/21+

uAIIIIIuAII

CCCC

CP

5.9192,1

4321

10

==≅==∴=≅>>β

Q1 through Q4,Q8 and Q9 : negative feedback loop

Figure 9.16 The dc analysis of the 741 input stage.

To stabilize 2/10CII ≅

↓↓

⇒↑⇒↑⇒↑ III CC)(/98

β

9/23/2007 (c) 2007 DK Jeong 40/78

↓⇒↓ IconstII CP )(/2 10β


If neglect the base current of Q16,Q7

Q7 bias current

III CC ≅= 56

IRVI2

uAVIVV

RIRVIII BE

NEC

5175.9l25l

2

3

2677

++=≅

β

uAI

mA

mVI

VV

C

STBE

5.10

51710

ln25ln

7

146

=⇒

=== −

Input Bias and Offset Currents

Figure 9.17 The dc analysis of the 741 input stage, continued.

p1 2 9.5 / 200 47.5

2B B

BN

I I II nAβ

+= = = =

9/23/2007 (c) 2007 DK Jeong 41/78

1 2input offset current OS B BI I I= −


Input offset voltageVOS : differential input voltage to reduce the output current to zero

Input common-mode rangeInput common mode rangeInput stage remains in the linear active modeThe upper end by saturation of Q1, Q2Th l d b t ti f Q Q

9/23/2007 (c) 2007 DK Jeong 42/78

The lower end by saturation of Q3, Q4


Second-stage bias (Fig. 9.13)Second stage bias (Fig. 9.13)

13 17

17

0.75 (by Emitter Area Ratio)=550 A=

l 618

C B REF C

C

I I IIQ

μ=

1717 17 17 17 8 17

1716 16 16 17

ln 618 ,

16.2

CBE T B E BE

S

BC E B

IQ V V mV V I R VI

VQ I I I Aμ

⇒ = = = +

⇒ ≅ = + =

9/23/2007 (c) 2007 DK Jeong 43/78

16 16 16 179

16.2C E BQ I I I ARμ⇒ ≅ +


If neglect the base current of Q14,Q20Output-stage bias

If VBE18 is 0.6V,

uAIII REFEC 18025.02323 =≅≅

60 uAk

IR 15406.0

10 ==

IIuAI CE 16515180 1818 ≅=−=

uAIQin

mVI

IVV

B

S

CTBE

8.0200/165,

588ln

1818

1818

==

≅=⇒

mVI

IVVQin

uAII

S

CTBE

EC

530ln,

8.15

191919

1919

==

=≅⇒

Figure 9.18 The 741 output stage without the IIV

IIVVAlso

VVVV

S

CT

S

CTBB

BEBEBB

lnln,

118.1530588

20

20

14

14

1918

+=

=+=+=

9/23/2007 (c) 2007 DK Jeong 44/78

short-circuit protection devices. uAIIII

CC

SS

15420142014

==⇒

9.5 Small-Signal Analysis of the 741 Input Stage

The Input Stage

dDifferential signal vig iFour emitter resistances connectedin series 4·re

Input resistance

kΩ 2.639.5uA25mV

IVr ,

4rvi Te

e

ie ====

Input resistance

MΩ1.2r)1(β4R eNid =⋅+⋅=

Figure 9.19 Small-signal analysis of the 741 input stage.

9/23/2007 (c) 2007 DK Jeong 45/78


7 : neglectedbi 7 : neglectedbi

5c ei iα=

5 6 5 6, identical : c cQ Q i i=5 6 5 6, c cQ Q

output node : 2o ei iα=

Figure 9.20 The load circuit of the input stage fed by the two

complementary current signals generated by Q1 through Q4 in

Fig. 9.19. Circled numbers indicate the order of the analysis steps.

The transconductance of the input stage

)V/A265/1G1k632(i2iG eo ≅Ωα⋅α⋅

9/23/2007 (c) 2007 DK Jeong 46/78

)V/mA26.5/1G,1,k63.2r(r2ir4v

G 1meeee

e

i

o1m =≅αΩ==⋅⋅

=≡


Output resistance(current source of Q44) || (output resistance of Q) || (output resistance of Q66))

50V V

Virtual ground 4

50 5.269.5

[1 ( // )] 10 5

Ao

V Vr MI A

R r g r r Mμ

= = = Ω

= + = Ωground 4 [1 ( // )] 10.5o o m eR r g r r Mπ+ Ω

0bi ≅

6 2[1 ( // )] 18.2b

o o mR r g R r Mπ= + = Ω

Figure 9.21 Simplified circuits for finding the two

components of the output resistance Ro1 of the first

stage

1 4 6 // 6.7o o oR R R M⇒ = = Ω

9/23/2007 (c) 2007 DK Jeong 47/78

stage.


Small signal equivalent circuit of input stage

Figure 9.22 Small-signal equivalent circuit for the input stage of the 741 op amp.

2.1idR M= Ω

1

1

1/ 5.26 /6.7

m

o

G mA VR M

== Ω

9/23/2007 (c) 2007 DK Jeong 48/78


Example 9.3RΔ

1 2

6

, where 0.02

current in Q I decreasing

RR R R R RRΔ

= = + Δ =

⇒ Δ

5 6

5 6

3

( )( )( )

BE BE

BE BE e

V IR V I I R RV V I R I R R Ir

I R

+ = + −Δ + Δ

− = Δ −Δ + Δ ≅ Δ

Δ Δ 3 5.5 10e

I RI R R r

−Δ Δ∴ = = ×+ Δ +

T d t hIΔ

3

To reduce to zero, we have to apply an input voltage

5.5 10 0 3

OS

IV

I IV mV−

Δ

Δ ×≅

1 1

0.3OSm m

V mVG G

= = ≅

9/23/2007 (c) 2007 DK Jeong 49/78


The second stage

++++= RrRrR eei )])(1(||)[1( 81717916162 ββ

Input resistance by inspection

Ω≅ MRi 42Transconductance

Figure 9.24 The 741 second stage prepared for

2

13

- : short-circuit output current to input voltage( 0)

m

c B

Gi

i i G=

⇒Figure 9.24 The 741 second stage prepared forsmall-signal analysis.

17 2 2

17 9 1717 17 2

17 8 9 17 16

( // ),

( // )

o c m i

b ic b i

e i e

i i G vv R Ri v v

r R R R rα⇒ = =

= =+ +

Figure 9.25 Small-signal equivalent circuit

17 8 9 17 16

17 17 17 8

172

( )( 1)( )

6.5 /

e i e

i e

cm

R r RiG mA V

β= + +

∴ ≡ =

9/23/2007 (c) 2007 DK Jeong 50/78

g g q

model of the second stage.2

2m

iv


The second stage

2- : output resistance of second stageoR

Output resistance

2 13 17

13B

( // )- Q base and emitter grouned

=90 9k

o o B oR R R

R r

=

= Ω

Figure 9.26 Definition of Ro17.

13 13

17 17 17 8 17

16 9 17

=90.9k- [1 ( // )] 787 ( // )

o B o B

o o m

e

R rR r g R r kr R r

π

π

= Ω= + = Ω


The output stageOutput voltage limits- when Q13A is saturated

VVV

when Q is saturated

)VbelowV1(VVVv

CC

14BECEsatCCmaxo −−=

- when Q17 is saturated

)VaboveV51(VVVVv

EE

20EB23EBCEsatEEmino

−+++−=

)VaboveV5.1( EE

Figure 9.28 The 741 output stage.

9/23/2007 (c) 2007 DK Jeong 52/78


The output stageO i it t t ltOpen-circuit output voltageof the second stage

2222 iomo vRGv −=

Figure 9 29 Model for the 741 output stage This model

Second stage voltage gain

2222 iomo

33 ii RGvAFigure 9.29 Model for the 741 output stage. This modelis based on the amplifier equivalent circuit presented in

Table 5.5 as “Equivalent Circuit C.”23

322

2

32

oin

inom

i

i

RRRRG

vvA

+−=≡

Input resistance

RfR Lin )(3 =

VVAMkRkRrQofemittertheinnceresistatotal

negligibleQQofRkrkRrRactiveQassume

bAo

AoLb

/515737474||:

,280,100,

201323

19181320232020

ΩΩ×≅∴Ω=⇒

−Ω≅Ω≅+=

β

βπ

9/23/2007 (c) 2007 DK Jeong 53/78

VVAMkRin /515,7.374 2233 −=Ω=Ω×≅∴ β


The output stageOpen-circuit overall voltage gain of the output stage

1vGR o ≅=→∞= 1v

GRLR2o

3voL ≅=→∞=∞=

Output resistancep

20 14 20 14

20

( ,Q on/Q off or Q off/Q on): negative Q

o LR f Iv active

=⇒ 20

223 23 13

23

: negative Q

1.73


The output stage

Output short-circuit protection

- Output terminal Short-circuited ⇒ Large current ⇒ Burnout of IC

- Short-Circuit Protection⇒ limit the current in the output TR

- In Fig 9.13

limit the maximum current that OP Amp

14 14 15 6 14

15 15 14

If > 20mA(Q ), > 540

Turn on Q , E BE E

C B

I V R I mV

I I

=

⇒ ↑ ⇒ ↓

limit the maximum current that OP Amp can source to about 20mA

9/23/2007 (c) 2007 DK Jeong 55/78

9.6 Gain, frequency response, and slew rate f th 741of the 741

Small-signal gain

Figure 9.31 Cascading the small-signal equivalent circuits of the individual stages for the evaluation of

the overall voltage gain.

2 2

2 2

o i o o

i i i o

v v v vv v v v=

1 1 2 2 2 3 ( // )( )

476.1 ( 526.5) 0.97 243,147 V/V

Lm o o m o vo

L o

RG R R G R GR R

= − −+

= − × − × =

9/23/2007 (c) 2007 DK Jeong 56/78

476.1 ( 526.5) 0.97 243,147 V/V 107.7 dB=

9.6 Gain, frequency response, and slew ratef th 741of the 741

Frequency responseUsing Miller’s theorem in second stage CC, the effective capacitance

2(1 ) 30 (1 515) 15480in CC C A p pF= + = + =A2 : the second-stage gain

This capacitance is quite large, we neglect all other C between Q16 and signal ground

The total R between this node and ground

Th d i t l

1 2( ) (6.7 4 ) 2.5t O iR R R M M M= = Ω Ω = Ω

The dominant pole1 4.1

2p in tf Hz

C Rπ= =

9/23/2007 (c) 2007 DK Jeong 57/78

in t


Frequency response

The unity-gain bandwidth ft

The phase shift at ft is -90°

0 3 243147 4.1 1t dBf A f MHz= = × ≅

p t

The phase margin is 90°

This phase margin is sufficient toprovide stable operation for closed

Figure 9.32 Bode plot for the 741 gain, neglecting

nondominant poles.

loop amp with any value of βp

9/23/2007 (c) 2007 DK Jeong 58/78


The high-gain second stage (Cc)

Figure 9.33 A simple model for the 741 based

The second stage gain is large (The output R of the input stage and the input

on modeling the second stage as an integrator.

The second stage gain is large (The output R of the input stage and the inputR of the second stage have been omitted)

1( )( )( )

o mV s GA sV C

≡ = 1( ) mGA jwj C

=

The magnitude of gain becomes unity at ω=ωt

( )i CV s sC Cj Cω

1mGω = 1tf MHzω= ≅ 1( 1/ 5.26 / 30 )CG mA V and C pF= =

9/23/2007 (c) 2007 DK Jeong 59/78

tCC

ω 12t

f MHzπ≅ 1( 1/ 5.26 / 30 )m CG mA V and C pF


Slew rateThe large input voltage causes the input stage to be overdriven, and its small-signal model no longer applies

Figure 9.34 A unity-gain follower with a large step

Figure 9.35 Model for the 741 op amp when a large

positive differential signal is applied.g y g g p

input. Since the output voltage cannot change

instantaneously, a large differential voltage appears

between the op-amp input terminals.

2( )C

o cC C

i dt Iv t v tC C

= = =∫

1 3 2 4

0 , 10, ,

t V V VQ Q on and Q Q off

++ −> − =

⇒

C C

2 2(9.5 ) 0.63 /30C

ISR V sC p

μ μ= = =1,

mt

C

GC

ω =

9/23/2007 (c) 2007 DK Jeong 60/78

3 6 2c cI I I⇒ = =


Relationship between ft and SR

re: Emitter resistance of each of Q1 through Q4112

4m eG

r=

1, 2T

e mV Ir GI V

= =1 2e m TI V

I SRh f2 4t C T T

thereforeC V V

ω = =

3 63 64 4 25 10 2 10 0.63 / ( 741)T tSR V V s forω π μ−= = × × × × =

9/23/2007 (c) 2007 DK Jeong 61/78

9.7 Data Converters – An Introduction

Digital processing of signals

Convert the signal from analog to digital form and then use digital ICs to perform digital signal processingICs to perform digital signal processing

The digital signal processor can perform a variety of arithmetic and logic operations that implement a filtering algorithm

Analog to digital converter (ADC)Analog to digital converter (ADC)Accept an analog sample and produce an N-bit digital word

Digital to analog converter (DAC)Accept an N-bit digital word and produce an analog sample

9/23/2007 (c) 2007 DK Jeong 62/78


Sampling of analog signalsSample-and-hold

Figure 9.36 The process of periodically sampling an analog signal. (a) Sample-and-hold (S/H) circuit.

The switch closes for a small part (t seconds) of every clock period (T). (b) Input signal waveform.

9/23/2007 (c) 2007 DK Jeong 63/78

The switch closes for a small part (t seconds) of every clock period (T). (b) Input signal waveform.

(c) Sampling signal (control signal for the switch). (d) Output signal (to be fed to A/D converter).


Signal quantizationConsider: 0~10VAssuming that we wish to convert this signal to digital form and thatthe required output is a 4 bit digital signalthe required output is a 4-bit digital signal

0V 0000 V 2102/3V 00016V 100110V 1111

VVresolution32

1510

==

10V 1111

Example: the case of a 6.2V analog level (between 18/3 and 20/3)18/3 (6V)18/3 (6V)

Quantization errorUse of more bits reduces quantization error

9/23/2007 (c) 2007 DK Jeong 64/78


The A/D and D/A converters as functional blocks

Figure 9.37 The A/D and D/A converters as circuit blocks.

9/23/2007 (c) 2007 DK Jeong 65/78


The A/D and D/A converters as functional blocks

Figure 9.38 The analog samples at the output of a D/A converter are usually fed to a sample-

and-hold circuit to obtain the staircase waveform shown. This waveform can then be filteredand hold circuit to obtain the staircase waveform shown. This waveform can then be filtered

to obtain the smooth waveform, shown in color. The time delay usually introduced by the filter

is not shown.

The analog samples at the output of a D/A converter are usually fed to a sample-and-hold circuit to obtain the staircase waveformThis waveform can then be smoothed by a low-pass filter, giving rise to the

9/23/2007 (c) 2007 DK Jeong 66/78

y p , g gsmooth curve in color in Fig.9.38

9.8 D/A Converter Circuits

Basic circuit using binary-weighted resistors

Figure 9.39 An N-bit D/A converter using a binary-weighted resistive ladder network.

bb b

bVbVbVi REFREFREF +⋅⋅⋅++=

1 21 22 2 2

NN

bb bN bit digital word D− = + + ⋅⋅⋅+

DR

V22b

2b

2b

RV2

bR2

bR2

bR

i

REFNN

221REF

N1N21o

=⎟⎠⎞

⎜⎝⎛ +⋅⋅⋅++=

+++= −

9/23/2007 (c) 2007 DK Jeong 67/78

DVRiv REFfoo −=−=⎠⎝


Basic circuit using binary-weighted resistors

The accuracy of the DAC depends on- The accuracy of Vref- The precision of the binary-weighted resistors- The perfection of the switches

Disadvantages- For a large number of bits (N>4) the spread between the smallest and

largest R becomes quite large.hi i li iffi l i i i i i i l- This implies difficulties in maintaining accuracy in R values.

A more convenient scheme exists utilizing a resistive network called the R-2Rgladder

9/23/2007 (c) 2007 DK Jeong 68/78


R-2R Ladders

VIVI

IIII

REFREF

NN

⎟⎞

⎜⎛ ⋅⋅⋅==

=⋅⋅⋅=== −

,,

242

21

1321

bR

VbR

VbR

Vi

RI

RI

NNREFREFREF

o +⋅⋅⋅++=

⎟⎠

⎜⎝

242

,4

,2

21

21

DR

VbbbR

V REFNNREF =⎟⎠⎞

⎜⎝⎛ +⋅⋅⋅++=

222 221

Figure 9.40 The basic circuit configuration of a DAC

utilizing an R-2R ladder network.DVRiv REFfoo −=−=

Because of the small spread in R values, this network is usually preferred to thebinary-weighted scheme discussed earlier, especially for N>4

9/23/2007 (c) 2007 DK Jeong 69/65


A practical circuit implementation

( )2N

NN BE

IV V Rα

⎛ ⎞= + ⎜ ⎟⎝ ⎠

12 N

N NIV V Rα−

⎛ ⎞= +⎜ ⎟⎝ ⎠

1

14 2N N

N NBE BE

I IV R V Rα α−

−⎛ ⎞= + = +⎜ ⎟⎝ ⎠

1 12

N NBE BE N Nif V V I I

− −= ⇒ =

12 4 2NI I I I−

Figure 9.41 A practical circuit implementation of a DAC utilizing

11 2 32 4 2

NNI I I I∴ = = = ⋅⋅⋅ =

9/23/2007 (c) 2007 DK Jeong 70/65

an R-2R ladder network.


Current switches Each of the single-pole double-throw switches in the DAC circuit of Fig.9.41 gcan be implemented by a circuit as that shown in Fig.9.42 for switch Sm

Im: the current flowing in the collector of the mth-bit transistor

Qmr: the reference transistor

If bm>VBIAS Qms turn on, Qmr turn off Im through QmsFigure 9.42 Circuit implementation of switch Sm

in the DAC of Fig. 9.41. In a BiCMOS technology,If bm< VBIAS Qms turn off, Qmr turn onIm through Qmr

in the DAC of Fig. 9.41. In a BiCMOS technology,

Qms and Qmr can be implemented using MOSFETs,

thus avoiding the inaccuracy caused by the base

current of BJTs.

9/23/2007 (c) 2007 DK Jeong 71/78

9.9 A/D Converter Circuits

The feedback-type converter

The comparator circuit provides an output that assumes one of two distinctvalues

Figure 9.43 A simple feedback-type A/D converter.

valuesAn up-down counter is simply a counter that can count either up or downdepending on the binary level applied at its up-down control terminal

9/23/2007 (c) 2007 DK Jeong 72/78


The dual-slope A/D converter

Figure 9.44 The dual-slope A/D conversion method. Note that vA is assumed to be negative.

Close S2 (discharge C, v1=0) Open S2 and switch S1 to vA (I=vA/R flow away from the integrator )

9/23/2007 (c) 2007 DK Jeong 73/78

V1 rises linearly with a slope of I/C=vA/RC, as indicated in Fig. (b)


The dual-slope A/D converterReference to Fig. 9.44(b)

peak AV v=

At the end of this phase, the counter ist t

1T RC=

reset to zeroIn phase II, v1 decreases linearly with aslope of (Vref/RC)When v1 reaches zero, the control logicstops the counterThus the content of the counter, n, atpeak refV V Av⎛ ⎞⎜ ⎟ the end of the conversion process is thedigital equivalent of vA

2

peak ref

T RC=

2 1A

ref

vT TV⎛ ⎞

= ⎜ ⎟⎜ ⎟⎝ ⎠

Avn n⎛ ⎞

= ⎜ ⎟⎜ ⎟

9/23/2007 (c) 2007 DK Jeong 74/78

refref

n nV

= ⎜ ⎟⎜ ⎟⎝ ⎠


The parallel or flash converter

Figure 9.45 Parallel, simultaneous, or flash A/D conversion.

Very fastA rather complex circuit

9/23/2007 (c) 2007 DK Jeong 75/78


The feedback-type converterSuitable for CMOS implementation

Figure 9.46 Charge-redistribution A/D converter suitable for CMOS implementation:

(a) sample phase

(a) sample phase: SB closed, v0=0, sA=vA, Q=-2CvA

(a) sample phase

9/23/2007 (c) 2007 DK Jeong 76/78



(b) hold phase: SB open, Si to GND, SA to Vref, vo=-vA

9/23/2007 (c) 2007 DK Jeong 77/78



(c) charge-redistribution phase(c) charge redistribution phase

S1 to Vref : vo=-vA+Vref/2,- If vA>Vref/2, voVref/2+Vref/4, vo

Documents

CHAPTER 9 Operational-Amplifier andAmplifier and Data …ocw.snu.ac.kr/sites/default/files/NOTE/1678.pdf · 2018. 1. 30. · 2007 Fall: Electronic Circuits 2 CHAPTER 9 Operational-Amplifier