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8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
1/8
Deep-Ocean
Tests
of
an
Acoustic
Modem
Insensitive to Mult1path Distortion
Winfield Hill, Gerald Chaplin, David Nergaard
Sea
Data,
Inc., A Pacer Systems Company
1 Bridge
Street, Newton, Massachusetts,
02158
concept,
design
and ocean
testing of a new low-power
modem
is
presented. The telemetry
system
em
a novel chirp frequency sweep and has
other
fea
allow
operation
in the presence of multi
path
inter
The chirp system uses fsk data modulation and
a carrier sweep starting at 9 or 31kHz, depend
upon the model, to obtain the benefits of frequency
without requiring a frequency synthesizer, multi
or
a
FFT
analyzer.
Intended
for
retrofittable
to
instruments,
the
new
system
is designed for use in
deep
ocean
and
the continental
shelf over dis tances
to
were performed in
about
4000
meters of
using
the
low frequency version. Additional shalIow
are
planned,
including a typical harbor.
acoustic
telemetry
diversity modem
Introduction
multipath
chirp
of data is a commonly desired capa
is not commonly available in undersea oceano
instruments. Although convenience and peace-of
occaisional motivations for these desires, strong
have been
made
for
the
value of this capability(1).
include use in
real-time
operational systems, multi
deployments
(where
it s
impractical
to
wait until the
for the data),
performance
monitoring, repair flexibil
and
expendible instrumentation.
We
report
here on
the
and
initial ocean tests of
a new chirp acoustic teleme
method, which
has
simplicity and reliability properties
for fitting acoustic telemetry data links to existing
designs.
Background
- Ocean Acoustic Telemetry
travelling a substantial distance in
the
suffers from severe amplitude fluctuations
and
phase
Acoustic temporal incoherence may by caused
by multiple
sound
pathways,
bottom
and surface scattering
and moving inhomogeneities in
the
ocean(2). However the
repeated observation
of
such
degradation
has obscured the
fact that sound transmission quality over direct vertical or
slanted pathways (other than in a sound channel) may be
quite
good(3).
Kearney
and Laufer(4)
demonstrated
this
point
while de
livering a paper at Oceans '84, by playing a
cassette
tape
recording
of
voice and music transmitted from 1500 me
ters
depth
to
a
shipboard
recorder; my
memory
is
that
the
primary degradation was due to the use of a very poor cas
sette recorder. Designers of acoustic high-resolution pos
tioning systems have long
taken
advantage of good direct
transmission paths by detecting
the
arrival of
short acou;tic
pulses using narrow-band Q
>
30) filters(5). When used
at
low frequencies (10kHz), these positioning systems re
quire several milliseconds
of
phase coherence in
the
leading
edge
of the
pulse.
Short-range «
400m) acoustic
telemetry
systems have been constructed(6,7) using simple frequency
shift keying (fsk) modulation in the expectation of a reliable
acoustic path, with some success. One system transmits at
the
very slow
rate
of 1 bit-per-second(8) to achieve
up to
a
1000m range.
2.1
Multipath. Despite the good quality of a direct path
or
reliable-acoustic-path signal transmission channel, most
practical underwater acoustic systems must contend with
strong undesired signals
scattered
from the surface or the
bottom. This is especially true when one of the acoustic
transducers is near
the
ocean surface. Superimposing the
surface-scattered signals
upon the direct-path
signal causes
fading and phase instabilities, possibly including
complete
cancellation of the
desired signal for a few milliseconds from
destructive
interference. Therefore, a pulsed one- or two
frequency signal, which begins with
good receive quality,
deteriorates as multi
path
interference arrives.
As
an
example of simple two-frequency fsk
telemetry
per
formance
when
surface and bottom scatter have a strong
influence, consider the experiences
of
Ryerson
at
Sandia
Labs(6). Transmission from a 10m subsurface buoy
with
a
slant
range of 180 to 280 meters to a surface buoy was
desired. Water depth was 200 meters. Optimum perfor
mance was obtained only after a variety of system-tuning
CH2585 8 88 0000
275 <
.l
988 IEEE
~ H e L J . I ~ ~ ~ \e:E:e {MfS C o J ; \ - e ~ a ,
OCellV\S '
66
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
2/8
were made.
Operating
frequencies were selected
near
50kHz)
to
reduce
transducer
backside
and
side-lobe
esponse
and to attenuate
long, multiple-reflection paths.
n mid-experiment, the
receive
transducer depth
was in
reased by one meter. Also, lower error rates were achieved
with
a -12dB power change (0.6
watts
instead of the design
level of 10
watts).
An 85 to 90% success
rate
was achieved.
I we
-
h
•
1
2.2 Surface Multipath.
A common surface-path
sit
uation is illustrated
in figure
1.
The
offending surface
scattered
backside arrival) signals clearly travel a longer
path
than the direct
path
signal and therefore take a longer
time to
arrive.
The
earliest-arriving
scattered
signals take
an
extra
delay
time
Td)
to
travel
an extra path
delay
P
d
)
as follows:
1)
d ..; h d
P
d
= B h
2
[ h - d) tanA - dtanBj2
os cos A
2a)
where d is
the
receive
transducer depth,
h
is the transmitter
depth,
A
is
the transmitting
slant
angle, B
is the scattered
signal
receive
angle
both
angles are measured from
the
vertical) and e is the speed of sound in seawater,
about
1.5m/ms. The surface
watch-circle radius
we)
is
related
to the slant angle
by
we = h -
d)
tan A.
If
the
watch circle
radius is known instead
of
the slant
angle
A, equation
(2a)
can be written:
P
d
= _ d _ Jh2 we
- dtan B)2
- J h -
d)2
wc
2
cosB
2b)
When
the
transmitter is
straight
below
A =
0 and
we
=
0), equations (2)
above
simplify to:
P
d
= _ d _
.
Ih2 dtanBp - h - d)
2c)
cos
BY
The
first delayed
surface-scatter multipath
arrival occurs
at Td
=
2d/e, when
the
arrival angle B
=
0 (surface angle
= 90°), followed by
more
sound arriving for B
>
O
The
first arrival delay
is about
27ms for a receive hydrophone
depth
of 20m.
For
slanted sound paths A >
0) equation (2a) shows
that
the surface-scattered
first-arrival delay time is slightly faster
than
for the direct overhead case; the shortest
path
occurs
for equal angles of incidence and
scatter
at
the
surface. As
an example,
for a transmitter in 3000m of water,
to
a
60m
deep hydrophone
at
a 2000m watch circle
distance, A = 34°
so the surface
incident
angle (given by
90°-A) is
about
55°.
Sound
scattered
at 55°
from
the surface
B
= 35°) will ar
rive with a 67ms delay compared
to
80ms for
the
straight
below case). Straight-line sound travel
has
been assumed
throughout,
even
though
for a
slanted
direct
path
sound
travel
is actually slightly
curved, due
to
refraction by
the
sound-speed
depth
profile;
This
does
not
affect
our
conclu
sions.
After the first
multipath
arrival, sound travelling longer
paths continues to arrive for a substantial period of
time;
this additional sound
constitutes most
of the
multipath
in
terference. Some of
the sound has
travelled very complex
pathways, involving
volume scatter
as well.
Although the
multipath
signals suffer surface-scattering losses( 6)
of
10
to
20dB, the beneficial effects of these losses are reduced by
,he
large
area of the
surface.
The
desirable losses are fur
ther
reduced during high sea-state conditions, when acous
tic surface
scattering
increases (e.g. see
the
backscattering
curves in ref 15 p. 264). However, for frequencies
above
20kHz (e.g.
33kHz), wind
velocities above 10
to
15
m/s
may actually cause reduced surface-scatter sound due
to
sound
attenuation
by small-bubble p o p u l t i o ~ in
the top
5 meters of
the
ocean( 7).
2.3 Fighting Multipath.
Several
methods
have
been
sug
gested
to reduce
signal
degradation
by
multipath
interfer
ence. One
is
to
use a
transducer with
high
back rejection
(or use a baffle).
In
the
3000m
example
above,
the
first
offending sound arrived at angle of 145.
0
from
the
trans
ducer forward direction (given by 180°
- A,
assuming the
transducer is pointed
down). A second
method
is
to
cre
ate a highly-directive receive transducer array(18). These
approaches increase
the
cost of
the
system, are painful
to
implement at low frequencies
and
have
limited
utility for a
variety of reasons. Furthermore, in shallow
water, directive
sensors may not be very helpful.
276
Acoustic transmission in shallow
water s
much more dif
ficult
than
in deep water, since it suffers from the exis
tance of many strong sound pathways to the destination,
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
3/8
numbers of surface
and bottom
reflec
Computer modelling(T,18) indicates
that
for 10kHz
in 200m deep water,
the Direct-to-Multipath
Ratio
(DMR)
may be
as
poor 88
6
dB at
ranges
of
than lOOOm. Actual measurements in
the
ocean may
poorer DMR. Higher-frequency transmissions will
sea-water absorption attenuation for
longer
multiple-bounce
pathways,
but
less
than 5dB of
is
calculated
at
50kHz,
due to this
effect.
sound transmission
methods
have been
to solve
the multipath
problem. Systems
many frequency channels(l1,12,IS) have been
proposed,
up to
32 frequencies(H), so
that the system can
switch
a new frequency before the multipath interference ar
In a common approach, the frequencies in use
are
every 50 to lOOms, allowing
the
multipath
energy
decay on the old channel. Since
the
decay
time
allowed
can be reused is proportional to the num
of available frequency channels, this may well be a true
more is
better .
Of course
the telemetry system
more
complex,
but
the improved results
that
be
obtained
in all environments are very
attractive.
f O ~
lot-to ~ I I I
1-
2
A
New Chirp Telemetry
Method
new Sea
Data chirp
acoustic
telemetry system
is
based
a variation
of
the
frequency-diversity idea: use
an
in
number of frequencies. This is achieved by sweep
the telemetry carrier
frequency while applying fsk data
(fig 2).
The
transmitted
signal
P t)
is a single
starting at II, and
changing
at
a
smooth rate
/dt, plus fsk frequency shifts with amplitude 12:
P t)
=
cos[w t)t]
3)
w t)
=
211 11
t t
12
M
t»)
4)
dl
=
s
-
dt
Ts
5)
where Ts is
the duration
of
the
sweep and 11
and
Is
are
the starting and
ending sweep frequency
and M
t)
=
0
or
I according
to the
data
bits. The
modulation ampli
tude, 12
is chosen large enough, e.g.
>
150Hz,
to
eliminate
doppler-shift spreading problems, which will be less than
40Hz (O.33Hz/kt per kHz). .
If the
receive frequency
is
accurately swept
to match the
transmitter,
a small receive
bandwidth (constrained by the
data
rate and
the
fsk 0,1 frequency shift) can be used,
just
as in a conventional fsk system. A small bandwidth will im
prove
the
signal-to-noise
ratio
(SNR)
not
only by rejecting
ambient noise
but
also by rejecting
the
(delayed)
multipath
energy from
the
old-channel frequencies.
n
a chirp
telemetry
system, the effective frequency-diversity
channel usage time Tu)
can
be equal to
the
time required
for
the carrier
frequency sweep to change by
more than the
receive
bandwidth
BW),
as follows:
Ts
Tu BW
3 - I
6)
The
usage time
can
be easily
set at
under 50ms (e.g. BW
= ~ O O H z
sweep 4000Hz in 650ms), allowing excellent re
jection of multipath signals.
As
an
added benefit,
the
new
chirp telemetry approach
can
be inexpensive, compared to other frequency-diversity
methods,
since multiple frequencies are not required (i.e.
no synthesizer)
and
receive decoding
can be
simplified (i.e
o multiple
filters
or FFT
analyzer). To
understand our
approach and
the
role
of
all
the
elements in
the
sweep wave
form
of
figure 2, we'll
start
by considering how
the
receiver
works (see figure 3).
277
3.1
Signal
Description.
Because a
telemetry
receiver
contains many circuit elements
that
consume electrical power,
it's desirable to switch
the
power to a
portion of these
cir
cuits.
n the
receiver design above, a
number
of
components
are continuously
powered
in order
to
detect the
arrival
of
an alert
signal.
These
are
the
preamp, AI,
a
bandpass
fil
ter, BP, the 10 detector and a power-control
circuit
(for
our
experiment,
10
=
9.0kHz). The
bandpass
filter is designed
to pass signals over
the
entire 10 to
f3 range
of
the
system
(a
double-
or
triple-tuned filter)
and to
reject
intense
low
frequency noise from shipping, etc
(extra LF
cutoffs).
The
fa energy detector operates on a principle similar to that
used by
many
high-resolution acoustic positioning
systems
(5):
an amplitude
limiter (to
establish constant power),
a
sharp fo
filter
and
a
comparator with
a
time constant,
work
together
to determine if the fo energy present is above
the
background noise adjacent to fo by a
threshold
amount.
When
the
10 alert-tone energy
is detected, the remainder
of the
receiver, including
the
microprocessor,
is turned
on.
After a short time,
to, the
transmitter shifts its frequency to
(for
our test
11
=
10 600Hz),
creating
the data-trigger
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
4/8
tone. This trigger tone start pulse is detected by the
11 energy detector (similar to
the
1 energy detector)
and
is
used to start
a sweep
generator
and voltage-controlled
oscillator (VeO). The resulting frequency ramp is designed
to
precisely
track the transmitter's
sweep
with
a fixed offset,
hr where frr
>
13 - 10)/2 to avoid images.
This
ramp
frequency
is
the local oscillator (LO) input to a mixer, and
has a frequency, lLO , similar to equations (3) to (5)
except
as follows:
dl
fLO = f t) =
I i dt
t IIr
7)
The resulting
intermediate
frequency (IF) output from the
mixer after IF-stage
filtering is:
v t)
=
sin[27rI t)t] n t)
8)
f t) = fIr 12M t)
9)
where n t) is the received noise, with a noise bandwidth
given by the IF-stage bandpass. This signal is limited and
applied to a frequency
discriminator
to
track 12 and
deter
mine whether M
=
0 or
1.
A
data
precursor time delay, t
in fig 2, allows the circuits
to
settle before data discrimina
tion must start. Also a warmup time, tw = to less the
1
detect
time,
is available
forthe
crystal
in
the
receiver's
microprocessor
to
start, etc.
The
frequency discriminator in fig 3
is
a phase-locked loop
(PLL) circuit, which forms a tracking filter
to further nar
row the noise
bandwidth
of
the
receiver.
The
input
stage
of the PLL
is
a
limiter that
responds to the
strongest
sig
nal within
the
IF bandpass and acts
to
reject any weaker
signals,
thereby
further rejecting (quieting)
unwanted
mul
tipath signals . A full-wave mixer phase-detector circuit
(exclusive-OR) and the PLL loop filter act in a
+5
BP
\
FO
DETECTOR
POWER
CONTROL
SWEEP
GEN
+5
=
SWITCHEC
POWER
3
manner
to
maintain the v o
output
frequency
near
wet)
in formula (9). Further filtering of the
input
to
the v o
-
a varying dc voltage
vet) ex wet)
- along
with
ac-coupling
and clamping, yields the original data-stream signal, M t).
The
PLL loop filter and low-pass filtering of M t)
set
the
noise bandwidth,
BW,
of the
telemetry
receiver. The re
ceiver
should be
able
to operate
with very low SNRs, al
though
the
data error
rate
may not then be zew.
4.
Transmitter
The transmitter (figure 4) helps illustrate the
simplicity
of the chirp telemetry scheme. A few low-data-rate con
trollable
outputs
from the instrument's microprocessor
are
sufficient
to
operate the transmitter. These outputs include
the sweep
generator power
control,
an
enable for
the output
driver as soon as
the
v o
is stable,
a
start
pulse (SP) shift
ing
the
frequency for a data trigger, a sweep
enable
(SE)
and
the
data bit (DB)
modulation
signal.
Another
line
sets
the sweep rate (SR) to allow
optimizating the
system for
deepsea or
shallow-water
use.
BATTERY
LI
SYSTEM
,p
SP SWEEP VB0
I . : . . . . = ~ = ~ = M = O = D G UE L = A ~ = O = R = - _ - = : ? r n c ~ _ - , rm
R
ENABLE TI
4
278
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
5/8
At low frequencies, e.g. 10kHz, obtaining a transmit op
erating range of
5kHz
is
a challenge, due
to the
narrow
band nature
of a
tuned
acoustic
transducer. In
figure 4,
the
reactive component of
transducer
Xl is removed us
ing tuning coil L1, with a series resistor R1 to increase the
frequency range.
In the
9
to
14kHz experiment
to
be de
scribed, a modified ITC
type
3013
transducer
(which nor
mally
has
transmit-voltage-response
peaks at
9 and 14kHz)
was used with a 22mHy choke
and
a 50 ohm
damping
resis
tor.
A very
acceptable
calculated
network
output
flatness
(+ 140±2dB/V) was obtained over a 7.5 to 14kHz range,
and verified
with
pulsed measurements in
the
local
YMCA
swimming pool. If necessary, a more complex network could
be devised. When operating
the
system with 5kHz sweeps
at 33kHz, using a custom-designed
transducer,
a
damping
resistor
is
less important.
4.1 Power.
In
the
deep ocean
test, the output
stage con
sisted of a pair of VMOS transis tors driving a
center-tapped
transformer
with
a regulated 12V input.
This
provided
about 20 watts of power into the transducer network and
yielded a
modest
calculated source level of + 179dB re l/LPa
at
1m, confirmed
in
the
pool test.
The current drain
from
the instrument battery was less than 2A during transmis
sion, a very acceptable level for any instrum.nt with several
,tacks of alkaline
batteries.
Although lower power levels may be used in practise, our
thought
was
to
get good quality eata on
the experiment
OAT
tape and
subsequently degrade it
with
noise when we
tested transmit codes and receiver designs in the lab. How
~ v r the
higher-power energy usage
is not unattractive:
At
300 baud, less than 0.1 Joules per bit is required, including
alert tone,
etc. Since a single stack of alkaline D-cells con
tains about 0.5MJ of
energy,
it could
power
about
100,000
transmissions of 50-bit data blocks.
SR
IK
R4
1
R5
o R6 R7 2V
2 K 165K 1
R3
R8
4.99K
rl
1
RO
RI
R2
4.2
Design Simplicity. Because I always miss
the
ab
sence of electronic-circuit schematics at IEEE conferences,
I ll be sure to include one here. Figure 5 shows details of
the transmitter
sweep
generator and
serves
to further
illus
trate
the
simplicity of our new approach, while giving me
a chance to dispel any concerns over drifts, tuning, etc.
The most
important
component is the voltage-controlled
oscillator (VCO) chip U4,
an
Analog Devices AD537, which
operates
at
twice
the transmitter output
frequency.
This
VCO chip creates a very stable frequency and has low
power-supply and temperature drift coefficients (0.01 /volt
and 0.03 /10
degrees C).
When
used with
stable
compo
nents (capacitor
Cl and
resistor RIO are low-tc compo
nents), the
AD537 may allow a circuit with lifetime factory
calibration. The
VCO follows
the
formula I = Vs/[1O(R9
+
RlO)Cl].
Here R9 sets
the
exact coefficient for
the
VCO
frequency-programming voltage, V
3
which comes from am
plifier A3 (LMIO, chosen for ImA sink capability when
V
ou
=
0.2V
at
the end
of
the
sweep).
This
amplifier s
summing junction allows the telemetry system
operating
parameters to
be exactly ratiometrically determined by pre
cision resistors RO, Rl
and
R2 according
to the
following
formula:
(10)
where
Fo
=
I/RO
sets
the 1
alert frequency,
1
= I /Rl
sets
the
11 - 1 data trigger frequency shift,
2
= I/R2
sets
the h fsk
modulation
level and
s
=
1/R3
sets the
sweep
rate
(and hence Is). Amplifier
Al
(OP-20, chosen
for low offset voltage) creates a reference I-volt above
the
amplifier-reference signal (also 1 volt), so
that
k
= R8.
In
the
experiment, an electronic switch selected two values of
R2
to
allow two fsk
modulation
levels.
Amplifier A2 (OP-90, chosen for low input
current
and off
set voltage) is a ramp, which operates (when switch SE
opens) with
an integration
constant =(R4+R5)C2.
The
integrator uses voltage source trim R6
to
allow two cali
brated sweep rates according to the resistor ratio
R4/R5
CI
2f
5
1 E t
VR
279
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
6/8
IIwitch
SR. The
sweep
generator operates on
5.0 volts
supply,
a.nd
the entire circuit requires only two
sim
calibration points, yet
we're able
to
get our infinite
channels.
. Signal Propagat ion
Loss,
Noise, S N R
expected
signal-to-ambient-noise
ratio
(SNR)
can be
.
alculated
(in
dB) by subtracting the speading
and
attenu-
losses and the background noise level from the- trans
source level:
SNR =
SL -
20Iog(r)
-
r
_
NSL
- lOlog BW) 11)
1000
SL
is the
source level (dB
re
1tLPa
at
1
yard),
cor
for
the transducer
directivity
index, r
is the range
-
not
km), a is
the
seawater attenuation coefficient
than 1dB/km
for frequencies below 15kHz), NSL is
the
spectral
level
(dB re
1tLPa/v1fz ),
and BW
the receiver
bandwidth
(Hz).
The
equation assumes a
preamp
and
does not include the
improve
a
directive
receive
transducer
will
provide
in
rejecting
ambient
noise, which could exceed
seawater
attenuation
is due to
magnesium-sulfate ionic
with
an
absorption
coefficient of
about 0.7 and
dB/kyd
(at
10°C
and zero depth)
for 12
arrd
35Hz, re
(see ref. 15,
page
109
and
ref. 22).
Over
the
of 8
to
50kHz,
the absortion
coefficient increases
by
square of the
frequency, decreases
about
7% for
each
of depth
and
increases
about
2% for each °C of tem
decrease. The latter two effects
tend to
cancel
other
out in
the top half
of
the
deep ocean. Applying
formulas
to
expected ocean conditions yields
the
values
which
can be integrated
over
the
sound propaga
to determine the absoption
loss for various
applications.
Depth (m)
Temp OC)
Attenuation (dB/kyd)
@10kHz
@33kHz
0 20
0.65
4.S
3000
4
O.SO
3,3
6000
4
0.38
2.4
using
the system in
deep
water, with
a Skm
path,
we
calculate
a
16dB expected SNR
for (poor)
20m/s
wind
as follows: Given
the TVR of
the
transducer
t
+141
dB
per
volt, and considering
a
l.SdB
loss for
the
tuning
resistor,
we can calculate an output
acoustic
of +178dB, for 20 watts
(this
was confirmed in
pool
test).
We lose -74dB from Skm spreading and -
from attenuation
(at
10kHz). The resulting calculated
280
signal strength
of
+99dB
re 1J LPa near
the surface,
is about
16dB
louder than the wind
noise for
20m/s
(NSL
= +S8dB
at 10kHz), assuming a 300Hz receiver
bandwidth (+25dB)
and
an isotropic receive
transducer
(Dr
=
OdB).
5.1
Shallow
water .
Using 33kHz in shallow
water
at
lOoC, 3km of range will result in
about -22dB
of absorption
loss,
assuming the actual (scattered and
reflected) path
is
30% longer
than
the
range. Since
the sound is
in a channel
the spreading
loss
may be
less
than the
-69dB value
r o ~
formula
(11),
say
+lOdB
for
30m water depth
(2).
The
fi-
nal system SNR is similar to the case above since NSL is a
bit lower at 33kHz. Because 3km of range in shallow water
will be subjected
to
severe
multipath
interference,
the
sweep
rate
may increased and
the
data
rate
may
be
decreased to
combat this. Also,
the
telemetry system's processor
can
easily allow using slower
data
rates, with 25ms dead
peri
ods in between
each
bit,
to
allow
the
immediate
multipath
energy to
decay.
6.
Sea Trial using
a
DAT Recorder
The
experiment
was performed on 16
to 17
June
1988
dur
ing cruise
OC200
of
the
WHOI
vessel R.V.
Oceanus, t
a
site approximately
400 miles
east of Cape Hatteras, just
north
of the Gulf Stream, in 3775m of water.
The
undersea
transmitter
for
the
experiment
operated
over a range of 9
to
14.5kHz and was installed in a Sea
Data
model 1665 In
verted Echo Sounder (rES), deployed on
the bottom. The
receive hydrophone was
the standard
EG G
acoustic
re
lease deck-set sensor (an
ITC
3013
transducer),
suspended
over
the
side of
the
ship
about 18m
below
the
surface. A
custom-built preamp
with a 5kHz 2nd-order
bandpass
filter
was used with
the
hydrophone.
To
test the
new telemetry system, we elected
to
transmit
test
signals from
the
ocean
bottom to
various lab-based re
ceiver circuits
via
a
shipboard
precision audio recorder. In
this way we
could
perform receiver tests in the lab with var
ious noise levels and different types of interference, using a
TEST
;
~
TR NSMIT
;; 7 } > I p 1
8/17/2019 Deep-Ocean Tests of an Acoustic Modem Insensitive to Multipath Distortion
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large variety
of transmission types
as
they
were
actually
received in
the
ocean. We used a small
portable
16-bit dig
ital audio
tape
(DAT) recorder (Technics model
SV-MDl,
complete
wit.h a
manual entirely
in
Japanese). Thus
we
were able
to obtain
perfect (90dB
dynamic range,
flat
to
18kHz, 0.01% time
stability)
digital analog recordings
of
the received hydrophone signals.
During the experiment the transmitter
variables were cy
cled
through
a
variety
of combinations
using a
parameter
table
in
the
microprocessor's
program.
The parameters
in
cluded: data
rates
(100
to
360
baud),
modulation
index
(300 and 500Hz), chirp sweep rate (10
and
40kHz/s), chan
nel decay quiet times (62ms to 14s), transmit
duration
(1
to
10
bytes/record and
3
to
80 records)
and the transmitted
data patterns.
During
the experiment the
receive variables included
slant
range
and
Dolphin
activity
level.
Winds of
lOkts and a
steady
rainfall
both occurred at
various
times
during the
experiment. The experiment
was
performed with
the
low
frequency version (to = 9kHz), since all
the
available com
ponents
(IES
transmit
stage, receive hydrophone
and
DAT
recorder) weighed
against
the
33kHz version.
Initial oscilloscope examination of
the
DAT tapes shows 3
to
lOdB of
fading after the
sweep
was under
way
(due to
multipath?), -6
to -lOdB of
delayed (obvious) multi
path
interference
and
a + 10 to +20dB
S i ~
(4kHz noise band
width), depending upon surface conditions.
The concept of
using a DAT recording
to
provide receiver
test signals has proven
to be
very useful.
At
this
writing,
excellent
performance has
been
obtained playing back the
tapes
into
our prototype
receiver. In
this
fashion we will
easily be able
to
optimize
the
performance of
the
receiver
design
with tests using bench instruments, e.g. the SNR
can
be
degraded
with
noise
generators.
Already, we were able
to
painlessly
test the
improvement that a CMOS-switch
ana
log mixer provided over a
limiter/XOR-gate
mixer. Further
DAT
recorder ocean experiments are planned
in
shallow
water.
6.1 Dolphins. We experienced considerable interference
from dolphins, who were curious
about the ship and
enjoyed
playing with
the hydrophone.
A few dolphins used
their
variable-rate
pulse
sonar to locate and
"ping"
the
trans
ducer;
at
closest
approach they
increased
the
ping
rate to
buzz. Like
our hydrophone,
the
dolphins
could
hear
the
transmitter on
the ocean bottom.
Amazingly, they did a
good
job of
mimicking
the
9 to 14kHz sweep signal
of the
telemetry But we
haven't yet
decoded
their
transmissions
(Does anyone know, do they use ASCII code? And if so,
is it
Is
b first?).
t
was necessary
to
move
the ship
several
times,
and to turn
off
the
fantail lights. This may
be an
argument in favor of
higher
frequencies, such as
our
33kHz
version.
281
1
System Considerations
The
receiver and
transmitter of
the
acoustic
modem
each
occupy one
card,
as does
the
processor.
The
33kHz trans
ducer is very small, l.6-in (4cm) in diameter,
and
is con
structed with an O-ring
groove
and 3/4-16 stud with
em
bedded
wires,
to
allow it
to be
screwed directly
into an
endcap. Thus, the system can
easily
be added to many
existing designs. A standalone version mounted in a small
housing with
a
battery
is
planned
as well.
The
final
telemetry
system software will employ a data
transmission
protocol suited for systems applications, and
error checking features. A unique code can be sent from
each
transmitter
for identification. Controlled
redundancy
can greatly
reduce
the
error rate: block error-correction
codes
such
as
the
Reed-Solomon code(21) can allow for cor
rection (after reception) of up
to
15 errors within a ISS-bit
block while achieving
an
80% code
rate
(125 data
bits).
Although both
receiver
and transmitter cards
will often
be
located at both ends of
a
system,
creating a full
underwater
MODEM, telemetry
systems
can be substantially
simplified
if one-way data transmission is used. f stable timebase os
cillators are
employed(19), offset time-slot channels
may be
established
so that many
undersea instruments can trans
mit
to
a
central
receiver(20) on a single frequency, without
requiring
a
command
for
the
transmission. Furthermore,
studies
have shown(7) that a one way acoustic data
trans
mission
system
can
be optimal,
e.g. "Analysis of
the
ADTL
data indicated that command and retry
provided only
min
imal improvement in
the amount of
data passed
without
errors.
8. Conclusion
t is
our expectation
that considerable
improvement
over
other
traditional
methods
will
be
experienced with
our
new
swept-frequency telemetry, at a reduced cost.
t
is our hope
that our
work will help lead
to
a
greater and
happier
use
of
acoustic telemetry in
the
ocean.
9. Acknowlegments
One
of
us (GC) wrote some of the transmitter
software
and
singlehandedly
( )
performed
the
undersea experiment,
during the
wee hours when
the rest
of
the ship
was asleep,
while another (DN) modified
the
IES undersea transmitter
and constructed preamps and
prototype
receivers
to ana
lyze
the
DAT
tapes.
Special
thanks are due
to Kevin Boyce
for creating a
major portion
of
the
original IES micropro
cessor code,
to Dan
Frye
and others at WHO
I for
their
suggestions
and
review of
telemetry system
goals
and to
Prof. Randy Watts at URI
for his
encouragement.
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References
Parker,
B.B. (1985) Real-Time Oceanographic Model Systems: Present
and
Oceans
'85,
Proc.
IEEE-MTS
Conf.,
pp W.-214.
Urich,
ILJ.
(1982) Sound
Propagation
in
the
Sea, Peninsula Publishing, Los
chapter 10-12.
Coffey,
D.M.
and
PaquetteiI985): Aaura