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APPROVED: Hualiang Zhang, Major Professor Hyoung Soo Kim, Committee Member Shengli Fu, Committee Member and
Interim Chair of the Department of Electrical Engineering
Costas Tsatsoulis, Dean of the College of Engineering
Mark Wardell, Dean of the Toulouse Graduate School
DESIGN AND APPLICATION OF PHASED ARRAY SYSTEM
Han Ren
Thesis Prepared for the Degree of
MASTER OF SCIENCE
UNIVERSITY OF NORTH TEXAS
August 2013
Ren, Han. Design and Application of Phased Array System. Master of Science
(Electrical Engineering), August 2013, 57 pp., 4 tables, 50 figures, references, 48 titles.
Since its invention, phased array has been extensively applied in both military
and civil areas. The applications include target detecting and tracking, space probe
communication, broadcasting, human-machine interfaces, and remote sensing.
Although the phased array applications show a broad range of potential market, there
are some limitations of phased array’s development: high cost, complex structure,
narrow bandwidth, and high power consumption. Therefore, novel ideas are needed to
reduce these constraints. In this thesis, several new approaches about the design and
application of phased array are presents. First, the principle of phased array and
fundamental design equations are introduced. Second, a new application of phased
array antenna for radar respiration measurement is presented. By integrating a 4×4
Butler matrix with four-element antenna array, there will be four distinct main beams in
radiation pattern. This new approach can improve the measurement accuracy and
realize a high detecting rate. Third, a compact phased array antenna system based on
dual-band operations is introduced. Dual-band function can make N-antenna system
obtain 2N unique radiation beams (N is an integer) and achieve a significant size
reduction compared to the conventional single-band system. To verify the design
concept, a four-element phased array antenna working at 5GHz and 8GHz is designed
and fabricated. The measurement results make a good agreement with the simulations.
Finally, a novel architecture of steering phase feeding network by using bi-directional
series-fed topology is presented. This bi-directional series-fed network needs less
phase shifters and realizes steering phase function by applying control voltage.
Copyright 2013
by
Han Ren
ii
ACKNOWLEDGEMENTS
I would like to express the deepest appreciation to my major professor Dr.
Hualiang Zhang, who has the attitude and the substance of a genius: he continually and
convincingly conveyed a spirit of adventure in regard to knowledge and research, and
an excitement in regard to teaching. His expertise in RF and microwave region
improved my research skills and prepared me for future challenges. Without Dr. Zhang’s
guidance and persistent help, this thesis would not have been possible.
I would like to thank my thesis committee members, Professor Hyoung Soo Kim
and Professor Shengli Fu for their helpful suggestions and comments during my study.
In addition, I am grateful to Professor Changzhi Li of Texas Tech University for
providing his technology and support to my research.
I want to thank all of my lab mates for their help, support, and cooperation.
Finally, I want to express thanks to my family, especially my wife for her constant source
of inspiration and support.
iii
TABLE OF CONTENTS
Page
ACKNOWLEDGEMENTS……………………………………………………………………...iii
LIST OF FIGURES………………………………………………………………………..……vi
LIST OF TABLES……………………………………………………………………………….ix
Chapters
1. INTRODUCTION…………………………………………………………………………….1
1.1. Background………………………………………….…………………..1
1.2. Motivation…………………………………………….…………………..2
1.3. Overview of the Thesis……………………………….………………...3
2. PRINCIPLE OF PHASED ARRAY………………………………………………………...6
2.1. Introduction………………………………………………….…………...6
2.2. Structure of Phased Array………………………………….…………..6
2.3. Classification of Phased Array…………………………….…………..9
2.3.1. Parallel-fed Network…………………………………...11
2.3.2. Series-fed Network…………………………………….11
3. NEW APPLICATION OF PHASED ARRAY ANTENNA FOR RADAR RESPIRATION
MEASUREMENT…………………………………………………………………..……...13
3.1. Background and Motivation………………………….……………….13
3.2. Design and Analysis……………………………………….………….15
3.3. Fabrication and Measurement Results…………………….………..21
3.4. Conclusion…………………………………………………….………..24
iv
4. A NOVEL COMPACT PHASED ARRAY ANTENNA SYSTEM BASED ON DUAL-
BAND OPERATION……………………………………………………………………….25
4.1. Introduction……………………………………………………….…….25
4.2. Design and Analysis………………………………….……………….27
4.2.1. Dual-band 90° Coupler…………………….………….31
4.2.2. Dual-band 45° Phase Shifter…………………………32
4.2.3. Dual-band Antenna……………………………………33
4.3. Simulation and Measurement Results……………….……………...35
4.4. Conclusion……………………………………………….……………..39
5. A NEW ARCHITECTURE OF STEERING PHASE FEEDING NETWORK BY USING
BI-DIRECTIONAL SERIES-FED TOPOLOGY…………………………………………40
5.1. Introduction…………………………………………………….……….40
5.2. Bi-directional Structure………………………………….…………….41
5.3. New Phase Shifter Design……………………………….…………...45
5.4. Simulation Results………………………………………….………….48
5.5. Conclusion……………………………….……………………………..50
6. CONCLUSION AND FUTURE WORK…………………….........................................51
REFERENCES………………………………………………………………………………52
v
LIST OF FIGURES
Page
2.1 General configuration of N-element phased array………………………………………7
2.2 (a) The pattern of dipole antenna element……………………………………………….8
2.2 (b) The pattern of array factor……………………………………………………………..8
2.2 (c) The pattern of a dipole antenna array after pattern multiplication…………………8
2.3 General diagram of the space feeding………………………………………………….10
2.4 General diagram of the phased array applying parallel-fed network………………...11
2.5 General diagram of the phased array applying series-fed network………………….12
3.1 General schematic of phased array antenna integrated with radar system………...14
3.2 Configuration of 4×4 Butler matrix network…………………………………………….15
3.3 The topology of the proposed Butler matrix network in HyperLynx………………….17
3.4 (a) Simulation results of insertion loss of Butler matrix network with P1 as the input
port……………………………….....……...………………………………………….…...18
3.4 (b) Simulation results of return loss and isolation of Butler matrix with P1 as the
input port…………………………………….................................................…….……19
3.4 (c) Simulation results of phase response of Butler matrix with excited P1………….19
3.5 (a) Simulation results of insertion loss of Butler matrix network with P2 as the input
port……………………………….....……...………………………………………….…...20
3.5 (b) Simulation results of return loss and isolation of Butler matrix with P2 as the
input port…………………………………….................................................…….……20
3.5 (c) Simulation results of phase response of Butler matrix with excited P2………….21
3.6 Topology of proposed phased array antenna system…………………………………22
vi
3.7 Photograph of proposed phased array antenna system with SP4T switch…………23
3.8 (a) Measurement result of main beam angle with excited input port P1…...............23
3.8 (b) Measurement result of main beam angle with excited input port P2…...............23
3.8 (c) Measurement result of main beam angle with excited input port P3…...............23
3.8 (d) Measurement result of main beam angle with excited input port P4…...............23
4.1 Configuration of proposed compact phased array antenna system based on dual-
band operation…………………………………………………………………………….26
4.2 Configuration of the modified 4×4 Butler matrix………………………………………..28
4.3 Calculated radiation beam angle θ versus ratio parameter N………………………..29
4.4 Schematic of the dual-band 90° coupler………………………………………………..31
4.5 Dual-band 45° phase shifter with T-shape structure…………………………….…….32
4.6 Schematic of the dual-band circular antenna…………………………………………..34
4.7 (a) Photographs of the fabricated phased array system from top view…………......34
4.7 (b) Photographs of the fabricated phased array system from bottom view…………34
4.7 (c) Structure of the fabricated phased array system from cross-sectional view……34
4.8 (a) Simulated results of magnitude response of dual-band 4×4 Butler matrix with P1
as the input port……………………………………………………………………………35
4.8 (b) Simulated results of magnitude response of dual-band 4×4 Butler matrix with P2
as the input port……………………………………………………………………………35
4.9 (a) Simulated phase response of dual-band 4×4 Butler matrix with input from P1 at
5GHz………………………………………………………………………………………..36
4.9 (b) Simulated phase response of dual-band 4×4 Butler matrix with input from P1 at
8GHz………………………………………………………………………………………..36
vii
4.9 (c) Simulated phase response of dual-band 4×4 Butler matrix with input from P2 at
5GHz………………………………………………………………………………………..36
4.9 (d) Simulated phase response of dual-band 4×4 Butler matrix with input from P1 at
8GHz………………………………………………………………………………………..36
4.10 (a) Simulated and measured return losses of proposed phased array antenna
system at 5GHz…………………………………………………………………………..37
4.10 (b) Simulated and measured return losses of proposed phased array antenna
system at 8GHz…………………………………………………………………………..37
4.11 Simulated and measured radiation patterns at two frequencies……………………38
5.1 N-element phased array with a parallel feed network…………………………………42
5.2 N-element phased array with a series feed network…………………..………………42
5.3 Configuration of proposed feeding network integrated with antenna array…………43
5.4 Beam directivity versus coupling factor for an 8-element antenna array……………44
5.5 Input power dissipation versus coupling factor for an 8-element antenna array...…45
5.6 The circuit diagram of proposed phase shifter…………………………………………46
5.7 The curve of required inductance versus varactor capacitance……………………...47
5.8 The insertion loss of phase shifter versus the phase shift……………………………48
5.9 Topology of proposed bi-directional feeding network…………………………………49
5.10 Simulated phase difference of proposed feeding network versus the varactor
capacitance as the signal is input from the left end……………………………………50
viii
LIST OF TABLES
Page
3.1 Phase response of the output signal of Butler matrix network……………………….16
3.2 Radiation beam angles of the proposed system by exciting input ports…………….22
4.1 Phase difference (𝜟Ф) of the Butler matrix and beam angle…………………………30
4.2 Antenna gain measurements at two frequencies………………………………………39
ix
CHAPTER 1
INTRODUCTION
1.1 Background
Since Nobel Laureate Luis Alvarez applied the phased array (or phased array
antenna) in a rapidly-steerable radar system for “ground-controlled approach” during
World War II, phased arrays have been around for more than half a century. In the early
period, the phased array was the most advanced technology and traditionally used in
military (e.g. national missile defense and space probe communication). Especially in
radar system, the phased array is an important component to realize the target
detecting and tracking. As the commercial demand increases rapidly and the advanced
science becomes more open, the phased array has started to be applied in civil areas.
The great growth in civilian radar-based sensors and communication systems has
shown increasing interest in utilizing phased array technology for business purpose. In
recent years, the phased array has achieved a great development in many new
applications: broadcasting, human-machine interfaces, search and rescue, remote
sensing, imaging, etc. All of above sufficiently proves the potential benefit of the phased
array in human society and research.
Compared to the conventional mechanical scanning system, phased array
system has some remarkable advantages. First, the phased array can generate higher
scanning rate (or speed) due to the electrical tuning mechanism. Second, it is very easy
to change the direction of radiation beam by adjusting the frequency and phase. Third,
the accuracy of phased array is much better than mechanical scanning. Finally, the
1
phased array can provide stable performance without any mechanical failure. As a
result, it is envisioned that the phased array will completely replace the mechanical
scanning array in both military and civil areas sooner or later.
1.2 Motivation
Although the phased array applications show a broad range of potential market,
the technology of phased array has not been widely deployed in the commercial area.
There are several important drawbacks, which limit the development of the phased
array. The primary disadvantages of utilizing phased array are high cost, complex
structure, narrow bandwidth, and high power consumption. For aerospace applications,
these disadvantages can result in negative effects on the quantity of the prototypes. In
military areas, these constraints often restrict the accuracy of the radar system.
Moreover, the high cost and complex structure of phased array are two main
impediments to achieve their deployment in a large-scale commercial application.
Therefore, it is meaningful to find some novel ideas and approaches to reduce these
constraints.
As discussed in previous paragraphs, there are some disadvantages to limit the
phased array deployment in a wide range of applications. It is obvious that addressing
these technical impediments and reducing system’s cost and complexity can lead to
more comprehensive applications in military and civil areas. Thus, any concept and
design achieving significant reductions on the cost and complexity of phased array
system will be very attractive for practical purposes.
2
Despite these constraints, the phased array is still very powerful and applied as
an advanced technology. For instance, the primary advantage of the phased array in
communication systems is spatial filtering. The spatial filtering of the phased array can
not only inhibit the signals generated from undesired directions but also reduce the
fading and multi-channel interference. Although all signals are generated from a single
transmitter, the phases of these signals arriving at the receive antenna must be different.
Therefore, by applying the phased array, it is possible to receive the desired signal from
a specific path while eliminating the others on other directions. Moreover, applying the
phased array in a transmitter antenna can increase the transmit power level in a desired
direction without the need of increasing the total transmitted power. On the receiver side,
phased array can achieve a significant improvement on the sensitivity between the
signal and noise. Therefore, the phased array will have a great prospect and
development in the future.
1.3 Overview of Thesis
This thesis presents several novel approaches about the design and application
of phased array systems. The second chapter introduces the principle of phased array
and presents the fundamental design equations. In addition, the classification of the
phased array architecture is also discussed.
In the third chapter, a new application of phased array antenna for radar
respiration measurement is presented. Conventional respiration measurement are often
undesirable because they are either invasive to patients or do not have high accuracy.
This thesis provides a novel non-contact respiration measurement method by applying
3
phased array antenna. Four identical patch antennas are employed to construct an
antenna array. To provide optimal feeding to the designed patch antenna array, a 4×4
Butler matrix using microstrip line is applied. By integrating this 4×4 Butler matrix with
the four-element antenna array, there will be four distinct main beams in the radiation
pattern. This new approach can improve the measurement accuracy and realize a high
detecting rate. Finally, a phased array antenna system working at 5.8GHz is designed
and measured. Both simulation and measurement results can agree well with the
concept.
In the fourth chapter, a compact phased array antenna system based on dual-
band operations is presented. Unlike the conventional single-band phased array
antenna which increases the number of antenna element to get more radiation beam, in
the proposed design, a new compact phased array antenna can maintain the same
function as the conventional one and realize a great size reduction. Specifically by
applying dual-band function, it is possible to use an N-antenna system to obtain 2N
distinct radiation beams (where N is an integer). In addition, this proposed phased array
antenna can achieve a significant size reduction compared to the conventional single-
band system. As a proof-of-principle, a four-element phased array antenna operating at
5GHz and 8GHz is designed and fabricated. The measurement results are in good
agreement with the simulations, validating the proposed design idea.
Chapter 5 introduces a novel architecture of steering phase feeding network by
using bi-directional series-fed topology. This bi-directional series-fed feeding network
needs less phase shifters compared to any of the conventional architecture of phased
array. It is expected that this novel architecture can result in a significant reduction on
4
the cost and complexity of overall phased array. In the proposed phased array antenna
system, a novel compact phase shifter is also designed and utilized. Finally, the overall
system is able to steer the radiation beam by applying a single control voltage. To verify
the design concept, a 5.8GHz steering phased antenna array is designed. The
simulation results show good agreement with the design theory.
Finally, the last chapter concludes the thesis and outlines its achievement.
Furthermore, direction for the future work is also presented.
5
CHAPTER 2
PRINCIPLE OF PHASED ARRAY
2.1 Introduction
Phased arrays have been around for more than half a century. In antenna theory,
a phased array is an array of antennas in which the relative phases of the respective
signals feeding the antennas are varied in such a way that the overall effective radiation
pattern of the phased array is enhanced in a desired direction and suppressed in
undesired directions. A classical antenna array is a group of multiple identical antennas
fed by a common source to generate a radiation pattern with a special direction. In
principle, by shifting the phase of the signal emitted from each antenna element, the
direction of main beam can be steered. This forms the fundamental operating
mechanism of the phased array.
2.2 Structure of Phased Array
The general configuration of an N-element phased array is shown in Fig. 2.1. The
antenna array consists of N identical antenna elements with an equal spacing d along
an axis. In the transmitting mode, it is assumed that all antenna elements have identical
amplitudes but each succeeding antenna element has a progressive phase of β to lead
current excitation relative to the preceding one (β represents the phase by which the
current in each element leads the current of the preceding element). The function of
feeding network is to generate the identical magnitude response and the β progressive
6
Fig. 2.1 General configuration of N-element phased array.
phase response to the antenna array. After the signal goes through the feeding network
and is delivered by the antenna array, a radiation pattern with a special angle θ will be
achieved. At the θ direction, the signal intensity is always the strongest among all
regions.
Based on the structure of phased array, the total field of the array is equal to the
field of a single element positioned at the origin multiplied by an important factor which
is widely referred to as the array factor (AF):
E(total) = [E(single element at reference point)] × [array factor] (2.1)
d
Feeding network
Signal Input
β 0(N-1)β (N-2)β (N-3)β (N-4)β Variable Phase Delay
Antenna Array
θ
Radiation Pattern
7
(a) Antenna element (b) Array factor
(c) Total radiation pattern
Fig. 2.2 The pattern of (a) antenna element only and (b) array factor only (c) a dipole
antenna array after pattern multiplication.
Eqn. 2.1 is the fundamental rule for the pattern multiplication of phased array,
and it applies only for arrays of identical elements. For instance, Fig. 2.2 shows the
configuration of the proposed pattern multiplication of a dipole antenna array with zero
phase difference (i.e. β = 0). Due to the cardioid form of the array factor, the final
direction of main beam is at zero degree.
-90⁰ +90⁰
0⁰
-90⁰ +90⁰
0⁰
×
+90⁰
0⁰
-90⁰=
8
For the N-element linear antenna array with uniform amplitude and spacing, the
final array factor (AF) can be written as Eqn. 2.2.
𝐴𝐹 = 1 + 𝑒𝑗(𝑘𝑑 cos𝜃+𝛽) + 𝑒𝑗2(𝑘𝑑 cos𝜃+𝛽) + ⋯+ 𝑒𝑗(𝑁−1)(𝑘𝑑 𝑐𝑜𝑠 𝜃+𝛽)
𝐴𝐹 = ∑ 𝑒𝑗(𝑛−1)(𝑘𝑑 cos𝜃+𝛽)𝑁𝑛=1 , 𝑘 = 2𝜋
𝜆 (2.2)
Here, 𝜆 is the wavelength of the signal, 𝑑 and 𝛽 is the distance and phase difference
between two adjacent antenna elements respectively, 𝜃 presents the direction of the
radiation beam. Finally, the above equation can be summarized to Eqn. 2.3.
𝐴𝐹 = ∑ 𝑒𝑗(𝑛−1)𝜓𝑁𝑛=1 , 𝜓 = 𝑘𝑑 cos𝜃 + 𝛽 (2.3)
In most of phased array design, the direction of main beam in radiation pattern is
an important and necessary parameter. To get the exact value of main beam’s angle,
the maximum value of array factor is the solving key. According Eqn. 2.3, when the
design parameter 𝜓 is equal to 2𝑚𝜋 (𝑚 = 0, ±1, ±2, ±3 … …), the array factor can always
reach its maximal value. As a result, the main beam’s angle can be written as Eqn. 2.4.
𝜃𝑚𝑎𝑖𝑛 = cos−1 �2𝑚𝜋−𝛽𝑘𝑑
� , 𝑚 = 0, ±1, ±2, ±3 … … (2.4)
Eqn. 2.4 is a very important equation and shows clearly the relationship between
beam angle 𝜃, phase difference 𝛽, and element spacing 𝑑. These three parameters are
applied extensively in the phased arrays presented in subsequent chapters.
2.3 Classification of Phased Array
As discussed in the last section, a general phased array is usually composed of
a feeding network and an antenna array. In phased array, the feeding network is used
to distribute the input signal into each radiation element in an antenna array. Based on
9
Fig. 2.3 General diagram of the space feeding.
different design purpose, the feeding network can generate various magnitude
distributions and phase difference. Generally speaking, the classification of phased
array is mainly based on the architecture of the feeding network. For the design of the
feeding network, there are almost as many ways as there are arrays in existence. There
are three basic categories of feeding network: space feeding, constrained feeding, and
semi-constrained feeding which is a hybrid of the space and constrained feeding [1]. In
the space feeding (as shown in Fig. 2.3), it is usually achieved by a separate feed horn
located at an appropriate distance from the antenna array [2]. Because of the free
space between the feed horn and antenna element, this type of feeding network is not
good for planar phased arrays. On the other hand, the constrained feeding is
considered as the simplest way of feeding the antenna array. This category generally
includes some transmission lines and passive microwave devices. The constrained
feeding itself can be classified into two basic types: parallel-fed and Series-fed [3].
These two types are the most commonly used methods to design the feeding network.
Their structures are discussed below.
Feed Horn
Transmitted Signal
Antenna
10
Fig. 2.4 General diagram of the phased array applying parallel-fed network.
2.3.1 Parallel-fed Network
In parallel-fed network, which are often called corporate feed, the input signal is
divided by a corporate tree network to feed into all the antenna elements as shown in
Fig. 2.4. This parallel-fed network typically employs power dividers and phase shifters
[4]. In an N-element phased array using parallel-fed network, the required amount of
power divider and phase shifter is 𝑁 − 1 and 2𝑁 respectively. As a result, the
performance of phased array with parallel-fed network critically depends on the
architecture of the power dividers and phase shifters. These two components are also
the major source of the phased array’s insertion loss.
2.3.2 Series-fed Network
In a feeding network using series-fed, the input signal is fed from one end of the
feeding network and coupled serially to the antenna elements as shown in Fig. 2.5. The
phase shifter is just deployed between two adjacent antenna elements. The main
advantage of series-fed network is it features more compact size than the parallel-fed
FeedingNetwork
Input Signal
PowerDivider
Phase Shifter
Antenna Array
11
Fig. 2.5 General diagram of the phased array applying series-fed network.
network, which can result in less insertion and radiation losses of the phased array [5].
Moreover, the series-fed network can relax the design constraints on the phase tuning
range of the phase shifters. In an N-element phased array with series-fed network, the
required number of phased shifters is usual less than that of the phased array with
parallel-fed network (normally by a factor of (𝑁 − 1)). However, the loss introduced by
the consecutive phase shifters indicates that it is impossible to generate identical
magnitude to antenna elements using such as topology. The cumulative loss of phase
shifters should also be an issue in the design of series-fed phased arrays with a large
amount of antenna elements.
Phase Shifter
Input Signal
Feeding network
AntennaArray
12
CHAPTER 3
NEW APPLICATION OF PHASED ARRAY ANTENNA FOR RADAR RESPIRATION
MEASUREMENT
3.1 Background and Motivation
Lung cancer is a disease characterized by uncontrolled cell growth in tissues of
the lung. The most common cause of lung cancer is long-term exposure to tobacco
smoke, which cause 80-90% of lung cancers. Besides smoking, genetic factors, radon
gas, asbestos, and air pollution including second-hand smoke are also the causes.
According to the American Cancer Society, 15% of people in the United States
diagnosed with lung cancer survive five years after the diagnosis. Worldwide, lung
cancer is the most common cause of cancer-related death in human being, and is
responsible for approximate 1.38 million deaths annually.
In the modern medical technology, there are several approaches for treating lung
cancer. The common treatments include palliative care, surgery, chemotherapy, and
radiotherapy. According the clinical examination, the radiotherapy can provide a better
treatment with less harm and shorter time period compared to other methods. Moreover,
studies have shown that an increased radiation dose to the tumor will improve the local
control and survival rates. However, the location of lung tumor is not fixed because of
the human respiration. In fact, the lung tumor can move significantly by 2-3cm with
respiration. The respiratory tumor motion has been a major challenge in radiotherapy to
deliver sufficient radiation without damaging to the surrounding healthy tissue [6]-[7]. In
order to address the above issue, some methods have been designed and presented in
13
the literature [8]-[9]. To improve the performance of radiotherapy, the accurate tumor
tracking should be realized.
Radar technology has been extensively used in both civil and military
applications including remote sensing, search and rescue, and imaging [10]. Recently,
radar technology is found to be an effective approach to achieve the tumor tracking
function with acceptable accuracy. By applying radar technology, it is able to obtain
non-invasive and non-contact detections of many useful parameters for medical
diagnosis and treatment. Up to now, several radar prototypes have been developed for
this purpose [11]-[12]. To further improve the performance of the radar based technique
for respiration measurement and tumor tracking, phased array antenna will be applied in
the radar system. In this way, it is available to steer the radar beam to simultaneously
measure the physiological signals at the patients’ chest and abdomen (as shown in the
top of Fig. 3.1), from which accurate tumor location can be derived.
Fig. 3.1 General schematic of phased array antenna integrated with radar system.
Radar System
Phased Array
Feeding Network
Beam Scanning
14
Fig. 3.2 Configuration of Butler matrix network (P1-P4 are input ports, and P5-P8 are
output ports).
3.2 Design and Analysis
The general schematic of proposed radar system is shown in Fig. 3.1, where the
developed phased array is integrated with a Doppler radar [13] for tumor tracking. By
applying this method, multiple beams can be achieved in the final radiation pattern. This
feature can help doctors measure multiple locations on the patients’ body in real time.
As a result, this beam-scanning radiotherapy system can significantly improve the
accuracy of tracking the tumor’s motion.
To realize the proposed multiple beam scanning, a Butler matrix is applied as the
feeding network in the phased array system [14]. The Butler matrix is a type of beam-
forming network [15]. From Fig. 3.2, the common 4×4 Butler matrix consists of four 3dB
hybrid couplers, two 45° phase shifters, two 0dB crossovers, and some phase-adjusting
transmission lines. Arranging these components with a special layout, the Butler matrix
can generate four output signals with equal power levels and the progressive phases
15
Table 3.1
Phase response of the output signal of the Butler matrix network
(Table 3.1). By integrating the Butler matrix with a four-element antenna array, four
distinct radiation beams will be achieved. Hence, one can switch the direction of the
radiation main beam from one to the other by exciting the designated input port. The
proposed phased array antenna system with the Butler matrix network has the following
typical characteristics. First, the number of radiation beams is equal to the number of
the antenna elements. Second, the Butler matrix’s inputs are isolated from each other.
Third, none of the inputs can provide a broadside beam. Finally, there is low insertion
loss in the whole system.
As discussed in the previous paragraphs, the proposed phased array antenna
system consists of a 4×4 Butler matrix and an antenna array. For easily integrating
Butler matrix with the antenna array on the same substrate, the RT/duroid 5880
substrate with the substrate thickness of 0.787mm and the dielectric constant of 2.2 is
employed to implement the whole system. To verify the proposed design concept, a
Butler matrix working at 5.8GHz is simulated in HyperLynx (previously called IE3D).
Since the line width of 50Ω microstrip line is too wide to effectively design the
components in the Butler matrix network, the whole Butler matrix is first realized by
100Ω microstrip lines (will be transformed back to 50Ω later using transformers).
Input Phase difference
P1 +45°
P2 -135°
P3 +135°
P4 -45°
16
Fig. 3.3 The topology of the proposed Butler matrix network (P1-P4 are input ports, and
P5-P8 are output ports).
Fig. 3.3 shows the topology of the proposed Butler matrix network in HyperLynx. Since
100Ω microstrip lines are applied in the Butler matrix, some transformers are employed
at input ports and output ports of the Butler matrix to match the 50Ω ports and antenna
array. The 90° coupler is realized by two pairs of 100Ω and 70.7Ω quarter-wavelength
microstrip lines. The crossover is formed by a novel fully symmetrical structure [16]. For
both the outer lines and the inner crossed lines, the characteristic impedance is 50Ω.
For the electrical lengths, one-quarter section of the outer line is 54.73°, and one-
quarter section of the inner line is 90° at the working frequency (5.8GHz). The phase
shifter is realized by a common delay line with the 50Ω impedance and the 45° electrical
length. The transformer consists of two quarter-wavelength transmission lines with
impedances of 59.46Ω and 84.08Ω respectively. Moreover, the meandered microstrip
lines are employed to adjust the output phases of the Butler matrix, and then connected
with the antenna array.
Transformer45°phase shifter
90°coupler
Crossover
Phase-adjusting
P1
P2
P3
P4
P5
P6
P7
P8
17
By applying the HyperLynx, the simulation results of the Butler matrix are
obtained and plotted in Fig. 3.4. When the input port P1 is excited, the insertion losses
at four output ports (P5-P8) of the Butler matrix are approximately -6.6±0.2dB at 5.8GHz
(as shown in Fig. 3.4(a)). For both return loss and isolation, the value is less than -25dB,
which is shown in Fig. 3.4(b). In addition, Fig. 3.4(c) shows that the phase difference
between two adjacent output signals is +45° at the working frequency. When the input
port P2 is excited, the value of the insertion losses at four output ports is -6.5±0.2dB (as
shown in Fig. 3.5(a)). In Fig. 3.5(b), both return loss and isolation are less than -23dB.
Finally, the phase response of four output signals is shown in Fig. 3.5(c), and the phase
difference between neighboring output signals is -135° at 5.8GHz. Based on the
symmetric structure of the Butler matrix, it is expected that the other two excited input
ports (P3 and P4) can generate the same magnitude response and complementary
phase response (-45° and +135°). Therefore, integrating this Butler matrix with an
antenna array, four distinct radiation beams can be realized.
(a)
18
(b)
(c)
Fig. 3.4 Simulation results of (a) insertion loss, (b) return loss and isolation, (c) phase
response of the Butler matrix with P1 as the input port.
19
(a)
(b)
20
(c)
Fig. 3.5 Simulation results of (a) insertion loss, (b) return loss and isolation, (c) phase
response of the Butler matrix with P2 as the input port.
3.3 Fabrication and Measurement Results
To verify the design concept, the proposed 4×4 Butler matrix is integrated with a
four-element antenna array to form the phased array antenna system (as shown in Fig.
3.6). In the antenna array, there are four identical circular patch antennas with 50Ω
feeding lines. The distance between the centers of two adjacent antenna elements is
about half-wavelength at the working frequency (approximate 23mm). Based on the
relation between the phase difference and the main beam angle (as shown in Eqn. 2.4
of Chapter 2), the final values of the radiation beam’s angle are illustrated in Table 3.2.
Four radiation beams with distinct direction can be achieved by the proposed system.
21
Fig. 3.6 Topology of proposed phased array antenna system.
Table 3.2
Radiation beam angles of the proposed system by exciting input ports
Fig. 3.7 presents the photograph of the fabricated phased array antenna system.
Due to the four input ports of the Butler matrix network, a SP4T switch (Hittite
HMC344LC3) is employed between the phased array antenna and the signal source. By
applying suitable bias voltages, fast switching can be realized for controlling the
direction of the radiation beam. The final measurement results of radiation beam angles
are shown in Fig. 3.8. As the input port P1 is excited, the main beam points to -20°. The
AntennaArray
Butler Matrix
P1
P2
P3
P4
Input Port Beam Angle
P1 -15°
P2 +40°
P3 -40°
P4 +15°
22
Fig. 3.7 Photograph of proposed phased array antenna system with SP4T switch.
(a) (b)
(c) (d)
Fig. 3.8 Measurement results of main beam angle with (a) excited P1, (b) excited P2, (c)
excited P3, and (d) excited P4.
SP4TSwitch
-20º+35º
-45º +15º
23
direction of the main beam switches to +35° as the input port P2 is excited. For input
port P3 and P4, the main beam angle is -45° and +15° respectively. Because of the
fabrication tolerance, there are some angle shifts between the measurement results and
the calculated values. For all radiation beams, the SLL is larger than 7dBi. Hence, the
measurement results match well with the design concept.
3.4 Conclusion
A new application of phased array antenna system for radar respiration
measurement is presented. The Butler matrix is employed to form the feeding network
in the phased array system. By integrating a 4×4 Butler matrix with a four-element
antenna array, four distinct radiation beams can be achieved. To verify the proposed
design concept, a phased array antenna system is designed to work at 5.8GHz. By
fabricating and measuring experimental prototypes, the performance of the proposed
phased array antenna system shows a good agreement with the design concept. It is
expected that integrating this phased array antenna with radar system can easily steer
the radar beam to simultaneously track the patient’s physiological signals at different
positions. Hence, this new technique can be applied for real-time tumor tracking.
24
CHAPTER 4
A NOVEL COMPACT PHASED ARRAY ANTENNA SYSTEM BASED ON DUAL-BAND
OPERATIONS
4.1 Introduction
As mentioned before, the phased array antenna system mainly consists of a
feeding network and an antenna array. After the signal goes through the feeding
network to feed the antenna array, multiple radiation beams with distinct directions can
be realized. In the proposed phased array antenna system, the function of feeding
network is to provide the specific phase responses and magnitude level for each
antenna element. One common way to construct the feeding network is to apply a
Butler matrix. The Butler matrix is a type of passive beam-forming network and has
symmetrical structure with identical number of inputs and outputs. For example, a 4×4
Butler matrix is to generate four output signals with progressive phases and equal
power levels. By connecting this 4×4 Butler matrix with a four-element antenna array,
there will be four distinct radiation beams, which is also verified in last chapter (chapter
3). In general, to generate more radiation beams using a single-band phased array
antenna, a common approach is to raise the order of Butler matrix (e.g. using a 8×8
Butler matrix, or even more). Some literatures have discussed the design of large order
Butler matrices [17]-[19]. However, raising the order of Butler matrix will lead to more
microwave components and transmission lines. As a result, the cost and size of the
whole system must be increased, which is undesired. Therefore, it is meaningful to find
25
Fig. 4.1 Configuration of the proposed compact phased array antenna system (Note:
dashed lines represent radiation beams generated at the lower frequency band; solid
lines represent radiation beams generated at the higher frequency band).
novel design concepts to address this issue.
In recent years, due to the rapid advance in modern communications, there are
several new design requirements for electronic components such as compact size, low
cost, and multiband operations. Especially, different types of dual-band microwave
devices have been presented in the literatures [20]-[22]. Based on these dual-band
microwave devices, several new dual-band phased array antenna systems are also
realized [23]-[27]. Inspired by these works, it can be expected to exploit the dual-band
characteristics to achieve size reductions. Specifically, it is found that an appropriate
design of special dual-band phased array antenna system can realize non-overlapping
radiation beams at two desired frequencies. As shown in Fig. 4.1, a dual-band 4×4
feeding network integrated with a four-element dual-band antenna array can generate
as many radiation beams as a single-band eight-element antenna array system. In
InputPorts
Dual-bandFeedingNetwork
P1
P2
P3
P4
BeamScanning
AntennaArray
26
general, an N-element dual-band phased array antenna system can be used to achieve
2N radiation beams, leading to a great size reduction.
In the proposed compact phased array antenna system (a 4×4 dual-band array
is studied), the 4×4 Butler matrix is employed as the feeding network. The typical
characteristics of the whole system are summarized as follows. First, the dual-band 4×4
Butler matrix integrated with antenna array can exhibit eight distinct radiation beams
across a wide frequency ratio range, as shown in Fig. 4.1. Second, the optimal
frequency ratio (𝑓2/𝑓1 ) plays an important role to achieve non-overlapping radiation
beams. Third, this phased array antenna can realize a significant size reduction
compared to the conventional single-band system. Finally, concise design equations are
applied for each constituting components of the proposed system. It is found that both
simulation and measurement results can match well with the design theory.
4.2 Design and Analysis
For the proposed compact phased array antenna system, the Butler matrix is an
essential component. As discussed in the last chapter, a conventional 4×4 Butler matrix
is composed of four 90° couplers, two 45° phase shifters, and two crossovers. However,
most losses of the Butler matrix are caused by these microwave components. To
achieve less losses and better size reduction, a novel modified 4×4 Butler matrix is
applied [28]. From Fig 4.2, this novel 4×4 Butler matrix only consists of 90° couplers and
45° phase shifters. In this way, the return losses introduced by crossovers can be
eliminated, and the complexity of the whole system is also reduced. Therefore, the
performance of the phased array antenna system can be improved.
27
Fig. 4.2 Configuration of the modified 4×4 Butler matrix with four inputs (P1-P4) and four
outputs (P5-P8).
The approach to realize the compact size is to apply the dual-band operation in
the whole phased array antenna system. To achieve dual-band operation, both the 4×4
Butler matrix and the antenna array need to be dual-band. In the proposed system, the
dual-band Butler matrix requires all microwave components to generate opposite phase
response and uniform power response at the two operating frequencies.
As mentioned previously, the frequency ratio is an important design factor in the
whole system. Based on the structure of Butler matrix, it is assumed that the phase
differences of the output signal are ±45° and ±135°. Then, by applying the array factor
equation (Eqn. 2.4), the main beam angle θ can be obtained as the following:
For phase difference β=±45°, cos 𝜃 = �𝑛 ± 18�𝑁 (4.1)
For phase difference β=±135°, cos 𝜃 = �𝑛 ± 38�𝑁 (4.2)
Where n is an integer number (0, ±1, ±2, … …), 𝑁 = 𝜆𝑑
with d as the inter-element
spacing and λ as the wavelength at working frequency (Note: in the proposed design,
since there are two working frequencies, there will be two different N corresponding to
P1
P2
P3
P4
90° coupler
Phase shifter
P5
P6
P7
P8
45°
45°
28
these frequencies). Based on calculations using the above listed equations, the relation
between radiation beam angle θ and the ratio parameter N is plotted in Fig. 4.3.
Fig. 4.3 Calculated radiation beam angle θ versus ratio parameter N.
As shown in Fig. 4.3, the different values of N can obtain different values of
radiation beam angle θ. Moreover, to avoid generating some grating beams, the final
optimal range of N is between 8/5 and 8/3 (out of this range, there will be multiple main
beam angles generated according to Eqn. 4.2). In the proposed dual-band based
compact phased array antenna system, there are two unique ratio parameters N1 and
N2 corresponding to the lower frequency 𝑓1 and the higher frequency 𝑓2, because d are
the same at these two frequencies. Therefore, according to Fig. 4.3, there should be
four distinct radiation beam angles at each of the dual operating frequencies (𝑓1 and 𝑓2)
for each phase difference of β=±45° and β=±135°. Consequently, with the dual-band
operation, the proposed array system can generate eight non-overlapping radiation
beams, achieving the same function as the conventional phased array antenna system
±10°
±20°
±30°
±40°
±50°
±60°
±70°
±80°
1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8
Rad
iatio
n be
am a
ngle
θ(d
egre
e)
Ratio parameter N
β=±π/4β=±3π/4
29
using single-band 8×8 Butler matrix.
To obtain the optimal range of frequency ratio (𝑓2/𝑓1), 𝜆1 and 𝜆2 are set as the
wavelengths at 𝑓1 and 𝑓2, respectively. Since d is the same at the two frequencies of f1
and f2 (𝑑 = 𝜆1𝑁1
= 𝜆2𝑁2
), the frequency ratio is 𝑓2/𝑓1 = 𝑁1/𝑁2. Based on the optimal range of
N as mentioned in the previous paragraph (i.e. N is between 8/5 and 8/3), the optimal
range of the frequency ratio can be obtained, which is 1< 𝑓2/𝑓1<5/3. Within this optimal
range of frequency ratio, non-overlapping radiation beams can be achieved for a given
inter-element spacing in the antenna array. In addition to that, the realizable frequency
ratio range of all constituting components of the proposed phased array antenna system
is larger than 1< 𝑓2/𝑓1 <5/3. Hence, all components can guarantee the stable
performance. In the designed prototype, 𝑓2 = 8GHz with 𝑁2 = 1.625 and 𝑓1 = 5GHz with
𝑁1 = 2.6 are chosen as the two operating frequencies. The final progressive phase
difference at output ports of the dual-band Butler matrix and the corresponding
generated radiation beam directions (assuming the inter-element spacing in antenna
array is 23mm) are illustrated in Table 4.1. In order to obtain the total layout of this
compact phased array antenna system, the design procedures of each constituting
component are presented in the following sections.
Table 4.1
Phase difference (𝜟Ф) of the Butler matrix and beam angle
Input Port 𝜟Ф@𝒇𝟏 Beam Angle 𝜟Ф@𝒇𝟐 Beam Angle
P1 +45° +18° -45° -11°
P2 -135° -51° +135° +34°
P3 +135° +51° -135° -34°
P4 -45° -18° +45° +11°
30
Fig. 4.4 Schematic of the dual-band 90° coupler.
4.2.1 Dual-band 90° Coupler
In a Butler matrix, 90° coupler is an essential element. To realize the dual-band
coupler, a branch line coupler with dual-band operation capability is employed [29]. Fig.
4.4 shows the schematic of the proposed dual-band 90° coupler. The electrical length of
each branch-line is selected to be integer multiple of quarter-wavelength evaluated at
the center frequency of the two operating frequencies (e.g. at (𝑓2 + 𝑓1)/2). To obtain all
values of design parameters, the following equations apply:
𝑓(𝛽) = 8𝐾𝑐+𝑐′
+ �1 + 2𝑐𝑐+𝑐′
� �1 + 2𝑐′
𝑐+𝑐′� = 0 (4.3)
𝑍1 = �𝑐+𝑐′
2 , 𝑍2 = 𝑍1
𝛽 (4.4)
where,
𝐾 = �cot𝜃 − 𝛽 tan𝜙2� �cot 𝜃 + 𝛽 cot
𝜙2�
𝑐 = 1 + 𝛽 tan𝜙2�2 cot𝜃 − 𝛽 tan
𝜙2� , 𝑐′ = 1 − 𝛽 cot
𝜙2�2 cot 𝜃 + 𝛽 cot
𝜙2�
𝜃 =𝑛𝜋2
(1 + 𝛿),𝜙 =𝑚𝜋
2(1 + 𝛿),𝛿 =
𝑓2 − 𝑓1𝑓2 + 𝑓1
P1
P4
P2
P3
31
n and m are positive integers; 𝑓2 and 𝑓1 are upper and lower bands respectively; Z1 and
Z2 are normalized values with reference to Z0. Based on the fractional bandwidth (𝛿), the
first step is to select appropriate values of m and n. Then, next step is using Eqn. 4.3 to
get the value of 𝛽. Finally, the characteristic impedance (Z1 and Z2) of each branch-line
can be obtained from Eqn. 4.4. This dual-band 90° coupler can exhibit low insertion loss
and good port isolation over a fractional bandwidth from 0.2 to 0.43. In the proposed
design, the characteristic impedance (Z1) and electrical length of the horizontal branch-
lines are 40.16Ω and 90° respectively, while the vertical branches have characteristic
impedance (Z2) of 75.4Ω and electrical length of 180°.
Fig. 4.5 Dual-band 45° phase shifter with T-shape structure.
4.2.2 Dual-band 45° Phase Shifter
Phase shifter is another imperative component in the proposed system. In the
proposed system, dual-band transmission lines are applied as the phase shifter. Fig. 4.5
shows the configuration of the dual-band 45° phase shifter with T-shape structure [30].
Z1 and Z2 are the characteristic impedances of series and shunt lines respectively, while
θ1 and θ2 are the corresponding electrical lengths at the lower operating frequency. To
obtain the design parameters, the ABCD-matrix of the T-shape structure is derived as
follows:
Z1, θ1
Z2, θ2
Z1, θ1P1 P2
32
2
21111
21
2 tancossinsincosZ
ZDA TTθθθ
θθ⋅⋅⋅
−−== (4.5)
2
2122
1111
tansincossin2Z
ZjZjBTθθ
θθ⋅⋅⋅
−⋅⋅⋅⋅= (4.6)
2
212
1
21 tancoscossin2Z
jZ
jCTθθθθ ⋅⋅
+⋅⋅⋅
= (4.7)
From Eqn. 4.5-4.7, replacing θ1 and θ2 with nπ - θ1 and 2mπ - θ2 (n and m are integer)
can make 𝐴𝑇 remain the same and 𝐵𝑇 get the inverse result. This is the key to realize
the dual-band operation of the proposed phase shifter. In the proposed design, the
electrical lengths of the two stubs are “θ1 and θ2” and “nπ - θ1 and 2mπ - θ2” at the lower
and higher frequencies respectively. After some calculations, the design formula of θ1
and θ2 can be derived to be 𝜃1 = 𝑛𝜋𝑓1 (𝑓1 + 𝑓2)⁄ , 𝜃2 = 2𝑚𝜋𝑓1 (𝑓1 + 𝑓2)⁄ at the lower
frequency. By applying the specific relationship which is AT = AP and BT = BP (AP and BP
are parameters in the ABCD-matrix of the ideal 45° phase shifter), the values of Z1 and
Z2 can be obtained. Finally, the proposed dual-band 45° phase shifter has θ1 = θ2 =
138.46°, Z1 = 22.08Ω, and Z2 = 41.84Ω.
4.2.3 Dual-band Antenna
To achieve a dual-band antenna array, a novel dual-band circular antenna is
applied [31]. Fig. 4.6 shows the structure of dual-band circular antenna with an offset
open-ring slot. The lower operating frequency depends on the radius R of the large
circular patch, while the higher one is dependent on the radius r of the patch enclosed
by the open-ring slot. The position of the offset open-ring slot needs to be adjusted to
keep both feed positions of the large circular patch and the additional one enclosed by
33
Fig. 4.6 Schematic of the dual-band circular antenna.
the open-ring slot at the same point. By selecting various values of the radius r and the
angle ϕ, the frequency ratio can be tuned. For the proposed system, the frequency ratio
of the antenna is 𝑓2/𝑓1 = 1.6. Correspondingly, the antenna design parameters are: R =
10.78mm, r = 6.78mm, and ϕ = 6°. The position of the feed point needs be adjusted to
make sure the similar broadside radiation patterns are realized at the two working
frequencies.
(a) (b)
(c)
Fig. 4.7 Photographs and structures of the fabricated phased array antenna system: (a)
top view, (b) bottom view, and (c) cross-sectional view.
R
r
feedφ
19.4cm
12.1
cm
P1
P2
P3
P4
Antenna array
Ground
Feeding network
Feed-through pin
Substrate
34
4.3 Simulation and Measurement Results
To verify the proposed design concept, a phased array antenna system operating
at 𝑓1= 5GHz and 𝑓2= 8GHz is designed and fabricated using RT/Duroid 5880 substrate
with the substrate thickness of 0.787mm and the dielectric constant of 2.2. Due to the
structure of Butler matrix, a two-layer layout is applied. In this layout, the dual-band 4×4
Butler matrix is on the bottom layer, while the dual-band antenna array is arranged on
the top layer. The antenna array is connected to the Butler matrix by employing two
back-to-back identical substrates with feed-through vias. Fig. 4.7 shows the
photographs and structures of the fabricated phased array antenna system. The
simulation results of the designed dual-band 4×4 Butler matrix are shown in Fig. 4.8.
When the input port P1 is excited, the insertion losses of four output signals are
7.2±0.5dB at 5GHz and 8.0±0.5dB at 8GHz (as shown in Fig. 4.8(a)). Meanwhile, the
insertion losses of four output signals are equal to 7.0±0.5dB at 5GHz and 8.2±0.2dB at
(a) (b)
Fig. 4.8 Simulated responses of output signals (magnitude) of the Butler matrix with (a)
P1 as the input port and (b) P2 as the input port.
35
8GHz when the input port P2 is excited (as shown in Fig. 4.8(b)). Based on the
symmetry of Butler matrix, similar results can be obtained when the other two input
ports (P3 and P4) are excited. In Fig. 4.9, the simulated phase responses of the dual-
band 4×4 Butler matrix are plotted. It is observed that, when the port P1 is excited, the
progressive phase difference at the four output ports is +45° at 5GHz and -45° at 8GHz.
By exciting the port P2, the phase difference is -135° and +135° at 5 and 8GHz,
respectively. For the input port P3 and P4, complementary phase differences will be
(a) (b)
(c) (d)
Fig. 4.9 Simulated phase responses of the Butler matrix with (a) input from P1 at 5GHz
(b) input from P1 at 8GHz, (c) input from P2 at 5GHz, and (d) input from P2 at 8GHz.
S51S61S71S81
S51S61S71S81
S82
S52S62S72
S52S62S72S82
36
(a) (b)
Fig. 4.10 Simulated and measured return losses of the proposed phased array antenna
system at (a) 5GHz and (b) 8GHz.
generated at two frequencies. These results agree well with the design goal listed in
Table 4.1. In the antenna array, the distance between two adjacent antenna elements is
equal to λ/2.6 at 5GHz. The overall performance of the fabricated prototype is
experimentally characterized. In Fig. 4.10, it shows the simulated and measured return
losses of the proposed system are below -16dB at 5GHz and lower than -20dB at 8GHz,
showing good matching of the overall system.
The simulated and measured radiation patterns of the proposed phased array
antenna system are shown in Fig. 4.11. At 5GHz, the main beam points to +25° and -50°
when the input ports P1 and P2 are excited respectively. At 8GHz, the main beams are
at -15° and +35° when P1 and P2 are excited. Due to the symmetrical structure of the
proposed phased array, when P3 and P4 are excited, complementary radiation beams
at +50°, -35°, -25°, and +15° are generated. Because of the fabrication tolerance, there
is beam angle shift between the measured and calculated values. For all main beams,
37
the side lobe level is below 4dB, and the antenna gain measurements are shown in
Table 4.2. Overall, the measurement results are in good agreement with the simulation
results, verifying the proposed design concept.
Fig. 4.11 Simulated and measured radiation patterns at two frequencies.
38
Table 4.2
Antenna gain measurements at two frequencies
Excited Input Port At 5GHz At 8GHz P1 6.2dB 5.3dB
P2 5.4dB 5.2dB
P3 6.1dB 5.0dB
P4 5.4dB 5.4dB
4.4 Conclusion
This chapter has demonstrated a novel compact phased array antenna system
based on dual-band operations. To realize the function of dual-band operations, a dual-
band Butler matrix is integrated with a dual-band antenna array. There are eight distinct
non-overlapping beams generated at two operating frequencies by using a 4-antenna
array. To prove the design concept, a phased array antenna system operating at 5 and
8GHz is designed and tested. Four main beams appear at ±25°, ±50° at 5GHz, while at
8GHz the main beams point to ±15°, ±35°. In addition, a significant size reduction
(about 53% at the higher frequency and 70% at the lower frequency) can be achieved
by the proposed compact phased array antenna system. The simulated and measured
results agree well, demonstrating a significant size reduction.
39
CHAPTER 5
A NEW ARCHITECTURE OF STEERING PHASE FEEDING NETWORK BY USING BI-
DIRECTIONAL SERIES-FED TOPOLOGY
5.1 Introduction
As mentioned before, the number of radiation beams mainly depends on the
range of phase difference at the feeding paths to the antenna array. The phase
difference is directly generated from the feeding network. In the feeding network, the
most important component is phase shifter. To get more radiation beams, the range of
phase difference should be increased. In other words, more phase shifters need to be
employed in the feeding network. However, large number of phase shifters and the
resulting high circuit complexity is one of the main reasons behind the high cost and
complexity of phased array system. Moreover, the cost and complexity of the individual
phase shifters in a phased array system also depend on the required range of phase
difference. Therefore, less phase shifters and smaller range of phase difference can
significantly reduce both the cost and complexity of phased arrays.
In the previous two chapters, the Butler matrix is employed as the feeding
network. The main limitation is the amount of radiation beam is directly proportional to
the maximum number of input ports in the Butler matrix. In this chapter, a new
architecture of feeding network is introduced which demands less phase shifter and
smaller phase difference range compared to any of the conventional feeding network
designs. In the proposed feeding network, two novel technologies are applied. First, a
new design of phase shifter is used to replace the conventional delay lines. The main
40
disadvantage of conventional delay lines is that the electrical length is fixed at the
working frequency. In contrast, the proposed phase shifter can steer the electrical
length by a single control voltage. Second, the proposed feeding network applies a new
bi-directional structure. This structure can decrease the require range of phase
difference at the output ports, leading to a significant size reduction. By applying these
techniques, the proposed feeding network can realize a compact size and steer the
phase difference. Hence, it is expected that more radiation beams can be achieved by
integrating this feeding network with antenna array. In the next two sub-sections, the
detailed analysis about the bi-directional structure and the novel phase shifter are
presented.
5.2 Bi-directional Structure
In general, phased array is designed based on either a parallel or a serial feed
network [32]. In parallel feed network, which is often called corporate feed, the input
signal is divided in a corporate tree network to all the antenna elements. In phased
arrays using parallel feed network, the phase shift at each individual element is
controlled separately through a phase shifter. In general, for an N-element phased array
with a parallel feed network, to achieve phase difference β, the maximum phase shift of
(𝑁 − 1)𝛽 is required from phase shifters as shown in Fig. 5.1.
Series feed networks are simpler, more compact and with lower feed line losses
compared to parallel network [33]. In a series feed network, fed from one end of the
feed network, the input signal is coupled serially to the antenna elements. Also the other
41
Fig. 5.1 N-element phased array with a parallel feed network.
Fig. 5.2 N-element phased array with a series feed network.
end of the network is usually terminated with a matched load. In a phased array with a
series feed network, the phase shifters can be inserted either in parallel or in series
along the feed line [34]. In an N-element serially fed array, placing the phase shifters in
series rather than in parallel reduces the required number of phase shifts by a factor of
(𝑁 − 1) as depicted in Fig. 5.2.
To further reduce the required amount of phase shift, the bi-directionally fed is an
effective approach. Bi-directionally series-fed networks have been implemented using
both hybrid [34]-[35] and integrated circuits [36]. In the proposed design, a new feeding
network applying the bi-directional series-fed topology can reduce the required phase
ParallelFeed
Input Signal
Phase Shifter
N-elementAntenna Array
(N-1)β (N-2)β β 0
Input Signal
Series Feed
N-elementAntenna Array
Phase Shifterβ β β β β
42
Fig. 5.3 Configuration of the proposed feeding network integrated with antenna array.
shifts to half of the phase shifts needed in the conventional series feed network as
shown in Fig. 5.2. Furthermore, this novel approach can allow the same phase shifter
design to be used throughout the entire feeding network, substantially reducing the
complexity of the whole system. Fig. 5.3 shows the general configuration of the
proposed feeding network integrated with antenna array. By using a matched single
pole, double throw (SPDT) switch, the whole phased array can be fed from either end.
In principle, if the beam angle can be steered from 0 to θ when the input signal is
propagated from left path, the input signal going through right path can generate the
beam angle between – θ to 0. Hence, the radiation beam can be steered from – θ to θ.
Due to the couplers connected in series as shown in Fig. 5.3, the antenna
elements receive the unequal power level from the proposed bi-directional series-fed
network. In order to determine the optimum coupling factor (C), its effect on the output
amplitude excitation is investigated. Assuming all couplers are identical and phase
shifters are lossless, the normalized signal level at each antenna element is shown as
the following:
Input Signal
PowerCombiner
AntennaElement
Phase ShifterCoupler
SPDTBi-directional series-fed Network
43
𝑆�̅� = (1 − 𝐶)𝑁−1𝐶 (5.1)
where N is the antenna element number.
According the Eqn. 5.1, the signal amplitude at each antenna element drops
along the feeding line. Hence, it results in a linear amplitude tapering and affects the
radiation beam’s directivity. Assuming the antenna elements are omnidirectional with
half-wavelength spacing, the final formula of beam directivity (D) is written as Eqn. 5.2.
𝐷 = 𝐶1/2
1−(1−𝐶)1/2 �1−(1−𝐶)𝑁/2
1+(1−𝐶)𝑁/2 (5.2)
As given by Eqn. 5.2, the beam directivity (D) depends on the antenna element
number (N) as well as the coupling factor (C). For eight antenna elements, the beam
directivity (D) versus the coupling factor (C) is plotted in Fig. 5.4. As can be seen, by
reducing the coupling factor, one can improve the array’s directivity. However, the
power dissipated in the matched termination within SPDT switch increases as the
coupling factor is reduced. The relationship between the normalized power dissipation,
the coupling factor and the number of antenna elements is given in Eqn. 5.3.
𝑃�𝑑𝑖𝑠 = (1 − 𝐶)𝑁 (5.3)
Fig. 5.4 Beam directivity versus coupling factor for an 8-element antenna array.
-20 -16 -12 -8 -4 0
2
4
6
8
10
Coupling Factor “C” (dB)
Bea
m D
irect
ivity
“D
”(d
B)
44
Fig. 5.5 Input power dissipation versus coupling factor for an 8-element antenna array.
Fig. 5.5 shows the plot of the dissipated power percentage as a function of the
coupling factor for an 8-element antenna array. Reducing coupling factor can increase
the power dissipated at the termination of the SPDT switch. Based on the requirement
of the beam direction and power dissipation, the needed number of antenna elements
and coupling factor can be determined. In the proposed design, an 8-element antenna
array and a coupler with -7dB coupling are employed. As a result, 9dB directivity and 18%
power dissipation can be achieved.
5.3 New Phase Shifter Design
One of the key and essential components in most of phased array systems is
phase shifter. As discussed before, a major cost and complexity of phased array system
is due to the expensive phase shifters. Furthermore, performance of phase shifters can
significantly affect the performance of the whole phased array system. Critical
parameters in phase shifter design include phase tuning range, insertion loss and return
-20 -16 -12 -8 -4 0
20
40
60
80
100
Coupling Factor “C” (dB)
Inpu
t Pow
er D
issi
patio
n (%
)
45
Fig. 5.6 The circuit diagram of proposed phase shifter.
loss [38]. In recent years, a variety of phase shifters have been applied in the phased
array. The main approaches to design phase shifters are based on a varactor loaded
line, vector modulated, reflective type, and switch based phase shifters. Varactor loaded
line is the most popular method in phase shifter design. In this method, the electrical
length of the line is varied by tuning the varactors. There are various varactor
technologies that have been used in phase shifters, such as BST varactors [39]-[41],
MEMS varactors [42]-[44] or solid state varactors [45]-[47].
In the proposed bi-directional series-fed network, the phase shifter is designed
based on the extended resonance technique [48], which is shown in Fig. 5.6. This
phase shifter consists of two varactors and one inductor with a π-shape structure. The
phase shifter is assumed to be matched to Z0 (Z0=1/Y0). A varactor with a susceptance
(jB) is connected in shunt at the input node. The admittance seen at this node (Y0+jβ) is
transformed to its conjugate value (Y0-jB) by applying a series reactance (jX). An
identical varactor with jB is added in shunt to cancel out the imaginary part of the
impedance. As a result, the final relation between design parameters of phase shifter
can be derived as the following:
Y0-jB Y0
Y0 Y0+jB
jB jB
jX
46
1𝑗𝑋+ 1
𝑌0+𝑗𝐵= 𝑌0 − 𝑗𝐵 ⇒ 𝑋 = 2𝐵
𝑌02+𝐵2 (5.4)
Applying the ABCD-matrix of the π-shape structure and Eqn. 5.4, the phase
response of the proposed phase shifter can be written as the following:
𝛥𝜃 = 𝑎𝑟𝑐𝑡𝑎𝑛 �1+2𝐵𝑍0−𝐵2𝑍02
2𝐵𝑍0−2� (5.5)
Fig. 5.7 The required inductance versus the varactor capacitance.
To tune the phase shift, the susceptance (jB) is controlled by the varactor bias
voltage. In general, as the varactors are tuned, the series reactance (jX) should also be
tuned to satisfy the matching condition. However, in the proposed phase shifter, the
series reactance will be fixed for the purpose of easy implementation. To avoid
increasing the insertion loss of the phase shifter, an appropriate value of inductance
should be taken into account. In the proposed design, the phase shifter works at
5.8GHz with the characteristics impedance of 50Ω. Fig. 5.7 shows the required value of
0 1 2 3 4 5
0.2
0.6
1
1.4
Varactor Capacitance (pF)
Indu
ctan
ce (n
H)
47
inductance versus the capacitance of varactor. To obtain a small insertion loss, the
value of inductance should be chosen at the peak point in Fig. 5.7. In other words, the
optimal value of series reactance should be equal to the characteristics impedance
(Xopt=Z0). The insertion loss of phase shifter (within the tuning range of varactor
capacitance) versus the phase shift is plotted in Fig. 5.8. As expected, as the phase
shift is increased, the insertion loss of phase shifter is also enlarged. In practice, by
choosing an appropriate range of varactor capacitance, the insertion loss can be held in
a small value.
Fig. 5.8 The insertion loss of phase shifter versus the phase shift.
5.4 Simulation Results
Based on the design procedure described above, a novel feeding network using
the bi-directional structure is designed and simulated at 5.8GHz. Fig. 5.9 shows the
topology of the proposed feeding network drawn in Advanced Design System (ADS). In
this feeding network, the phase shifter is implemented using varactor diodes and a
lumped inductor. The varactor capacitance used in the design is from 0.4 to 2.5pF with
the inductor value of 1.4nH. There are eight branches as the output ports, and each
60 80 100 120 140-12
-8
-4
0
Phase Shift (degree)
Inse
rtio
n Lo
ss (d
B)
48
Fig. 5.9 Topology of the proposed bi-directional feeding network.
branch includes a coupler and a power combiner. The coupler is CP0603 from AVX with
a coupling factor of -7dB, and the power combiners is from Mini-Circuits (SCN-2-65+) to
combine the power from the output ports of the couplers.
Fig. 5.10 shows the simulated phase difference through the proposed feeding
network when the input signal is input from the left end. As the value of varactor
capacitance is increased from 0.4pF to 2.5pF, the value of achievable phase difference
will be changed from -74° to -236°. Due to the bi-directional structure, the phase
difference from 74° to 236° can be realized by the input signal coming from the right end.
By integrating the proposed feeding network with 8-element antenna array, a wide
range of radiation beam can be achieved.
49
Fig. 5.10 Simulated phase difference of the proposed feeding network versus the
varactor capacitance as the signal is input from the left end.
5.5 Conclusion
A new architecture of feeding network for phased array systems is presented in
this chapter. Two novel techniques are applied in the proposed feeding network: bi-
directional structures and phase shifters using π-shape structures. The bi-directional
structure can reduce the required number of phase shifters, leading to a significant
reduction on the complexity of the whole system. Moreover, a novel phase shifter is also
introduced and designed. This type of phase shifter consists of varactors and inductor
with a π-shape structure. Using a single control voltage, the phase shifter can be tuned
easily. By integrating the proposed feeding network with an antenna array, one can
steer the radiation beam by controlling the tuning voltage and achieve a wide range of
beam-scanning. To verify the theory, a feeding network using bi-directional series fed
topology is designed and simulated. The simulation results verify the design concept.
0.5 1 1.5 2 2.5-250
-200
-150
-100
-50
Varactor Capacitance (pF)
Phas
e D
iffer
ence
(deg
ree)
50
CHAPTER 6
CONCLUSION AND FUTURE WORK
In this thesis, several novel approaches about the design and application of
phased array systems are presented. First, the principle of phased array is introduced,
and the fundamental design equations are also presented. Moreover, the phased array
is classified into different categories based on the feeding networks. For the application
of phased array antenna system, a new design is introduced for radar respiration
measurement. By integrating the phased array antenna with the radar system, multiple
radiation beams can be achieved to track the patient’s physiological signals at different
positions in real time. In addition to operating at single-band, phased array antenna is
also able to work at dual-band. By applying the dual-band function, N-element phased
array antenna system can generate 2N main beams in radiation pattern. It can achieve
the same function as the conventional single-band phased array antenna with doubled
antenna elements and realize a significant size reduction. Finally, a novel architecture of
phased array feeding network is presented. By applying the bi-directional structure and
a novel phase shifter, the proposed feeding network can allow one to steer the phase
shift by just applying a single control voltage.
In the future, in order to further improve the performance of phased array
systems, more different designs featuring characteristics such as compact size,
multiband, and multi-beam need to be investigated, the implementation of phased array
antennas in the Integrated Circuits (ICs) shows great potential for practical applications.
51
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