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Department of Science and Technology Institutionen för teknik och naturvetenskap Linköping University Linköpings universitet
gnipökrroN 47 106 nedewS ,gnipökrroN 47 106-ES
LiU-ITN-TEK-A--12/044--SE
Differential Six-PortTransceiver Design and
Analysis from a WirelessCommunication System
PerspectiveMuhammad Umar
Umair Yasir
2012-06-12
LiU-ITN-TEK-A--12/044--SE
Differential Six-PortTransceiver Design and
Analysis from a WirelessCommunication System
PerspectiveExamensarbete utfört i Elektroteknik
vid Tekniska högskolan vidLinköpings universitet
Muhammad UmarUmair Yasir
Handledare Magnus KarlssonExaminator Adriana Serban
Norrköping 2012-06-12
Upphovsrätt
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© Muhammad Umar, Umair Yasir
i
Differential Six-Port Transceiver Design and Analysis from a Wireless Communication System Perspective
Muhammad Umar
Umair Yasir
iv
Abstract
In modern telecommunication there is the demand of high data rates using wideband component
design. FCC has introduced the UWB spectrum for high speed data communication. UWB
systems have attracted the attention of researchers. Six-port transmitters and receivers are strong
candidates for UWB systems and research is being done on six-port modulators and
demodulators. In this work an effort is made to compare the performance of conventional single-
ended six-port transmitter and receiver with differential six-port transmitters and receivers.
In this thesis, single ended and differential six-port correlators are designed on 7.5 GHz using
Agilent Inc. EDA tool ADS and their performance is evaluated. A new wide-band differential
six-port correlator is implemented using rat-race couplers and double-sided parallel strip-line
phase inverter.
The designed six-port correlators are used for 8-PSK modulation and demodulation. For
transmitter-receiver system, mixed analog-DSP designing is used. The integral components of
the system are evaluated individually and behavioral modeling is used to evaluate the complete
transmitter-receiver system. The single-ended and differential systems are evaluated for noise-
figure, dynamic range, bit error rate and data rate.
v
Acknowledgements We would like to recognize the effort and support of following persons who have helped and
enabled us to successfully complete the thesis work:
Our examiner Dr. Adriana Serban for her support, guidance and giving us the opportunity to
work with her.
Our Supervisor Dr. Magnus Karlsson for his support in the thesis, especially in PCB fabrication
and measurements.
Mr. Gustav Knutsson for his help in PCB fabrication.
All WNE degree students specially Ionut-Alexandru Apolozan.
Never ending support of our beloved families who have always encouraged and supported us
throughout the life.
vi
List of Abbreviations
ADS Advanced Design System
ASK Amplitude Shift Keying
BLC Branch Line Coupler
BPF Band Pass Filter
DSP Digital Signal Processing
DSPSL Double Sided Parallel Strip Line
EDA Electronic Design Automation
EM Electromagnetic
EMI Electromagnetic Interference
FCC
Federal Communications
Commission
FSK Frequency Shift Keying
HPF High Pass Filter
I/Q In-Phase and Quadrature Phase
IC Integrated Circuit
IF Intermediate Frequency
LNA Low Noise Amplifier
LO Local Oscillator
LPF Low Pass Filter
NF Noise Figure
PA Power Amplifier
PCB Printed Circuit Board
PSK Phase Shift Keying
QAM
Quadrature Amplitude
Modulation
RF Radio Frequency
SMA Subminiature version A
SNR Signal to Noise Ratio
UWB Ultra Wide Band
VCO Voltage Controlled Oscillator
VGA Variable Gain Amplifier
VNA Vector Network Analyzer
WPD Wilkinson Power Divider
viii
Table of Contents
ABSTRACT ............................................................................................................................................................. IV
ACKNOWLEDGEMENTS .......................................................................................................................................... V
LIST OF ABBREVIATIONS ....................................................................................................................................... VI
TABLE OF CONTENTS ........................................................................................................................................... VIII
1 INTRODUCTION ............................................................................................................................................ 1
1.1 THE UWB COMMUNICATION TECHNOLOGY ............................................................................................................... 2 1.2 MOTIVATION AND OBJECTIVE OF THE THESIS ............................................................................................................. 3 1.3 METHOD ............................................................................................................................................................ 4 1.4 CONTRIBUTIONS ................................................................................................................................................... 4
2 THEORETICAL BACKGROUND ........................................................................................................................ 5
2.1 MODULATION SCHEMES ........................................................................................................................................ 5 2.1.1 Amplitude modulation ........................................................................................................................... 6 2.1.2 Phase modulation .................................................................................................................................. 6 2.1.3 Frequency modulation ........................................................................................................................... 9
2.2 TRANSCEIVER ARCHITECTURES ................................................................................................................................. 9 2.2.1 Transmitter designs ............................................................................................................................... 9 2.2.2 Receiver designs ................................................................................................................................... 12
2.3 DIFFERENTIAL SIGNALING ..................................................................................................................................... 14 2.3.1 Two wire signaling ............................................................................................................................... 15 2.3.2 Voltages and currents in differential signaling .................................................................................... 16 2.3.3 Differential impedance ........................................................................................................................ 17 2.3.4 Mixed-mode S-Parameters .................................................................................................................. 17 2.3.5 PCB structures for differential signaling .............................................................................................. 18
3 SIX-PORT CORRELATOR............................................................................................................................... 19
3.1 IDEAL SIX-PORT CIRCUIT ....................................................................................................................................... 19 3.1.1 Wilkinson Power Divider ...................................................................................................................... 20 3.1.2 Quadrature Branch line coupler ........................................................................................................... 21 3.1.3 180
o Branch Line Coupler ..................................................................................................................... 22
3.2 MODULATION USING SIX-PORT CORRELATOR ........................................................................................................... 23 3.3 DEMODULATION USING SIX-PORT CORRELATOR ....................................................................................................... 25
4 DESIGN AND IMPLEMENTATION OF SIX-PORT CORRELATOR ...................................................................... 29
4.1 SINGLE-ENDED DESIGNS ....................................................................................................................................... 30 4.1.1 Classical single-ended design ............................................................................................................... 30 4.1.2 Single-ended design with matching stubs............................................................................................ 35
4.2 DIFFERENTIAL DESIGNS ........................................................................................................................................ 38 4.2.1 Classical differential design ................................................................................................................. 38 4.2.2 Wideband differential design ............................................................................................................... 44
4.3 WIDEBAND 180O
COUPLER DESIGN ....................................................................................................................... 50
5 SIX-PORT MODULATOR AND DEMODULATOR DESIGN ............................................................................... 54
5.1 SIX-PORT MODULATOR IMPLEMENTATION ............................................................................................................... 54
ix
5.1.1 Variable port impedances .................................................................................................................... 55 5.1.2 8-PSK modulation using mixed analog-DSP designing ......................................................................... 55
5.2 SIX-PORT DEMODULATOR IMPLEMENTATION ............................................................................................................ 59 5.2.1 Diode modeling .................................................................................................................................... 60 5.2.2 Notch filter ........................................................................................................................................... 61 5.2.3 Digital judgment circuit ....................................................................................................................... 61
5.3 SIX-PORT TRANSMITTER-RECEIVER SYSTEM .............................................................................................................. 61
6 DESIGNED SIX-PORT TRANSCEIVER SYSTEM EVALUATION .......................................................................... 64
6.1 NOISE FIGURE COMPARISON ................................................................................................................................ 64 6.2 BER AND DYNAMIC RANGE COMPARISON .............................................................................................................. 65 6.3 BER AND DATA RATE COMPARISON ....................................................................................................................... 67 6.4 MODULATED SIGNAL CONSTELLATION DIAGRAMS AND POWER SPECTRUM COMPARISON ................................................ 67
7 CONCLUSION & FUTURE WORK .................................................................................................................. 70
7.1 CONCLUSION ..................................................................................................................................................... 70 7.2 FUTURE WORK .................................................................................................................................................. 70
8 REFERENCES ............................................................................................................................................... 72
1
1 INTRODUCTION
Wireless communication is ubiquitous nowadays. We can find its presence everywhere around
us. From indoor communication applications to long range communication systems which not
only connect the world together but can also communicate with the Rovers sent to Mars, wireless
communication has redefined the possibilities. There are numerous applications and standards
varying in data rate from some Kbits/s to Gbits/s and coverage distance from some meters to
hundreds of kilometers. The trend is to replace wired communication with wireless devices
which can communicate with similar efficiency and reliability. Advancements in the research
and design of wireless devices and techniques are going on for improving the reliability and
robustness in the communication and to introduce new innovations to the everyday life.
It all dates back to the prediction and description of Electromagnetic waves by James Clerk
Maxwell in 1873 [1]. Maxwell, on mathematical grounds, suggested that a varying Electric field
produces a varying Magnetic field and same in vice versa. Heinrich Hertz carried out a set of
experiments in 1887-1891 and validated the theory presented by Maxwell [2]. Guglielmo
Marconi, an Italian inventor, carried out first demonstration of wireless communication in 1895.
In 1940s, during the World War 2, research and development of RADAR (RAdio Detection And
Ranging) attracted much attention to the field of Wireless communication specifically
Microwave communication. Also, parallel advancements in the field of Solid state physics such
as invention of first Silicon transistor at Texas instruments in 1954 and its use as a switch in
electronic devices paved the way for development of complex digital communication devices.
The cellular mobile communication started in the late 1970s and early 1980s. The first
commercial analog cellular system was launched by NTT (Nippon Telephone & Telegraph) in
Tokyo, Japan in December 1979. In 1981 NMT (Nordic Mobile Telephone) introduced cellular
system in Nordic countries. Cellular mobile system started to evolve and expand rapidly. The
advent of 2G (2nd
Generation) systems in 1991 came with Digital version of cellular systems
which offered more spectral efficiency and security. The second generation got the name of
GSM (Global System of Mobiles).The consequent introductions of various data transmission
technologies and standards such as GPRS (General Packet Radio Service) and EDGE (Enhanced
Data rates for GSM Evolution) to GSM started a new era of wireless communication and spread
the use of cell phones dramatically.
In addition to these WANs (Wide Area Networks), short range wireless communication
networks or WPANs (Wireless Personal Area Networks) got evolved tremendously as well.
Bluetooth was introduced in 1999. It is a short range (1-100 meters) wireless alternative for
communication wires/cables. The technology’s latest version 3.0 + HS incorporates with 802.11
(Wi-Fi) and can serve up to data rates as high as 24 Mbit/sec. Before that version 2.0 + EDR was
able to provide 3 Mbit/sec data rate [3]. On the other hand IEEE (Institute of Electrical and
Electronic Engineers) developed specifications for 802.11 in 1997 [4]. 802.11 is a WLAN
(Wireless Local Area Network) specification and is popularly known as Wi-Fi. The technology
2
now has different versions namely 802.11 a, b, e, g and n with the latest in practice i.e. 802.11 n
can reach up to net data rates of 300 Mbit/sec.
Following is a summary of various short range communication standards and technologies:-
IEEE 802.11 a/b/g/n (Wi-Fi)
Bluetooth, IEEE 802.15.1
ZigBee, IEEE 802.15.4
UWB (Ultra wideband)
These technologies in terms of coverage distance and data rates are depicted in Figure 1.1
Figure 1.1 Different wireless communication technologies in terms of distance and data rates.
The last mentioned i.e. UWB is the focus of this thesis and will now be discussed in detail.
1.1 The UWB communication technology
Ultra-Wideband is a recent inclusion to the short range wireless communication technologies and
it was first authorized by FCC (Federal Communications Commission) of USA in 2002 [5]. FCC
allocated a wideband i.e. 3.1-10.6 GHz and with emission limitation on power spectral density of
-41 dBm/MHz. This limitation was implemented to reduce interference with other
communication technologies. A larger bandwidth and lower power offers a massive increase in
data rates compared to other narrowband short range standards such as Bluetooth and 802.11 Wi-
Fi.
The interrelation between channel capacity with bandwidth is described by Shannon equation:-
C = B × log2(1 + SNR)
Where C = channel capacity or data rate or throughput
B = channel bandwidth
SNR = Signal to noise ratio or
3
This means that for higher channel capacity we can either allocate more bandwidth or increase
the signal to noise ratio. Increasing SNR in wireless communication implies that we increase the
transmitting power which in turn can increase interference with other communication systems so
it is not desirable. Also its relation is logarithmic with the channel capacity. Then other option is
increasing bandwidth, which the UWB has employed.
The suggested bandwidth by FCC of 3.1-10.6 GHz is roughly 7 GHz. The first half of this band
overlaps with the unlicensed 5.725 – 5.875 GHz ISM (Industrial, Scientific and Medical) band.
Therefore in Europe, Asia and Japan there are further requirements of LDC (Low Duty Cycle)
and DAA (Detect and Avoid) techniques in the 3.1 – 4.8 GHz band to avoid interference with
existing technologies. EC (European Commission) has limited the UWB bandwidth for devices
without requirements for DAA mitigation techniques to 6-8.5 GHz. NICT (National Institute of
Information and Communications Technology) in Japan has divided the UWB band into separate
bandwidths of 3.4 - 4.8 GHz and 7.25 – 10.25 GHz. In the first band interference mitigation
techniques are required and further the allowed average transmission power is reduced to -70
dBm/MHz. In China, the approved bands for UWB operation are 4.2 - 4.8 GHz and 6 – 9 GHz
[6].
UWB communication can be categorized in two ways when it comes to transmission of channel
signaling i.e. (i) carrier free UWB communication (ii) carrier based UWB communication.
Carrier free UWB, also known as impulse radio UWB, the data is sent in the form of short or low
duty cycle pulses utilizing the whole allocated band. Carrier based UWB is further divided into
(a) single carrier (b) multi carrier. Example of single carrier is the use of DSSS (Direct sequence
spread spectrum) technique. Multi carrier UWB involves OFDM (Orthogonal Frequency
Division Multiplexing) technique in which signal is sent using multiple modulated orthogonal
carriers [7].
1.2 Motivation and Objective of the Thesis
Wireless communication in UWB requires new components designed and optimized for this
frequency band. A lot of research and design work has been going on designing LNA (Low
Noise Amplifier), Antenna (transmit/receive), Mixer, Band Pass filter, Power Amplifier, Six port
Direct carrier modulator and other components in the wireless communication system hierarchy,
for UWB band. Several studies have been carried out in Linköping University prior to this [6],
[8] and [9]. All of the above mentioned works were concentrated on communication system
architectures employing Six-port direct carrier modulator.
Six-port systems are made up of passive microwave components and are simpler than
conventional heterodyne radio transceivers involving IF (Intermediate Frequency) components.
The main advantages of Six-port correlator are of large frequency bandwidth and less power
consumption [10] [11]. Both these aspects make it a suitable choice for short range, high data
rate UWB systems.
As mentioned above, research in [8] was the study on implementing differential six-port
transceiver. Differential six-port promises the advantages of signal integrity, reduced common
mode noise, crosstalk suppression, compactness of the system and increased dynamic range [12].
4
The goal of this thesis is to implement 8-PSK (Phase shift keying) modulation/demodulation on
both single-ended and differential six port designs with a central frequency of 7.5 GHz. The
objective is the system level implementation of both designs and their comparison from a
wireless communication system perspective.
1.3 Method
The EDA (Electronic Design Automation) software tool used for the simulation tasks is ADS
(Advanced Design Systems) version 2011 from Agilent Inc. The bandwidth chosen among the
total UWB bandwidth is 6 – 9 GHz. This bandwidth is not subjected to the requirement of
interference mitigation techniques by any country’s laws. Individual components of Six-port
modulator/demodulator i.e. Wilkinson power divider and Branch line coupler are designed and
optimized for 7.5 GHz centre frequency. They are then combined together to make the six-port
correlator and the frequency, phase and amplitude responses were analyzed.
The layout components are generated for both single ended and differential designs and EM
(Electromagnetic) simulation are done using the ADS Momentum tool. The next part is the 8-
PSK signal modulation/demodulation. DSP (Digital Signal Processing) blocks in ADS are used
to produce/recover the baseband signals. A communication system as a whole is simulated and
system level parameters are analyzed.
1.4 Contributions
The work is done by two persons, Muhammad Umar and Umair Yasir. The individual
contributions are mentioned in Table 1.1.
Table 1.1 Individual contributions in work
Muhammad Umar Umair Yasir
Design of Six-port Correlators on
schematic and layout levels
Optimization of the designs for
modulation
Development of mixed analog-DSP 8-
PSK modulation scheme with port-5
signal digital control technique
Demodulation of the 8-PSK signal on
single-ended design
PCB fabrication and measurement
Report writing Ch 2,4,5
Design of Six-port Correlators on
schematic and layout levels
Study of ADS DSP blocks
DSP processing of the signal for mixed
analog-DSP simulations
Study of voltage-controlled switch to
replace the ideal switches in the
modulator
PCB fabrication and measurement
Report writing Ch: 1,3,6,7
5
2 THEORETICAL BACKGROUND
Telecommunication systems are an important constituent of life nowadays. A variety of
telecommunication systems are being used depending on the need of the situation. The simplest
scenario is a one way transmission and reception of information (depicted in Figure 2.1) where a
source wants to send the information to a sink through a channel [13]. To make the information
message appropriate for the channel the sender and detectors are used. The sender interprets the
source’s message to the form appropriate for the channel and detector interprets the message in
the channel back to original form. The task is to design this sender and detector to optimize the
speed, efficiency and cost.
Figure 2.1 A simple telecommunication model
Starting from the simple one way communication scenario the telecommunication technology is
advancing towards more and more complex architectures exploiting the sophisticated signal
processing techniques. The modern communication systems use several additional techniques
including source coding, channel coding interleaving, multiplexing and frequency spreading
[14]. All these efforts are being made to make the sender and detector (in Figure 2.1) more
efficient.
2.1 Modulation schemes
The appropriate utilization of a communication channel requires shift of the information signal
frequency into other frequency band suitable for transmission over the channel. For example a
radio system operates by converting audio signal of 20 Hz – 20 kHz to radio signal of 30 kHz
and upward. This process of shifting the range of the frequency to higher frequencies appropriate
for transmission over the channel is called modulation [15]. Usually modulation is performed by
varying the characteristics of a higher frequency sinusoidal wave according to the modulating
signal (modulating wave). On the receiver side the reverse process of the modulation is
performed known as demodulation.
Modulation types are classified as analog modulation and digital modulation depending on the
type of the modulating signal.
Source Sender Channel Detector Sink
6
2.1.1 Amplitude modulation
In amplitude modulation the amplitude of the carrier wave is varied according to the modulating
wave. In case of digital baseband data the modulating signal is input symbols in form of
distinctive voltage levels. So the modulated output wave also has distinct amplitude levels as
input symbols with the frequency of carrier wave. An amplitude modulated wave can be
expressed as:
(2.1)
SRF is the output modulated signal, Am(t) is the amplitude of the baseband signal and 𝛚c = 2πf is
the angular frequency of the carrier wave.
If the baseband signal is digital then this type of modulation is called Amplitude shift keying
(ASK). If the input signal is an ordinary bit stream with levels of 0 and 1, it will control the
modulated wave like a switch as shown in Figure 2.2. This type of modulation is also called on-
off keying (OOK) [16].
The demodulation of amplitude modulated signal can be done by passing the signal from a low
pass filtering circuit e.g. envelop detector, with removes the high frequency components of
carrier wave and baseband signal is retrieved.
Figure 2.2 Carrier wave, modulating signal and ASK modulated signal
2.1.2 Phase modulation
In phase modulation the phase of the carrier signal is varied according to the modulating wave.
In other words the information is inserted in the phase of the modulated wave. This type of
modulator is implemented using a multiplier.
Carrier wave
Modulating bits
ASK Modulated wave
7
(2.2)
Where AC is the amplitude of carrier and m(t) is modulating signal.
In case of the digital baseband signal where the baseband signal is represented by unique voltage
levels the output modulated wave takes discrete phase shifts. This is called phase shift keying
(PSK). For a single bit stream of 0s and 1s the simplest phase modulation is done by transmitting
the carrier with phase of 0o and 180
o to represent a binary 0 and 1 respectively. This kind of
modulation is called binary phase shift keying (BPSK). Figure 2.3 illustrates the BPSK
modulated wave.
Figure 2.3 Modulating signal, carrier wave, BPSK and BFSK signals
In order to utilize bandwidth more efficiently higher order PSK techniques are employed called
M-PSK. M represents the number of symbols carried by the modulated wave in form of identical
phase shifts. For this purpose the baseband data stream is divided into two or more parallel data
steams which modulate two orthogonal carrier waves which are called in-phase wave and
quadrature-phase wave. These two waves are then added together to get M-PSK modulated
signal. QPSK, 8-PSK and 16-PSK use 4, 8 and 16 phase shifts respectively. Each phase
represents an identical symbol in baseband. Figure 2.4 shows the QPSK and 8-PSK modulation
points in signal space diagram (constellation diagram).
Carrier wave
Modulating bits
PSK Modulated wave
FSK Modulated wave
8
Figure 2.4 (a) QPSK and (b) 8-PSK modulated constellation diagram. Phase difference between any two symbols is 90O
for QPSK and 45
O for 8-PSK
An M-PSK modulated wave is represented by the equation:
[
] (2.3)
where i represents the symbol number.
The data capacity per symbol increases with increased order modulation but on the cost of
increased probability of bit error rate because the Euclidean distance between the two symbols in
signal space decreases.
In digital modulation a combination of ASK and PSK can be used to increase the number of
symbols in signal space, increasing data rate [16]. This type of modulation is called Quadrature
Amplitude Modulation (QAM). In QAM two orthogonal carrier waves are amplitude modulated
and added together to get QAM signal. Figure 2.5 represents constellation diagrams for two
types of QAM symbols. A QAM signal can be represented by the equation:
[ ] (2.4)
Where XI and XQ are the baseband signals respectively called in-phase and quadrature-phase
components of the baseband signal.
I
Q
1
1
-1
-1
45O
I
Q
1
1
-1
-1
90O
(a) (b)
9
Figure 2.5 (a) 4-QAM and (b) 16-QAM modulated constellation diagram
2.1.3 Frequency modulation
Frequency modulation exploits the frequency shifting of the carrier wave according to the
modulating signal. The information to be transmitted is inserted in the frequency of the carrier
wave. Figure 2.3 shows a frequency modulated wave. For digital modulation the discrete
frequency values are used to represent the baseband symbols. It is called frequency shift keying.
The frequency shift between the two values must be as small as possible to save bandwidth. The
minimum shift which can be used is 1/2Tb, (Tb is the bit interval). FSK implemented using this
criterion is called minimum-shift keying or fast-frequency shift keying [16].
2.2 Transceiver architectures
The word transceiver is a combination of “transmitter” and “receiver”. Transceiver is a device
which can behave like transmitter as well as receiver. Efficient transceiver design in wireless
communication is crucial [17]. The design should be capable of supporting high data rates with
minimum errors maintaining communication over a long distance. A transceiver is supposed to
be able to combat channel noises, attenuation and fading.
2.2.1 Transmitter designs
The task of the transmitter is to mix the information signal with a higher frequency carrier to
produce a high power modulated signal in appropriate frequency band. The output power varies
from few mW up to several kW [17]. The generalized output wave equation can be written as:
I
Q
1
1
-1
-1
Q
1
1
-1
-1
(a) (b)
10
(2.5a)
(2.5b)
Usually two types of transmitter architectures are used. One technique is to modulate directly at
the transmission frequency and second is to do modulation at some lower frequency called
intermediate frequency (IF) and then upconvert the signal to some higher frequency for
transmission [6]. The former one is called Homodyne and latter one is called heterodyne
transmitter as shown in Figure 2.6.
Figure 2.6 (a) Homodyne transmitter, (b) Heterodyne transmitter architecture
In Figure 2.6a the VCO (LO) frequency is exactly equal to the carrier frequency; the modulation
and upconversion are directly done at a single stage simultaneously. While in Figure 2.6b a
relatively low frequency (f1) carrier is used for modulation and a second carrier with relatively
I
Q
Baseband
processing
circuit
VCO
sin cos
+ PA
LPF
LPF
BPF &
Matching
network
Self-modulation
LO-leakage
(a)
I
Q
IF VCO
sin cos
LPF
LPF
+
VCO
PA
BPF BPF
BPF &
Matching
networkBaseband
processing
circuit
(b)
11
higher frequency (f2) is used to further upconvert the modulated wave to a higher frequency
(f1+f2). In both type of designs the signal is amplified by a power amplifier (PA).
Quadrature Imbalance
For a homodyne transmitter it is difficult to produce LO signals perfectly at quadrature phase at
high frequency. The inaccuracy in the phase shift and mismatches in I- and Q-paths distorts the
resultant constellation [6]. In heterodyne transmitter the modulation is performed at relatively
low frequency reducing the quadrature errors.
LO leakage and Self Modulation
LO leakage is prominent when there is poor isolation between LO and RF ports of mixer causing
LO signal to escape towards the antenna. In homodyne transmitters the LO leakage signal cannot
be eliminated by the band-pass filter as it is exactly at the signal frequency [6]. While in
heterodyne transmitter LO leakage is not a notable problem. LO leakage causes unnecessary
power dissipation and constellation offset.
Self modulation is caused when the modulated signal is reflected by PA backwards and escapes
from mixer to VCO, disturbing the VCO spectrum. This problem is prominent in heterodyne
transmitters as the VCO is not at the carrier frequency [6].
We can summarize the advantages and disadvantages for both types of transmitter structures as
[18]:
Transmitter
Type Advantages Disadvantages
Homodyne Low cost
High integratibilty
Simple structure
Quadrature Imbalance
LO leakage
Self modulation
Heterodyne
Reliable
performance
No LO leakage
Expensive
Larger in size
Additional filtering
Increased power
dissipation
Table 2.1 Comparison summary of transmitter architectures
12
2.2.2 Receiver designs
The tasks of a RF receiver are to down-convert the signal frequency and demodulate it to get the
original information back. The receiver has to be able to detect a very low-power signal in a
noisy environment in the presence of other unwanted frequencies. Receiver front-end designs are
much complex than transmitter designs due to above mentioned requirements [6]. A simplest
receiver is a tuned radio receiver which contains a BPF, a LNA to amplify the signal power,
demodulator, LPF and a PA.
Depending on the scenarios a variety of receiver architectures are implemented including
homodyne (direct conversion or zero-IF), heterodyne, super-heterodyne and low IF receivers [6].
Figure 2.7 presents the structures for homodyne and heterodyne receivers.
Figure 2.7 Receiver architectures (a) Homodyne (direct-conversion receiver) (b) Heterodyne receiver
Homodyne receiver converts the RF signal to baseband in a single stage as presented in Figure
2.7a. The signal is first filtered then amplified by low-noise amplifier (LNA) to increase its
power to some detectable level. After amplification the signal is down-converted and
demodulated to baseband signal in a single stage. The signal is then low-pass filtered to remove
I
Q
Baseband
processing
circuit
VCO
sin cos
VGA
LPF
LPF
VGA
LNA
BPF
(a)
I
Q
IF VCO
sin cos
LPF
LPF
VCO
VGA
BPFBaseband
processing
circuit
VGA
LNA
Image reject
BPF BPF
(b)
13
high frequency components and variable gain power amplifier (VGA) adjusts the signal power to
appropriate level for analog to digital conversion.
Heterodyne receivers are more popular than homodyne. Heterodyne receivers works in two
stages as illustrated in Figure 2.7b. First the signal is filtered then amplified by LNA, after
amplification it is further filtered to suppress image frequencies. In first stage it is down-
converted to a lower frequency and in second stage it is demodulated to get the baseband signal.
The signal is further filtered and amplified by VGA.
A receiver can be evaluated in terms of its noise figure, sensitivity, selectivity and dynamic range
[17][19]. High performance components and careful design approach must be applied to design a
high fidelity receiver. The main RF receiver design aspects are:
Quadrature Imbalance
The imperfect quadrature phase shift and difference in amplitudes of LO signals cause distorts
the demodulated signal. This problem becomes prominent in homodyne receivers. In addition,
mismatches in I- and Q- signal paths also corrupts the signal constellation. In heterodyne
receivers the demodulation is performed at relatively low frequency reducing the quadrature
imbalance problem.
LO leakage
LO leakage is caused by the poor isolation between LO and RF ports of the mixer. In receivers
the LO signal escapes from the mixer toward the RF port get passed through the parasitic in the
LNA. It may get radiated by the antenna causing unwanted radiations by the receiver or it may
get back to the input of the LNA getting amplified and fed to the mixer. Mixing with original LO
signal it causes self-mixing producing DC components at the output of the mixer. In homodyne
receiver designs this DC component is superimposed on the baseband signals distorting them [6].
Image frequencies
In heterodyne receivers the carrier frequency is first down-converted to an IF frequency. During
mixing the frequencies at a distance of 2fIF from the carrier are also shifted to IF band causing
interference with the original signal. To eliminate image frequency an image rejection filter is
used to filter out the image frequencies prior to mixing. In homodyne receivers image
frequencies are not a problem.
Sensitivity
Sensitivity is defined as minimum signal level required at the antenna to obtain a defined signal
to noise ratio (SNR) at the receiver output [17]. Wireless communication range depends upon the
smallest level of the signal a receiver can process. Homodyne receives exhibit relatively good
sensitivity compared to heterodyne [17].
Selectivity
14
Selectivity is the ability of the receiver to receive a particular band while rejecting the adjacent
bands [17]. In multiband radio communication several radio transmissions are done on adjacent
bands with a guard-band. The band-pass filters in the receivers affect the selectivity most.
Heterodyne receivers have relatively good selectivity than homodyne due to increased filtering
[17].
Noise figure
Noise figure is defined as the ratio of input SNR to output SNR. The noise figure of the whole
receiver system is dominated by the noise figure of first active component. For that reason the
amplifier in the beginning of RF receiver is optimized for the low noise figure hence called Low
noise amplifier (LNA). The receiver sensitivity depends on the receiver’s noise figure.
Comparing homodyne with heterodyne receiver, the advantages and disadvantages can be
summarized as [18][17]:
Receiver Type Advantages Disadvantages
Homodyne
Low cost
High integratable
No image
frequency problem
Reduced filtering
Better sensitivity
Quadrature Imbalance
LO leakage
Low selectivity
Heterodyne
Reliable
performance
No LO leakage
problem
Better selectivity
Expensive and bulky
Additional filtering
Increased power
dissipation
Table 2.2 Comparison summary of receiver architectures
2.3 Differential signaling
Typically the electronic systems share a single conductor for current return path between
transmitter and receiver called ground. This use of a single reference conductor (ground) is
called single ended signaling. The IC package pins have resistance and parasitic causing shifts in
the ground plane. One receiver may be acting as a transmitter for other receivers adding further
shifts in the ground plane as explained in Figure 2.8a. Moreover noise exists between each two
points on the ground conductor. If the reference voltage on the ground conductor is shifted too
15
much the single ended signaling no longer works. The noise produced by unnecessary voltage
drop on the impedance of ground connection on signal return path is called ground bounce [20].
Figure 2.8 Signal transmission systems between two devices with package resistances ZA and ZB (a) Single-ended system with shared ground as current return path (b) two-wire system.
2.3.1 Two wire signaling
Two wire signaling can solve the ground shift problem at cost of one extra wire used as signal
return path instead of common ground as shown in Figure 2.8b. In high frequency circuits the
wires or PCB traces have coupling with the system chassis or other conducting materials. It
causes the current induced in the chassis which can be modeled as added parasitic. The
transmitted signal current finds this parasitic as an option for the return path. The current that
returns from the parasitic is called stray current [20]. In high speed circuits the stray current may
cause malfunctioning of the system.
TxTxRx
GndGnd
Shared Ground
Package connection resistance
Package connection resistance
ZA
ZB
Currentpath
Current to next device
currentreturnpath
(a)
TxTxRx
GndGnd
Shared Ground
Package connection resistance
Package connection resistance
ZA
ZB
Signal wireCurrent to next wire
currentreturnwire
Signal return
(b)
16
The solution of the above mentioned problem is the transmission of the signals mutually opposite
on the both conductor wires (or traces). This type of signaling is called differential signaling. If
both the conductors have the identical coupling with the reference (or the chassis) both wires
induce opposite signals cancelling the effect of each other. If both wires do not have the same
coupling or the signals are not perfectly complementary, some amount of current will flow in the
reference called common mode current.
2.3.2 Voltages and currents in differential signaling
Let denote the instantaneous voltages on the two wires as v1 and v2 with respect to an arbitrary
reference. The difference in the instantaneous voltages v1 and v2 is called differential voltage vd
and the average of the instantaneous voltages is called common-mode voltage vc [20].
(2.6a)
(2.6b)
Common-mode voltage causes production of common mode current. Common-mode current is
not cancelled by noise cancellation property of the differential signaling. Moreover it contributes
a lot in electromagnetic interference (EMI) radiations [21].
Another decomposition of the differential signaling is the even-mode and odd-mode voltages. An
odd-mode voltage in a conductor is one whose opposite exists in the second conductor. Odd
mode voltage vo is half of differential voltage vd. Even-mode voltage is one which is same on
both conductor wires. It is same as common mode voltage vc.
(2.7a)
(2.7b)
Voltage on one conductor is the sum of even- and odd-mode voltages and on the other it is the
difference. Current on one conductor is the sum of differential- and common-mode currents and
on the other it is the difference [20][21].
(2.8a)
17
(2.8b)
(2.8c)
(2.8d)
2.3.3 Differential impedance
Similar to voltage and currents, impedance for differential signaling is also categorized as
differential, common-mode, even and odd impedances. The definitions are as follows [22]:
Differential impedance is the impedance seen into a transmission line when exited in differential
mode.
Common-mode impedance is seen into a transmission line when excited with same signals on
both conductors.
Odd-mode impedance is the impedance of single conductor of transmission line while the other
is excited with opposite signal. Odd-mode impedance is half of differential impedance.
Even-mode impedance is the impedance of single conductor of transmission line while the other
excited with the same signal. Even-mode impedance is the double of common-mode impedance
2.3.4 Mixed-mode S-Parameters
S-parameters or scattering parameters are used to define the response of a microwave network at
RF frequencies. S-parameters for port n are defined by the normalized input and output power
waves (an and bn respectively) when all the ports are terminated in matched conditions.
S-parameters for a single-ended two port network can be defined by a 2×2 matrix. For to define a
two port differential network a 4×4 matrix is required as there is a pair of signals on each port
[23]. The differential signaling contains both differential- and common-mode signaling so single
ended S-parameters cannot be used to fully define the behavior of transmission in differential
signaling. For that purpose mixed-mode s-parameters are used. The mixed-mode s-parameters
for a differential two-port network are defined by [24]:
[
] [
] [
] (2.9)
18
Where adn and acn are the normalized incident power waves for differential- and common-mode
respectively and bdn and bcn are the normalized reflected power waves for differential and
common-mode respectively for the port n. The S-matrix provides information for various
transmission behaviors as explained as follows:
Sdd: differential-mode S-parameters
Scc: common-mode S-parameters
Scd: conversion from common- to differential-mode
Sdc: conversion from differential- to common-mode
With mixed-mode S-parameters the details of wave propagation in differential signaling and
mode conversion are represented. More details on mode conversion can be found in [23].
2.3.5 PCB structures for differential signaling
Different PCB configurations for differential signaling are in use including edge-coupled
differential microstrip, edge-coupled stripline, broadside-coupled stripline and double-sided
parallel stripline (DSPSL) [20][25]. Only double-sided parallel stripline structure is discussed
here. It is like double sided microstrip structure without ground plane with strips on the both
sides of PCB exactly on top and bottom of each other as shown in Figure 2.9c.
Figure 2.9 Cross sections of (a) microstrip and (b) double-sided parallel strip-lines. (a-c) showing conversion of a microstrip line into differential strip line
The design of DSPSL is related to design of simple microstrip line. The characteristic impedance
of a DSPSL of width w on a substrate of height 2h is double the characteristic impedance of a
microstrip line with same width on the same material substrate with height h [26]. It means if we
join two microstrip PCBs back-to-back without ground planes as shown in Figure 2.9, we get a
DSPSL with the characteristic impedance double of microstrip line. If we place a conductor
sheet of infinite size between these two PCBs it will not disturb the field distribution for the
transmission lines but it will convert the DSPSL to two identical microstrip lines with half of the
impedance that of DSPSL [26]. This technique can be used to convert the differential DSPSL
structure to single-ended structure, especially in the case of measuring the differential structure
with single-ended laboratory equipment.
+
_
+
_
(b)(a) (c)
19
3 SIX-PORT CORRELATOR
The six-port correlator was first used as laboratory instrument for measurement of reflection
coefficients and S-parameters of components from 1972-1994 [27] [28] [29]. It was first
demonstrated in 1994 by Ji Li, R.G.Bosisio and Ke Wu that six-port correlator can be used in
radio receiver operated at millimeter-wave frequencies [30]. They used six-port correlator to
demodulate the digitally modulated signal at microwave/millimeter-wave frequencies. Since then
intensive research work is going on designing radio systems employing six-port
modulator/demodulator. Different types of modulation schemes such as 16-QAM, 64-QAM and
QPSK have been successfully implemented. The six-port promises wide bandwidth so its use in
UWB applications has been the focus of the research [31].
3.1 Ideal six-port circuit
Six-port correlator, also known as six-port junction or network is made up of passive microwave
components such as Wilkinson power divider and Quadrature 90o branch line couplers joined
together through transmission lines. Six-port correlator is the fundamental component of the
direct-carrier six-port modulator/demodulator circuit which is an alternative radio transceiver
architecture approach for broadband wireless communication systems.
Based on different combinations and arrangements of WPD (Wilkinson power divider) and
BLCs (branch line couplers), various configurations of six-port correlator are available. One of
the most commonly used configurations consists of one WPD and three BLCs and is shown in
Figure 3.1 [32]. This configuration has been used in this thesis as well.
Figure 3.1 Six-port Correlator--made up of one WPD and three Quadrature BLCs [33]
λ/4
λ/4
50 Ω
Wilkinson Power Divider
QuadratureBranch Line Coupler
Port 1
Port 3
Port 4
Port 5
Port 6
Port 2
20
The individual components i.e. WPD and BLC will now be discussed in detail.
3.1.1 Wilkinson Power Divider
Wilkinson power divider is a three-port network made up of transmission lines and is used to
divide the power of a signal equally in two with introduction of 90o phase shift in both. The S-
parameters matrix of WPD is given below:-
(
√ ) [
] (3.1)
An ideal WPD would exactly divide the input power provided at port 1 into two equal power
outputs at ports 2 and 3. WPD with transmission line lengths is shown in Figure 3.2.
Figure 3.2 Wilkinson Power Divider with corresponding lengths and impedances of transmission lines
If Vn is the incident voltage wave at port n, and Vn
is the reflected wave from same port, then
ideal WPD has the following characteristics:-
if V1 = A cos(t)
V1 = 0 (ideal matching, no reflections)
then V2- = V3
- =
√ (cos(t) – 90
o)
ZO
ZO
ZO
Port 1
Port 2
Port 3
λ/4
λ/4
2ZO
21
3.1.2 Quadrature Branch line coupler
The Quadrature BLC is a four-port network made up of transmission lines and divides the input
power at port 1 into two equal but mutually 90o shifted output powers at ports 2 and 3. Port 4 is
isolated from port 1. The S-parameters matrix is given below:-
√ [
] (3.2)
Quadrature BLC is shown in Figure 3.3.
Figure 3.3 Quadrature Branch Line Coupler with corresponding lengths and impedances of transmission lines
In terms of incident and reflected voltage signals, we can write the characteristics as:-
if V1 = A cos(t) is the input voltage at port 1 and port 4 is terminated with Zo i.e. V4
= 0
V1 = 0, (ideal matching, no reflections)
then V2- =
√ (cos(t) – 90
o)
V3- = -
√ (cos(t) ) ±180
o phase shift
Port 4 is usually terminated with Zo, characteristic impedance of the transmission line. But as can
be seen in the S-parameter matrix, if a signal is applied to this port, it is divided in two and
experiences a phase shift of -90o at port 2 and -180
o at port 3. So in terms of individual voltage
components at ports 2 and 3, resulting from the voltage signals at ports 1 and 4, we have
V2 = V2- =
√ V1 -
√ V4
V3 = V3- =
√ V1 -
√ V4
λ/4
ZO
ZO ZO
ZO
ZO ZO
λ/4
Port 1 Port 2
Port 3Port 4
22
3.1.3 180o Branch Line Coupler
180o
Branch Line Coupler is also called Ring Coupler or Rat-race coupler. It has a different
orientation of transmission lines and their impedances than that of Quadrature BLC. It consists of
ring of transmission line with the total length of six quarter wave-lengths and characteristic
impedance of √ Zo. Four ports are connected with the ring in such a way that three mutual
distances are equal i.e. one quarter wave-length while one mutual distance is three quarter wave-
lengths. Ring coupler is shown in Figure 3.4.
Figure 3.4 Ring Coupler with corresponding lengths and impedances of transmission lines
Based on the selection of input and output ports, it can give two equal outputs of either the same
phase or 180o apart. The S-parameters matrix is given below:-
√ [
] (3.3)
If input is given at port 1, two equal amplitude signals with same phase are at ports 2 and 3,
while port 4 is isolated. If 2 is the input port, we get two equal amplitude but 180o phase apart
signals at ports 1 and 4 while port 3 is isolated. Both configurations are shown in Figure 3.5.
Figure 3.5 Ring coupler with different setting of input port and the corresponding outputs.
/4 3/4
Port1
Port2
Port3
Port4
ZO
Input
Output 0o
Output 0o
Isolated
Input
Output 180o
Output 0o
Isolated
23
3.2 Modulation using Six-port Correlator
For signal modulation using the Six-port correlator (depicted in Figure 3.1) first the S-parameters
of six-port are analyzed. Using WPD and BLC S-parameters matrices, S-parameters of six-port
correlator can be calculated as [6]:
[
]
(3.4)
So S-parameters matrix gives the following general relationships:-
(3.5a)
(3.5b)
(3.5c)
(3.5d)
(3.5e)
(3.5f)
The six-port modulator circuit with respective incident and reflected power waves and variable
impedance terminations at the ports is shown in Figure 3.6.
24
Figure 3.6 Six port Correlator being used a modulator with variable impedance terminations. ai and bi are input and output waves respectively from port i.
The six-port shown above is a common configuration of the Six-port correlator used as a
modulator. Port 1 is used as input port for LO (Local Oscillator) signal and ports 3 to 6 are
terminated with variable impedances controlled by the baseband signals at these ports. Port 2 is
the output port for RF modulated signal.
In Figure 3.6 Assume that ports 1 and 2 are perfectly matched i.e. there is no reflection; port 1 is
the input port for LO signal and port 2 is the output RF signal port. Assuming perfect matching
we get,
b1 = 0, a2 = 0.
Applying the LO signal at port 1,
( )
√ , as
√ (3.6)
and for the rest of ports,
(3.7)
where ai is the incident power wave and bi is the reflected wave at ports i where i = 3,4,5 and 6.
Then applying equations 3.5a - 3.5e, we obtain the output RF signal at port 2
(3.8)
a1
λ/4
λ/4
LO inRF out
b1
Zi1
Zi2
Zi1
Zi2
Zi1
Zi2
Zi1
Zi2
Γ3
Γ4
Γ5
Γ6
i = 3,4,5,6
b2 a2
50 Ω
6
1
2
3
4
5
25
In equation 3.8, the reflection coefficients Г3, Г4, Г5 and Г6 are realized by applying the baseband
signals to the variable impedances at these ports. These variable impedances can be any type of
switch which can take two or more values based on the state of the baseband signal. For higher
order modulation such as 16-QAM or 64-QAM more values or states of the impedances are
required such as 8 for 64-QAM [31]. For lower order modulation such as QPSK, only 2 states
(either short or open) for the impedance termination are enough [34].
In equation (3.8) Г3, Г4 represent the in-phase component (ГI) and Г5, Г6 represent the quadrature
component (ГQ). If for a particular case such as in QPSK we assume Г3 = Г4 and Г5 = Г6 then
equation 3.8 can be written in voltage signal form as
[ ] (3.9)
In time domain, equation 3.9 can be written as [6]:-
√
( (
)) (3.10)
Equation (3.10) describes an RF modulated signal. The amplitude and phase of this signal is
dependent on the variations of the reflection coefficients of the in-phase and quadrature phase
components.
Equation 3.10 can also be written in simplified form as
(3.11)
Where A(t) is the modulated amplitude and (t) is the modulated phase of the RF signal.
3.3 Demodulation using Six-port Correlator
The demodulation of the RF signal is achieved by mixing the received RF signal with the LO
signal which is equal in frequency to the LO signal at the transmitter. Schottky diodes are used
for the mixing purpose. The configuration is shown in Figure 3.7 [35].
26
Figure 3.7 Six-port Demodulator with Diodes, LPFs and Differential Amplifiers
The squared terms after the diode stage are then low pass filtered to reject the higher frequency
components and these low frequency signals are then fed into differential amplifier.
The LO signal is fed into port 1 and RF signal is fed at port 2. We assume perfect matching at
these ports so that
(3.12a)
(3.12b)
(3.12c)
where √
(3.13a)
and
(3.13b)
(3.14a)
(3.14b)
b2
a1
LO in
b1
1
LPF
LPF
LPF
LPF
+
+
I
Q
bi ai =Γibi
6
3
4
5
i = 3,4,5,6
(o)2
(o)2
(o)2
(o)2
50 Ω
RF in
a2
2
Six-portCorrelator
27
Now according to the S-parameters matrix in (3.4), we deduce the following signals at the ports
3, 4, 5 and 6
[ ( )] (3.15a)
[ ( )] (3.15b)
[ ( )] (3.15c)
[ ( )] (3.15d)
After squaring of the these signals by the diodes, they are passed to LPF (Low Pass Filter) to
reject the higher order harmonics such as 2LO(t), 2c(t) and LO(t)+ c(t). We have for V3 [6]:
(3.16a)
Similarly at other ports,
(3.16b)
(3.16c)
(3.16d)
Now to be able to recover the baseband signal from the resulted signals above, the first two
unwanted component terms are to be eliminated from each signal. In order to achieve this,
differential amplifier is used. Secondly the LO signal at the receiver end should be exactly equal
to the LO signal at the transmitter i.e. the carrier signal such that
ct - LOt = 0
Differential amplifiers give the output baseband signals i.e.
28
( ) (3.17a)
( ) (3.17b)
In this way both the in-phase and quadrature components of the baseband signal can be
recovered.
29
4 Design and implementation of six-port correlator
The task is to design single-ended and differential six-port correlator for UWB with centre
frequency of 7.5 GHz with minimum amplitude and phase imbalance for maximum bandwidth.
Various structures of six-port correlators are designed, simulated and fabricated. First, the
components of six-port are designed as standalone structures. These designs are simulated to
evaluate the performance, then optimized and integrated to make the six-port correlator. The six-
port structures are then optimized and fabricated.
Advance Design System (ADS) from Agilent Inc. is used for circuit simulations and evaluations.
Rogers 4350B substrate is used for PCB fabrication of the designs. Substrate specifications are
mentioned in Table 4.1. Single-ended designs are fabricated for 50 Ω and differential designs for
100 Ω port impedances. 50 Ω SMA female connectors for ports are used in PCB fabrication.
To measure the differential designs with available single-ended vector network analyzer (VNA)
a single-ended to differential conversion mechanism is used and a wideband 180O coupler is
designed and fabricated to be used as the converter.
The design and evaluation process is divided into three parts. First, the six-port designs are
simulated in ADS on schematic level to have a look on the ideal results. Second, the schematic
structure is converted to layout structure and simulated using ADS momentum field-solver. In
third step the layout design is modified to be suitable for fabrication and then fabricated for
practical evaluation.
Table 4.1 Rogers4350B substrate specifications
Relative dielectric constant 3.66
Substrate thickness 254 µm
Conductor thickness 35 µm
Metal conductivity 5.8×10
7
S/m
Loss tangent 0.004
Surface roughness 0.001 mm
30
4.1 Single-ended designs Initially single-ended designs are simulated, fabricated and analyzed for the performance. The
designs are optimized for 50 Ω port impedances and for best performance on 7.5 GHz. Two type
of single-ended structures are designed: i) classical design ii) design with matching stubs.
4.1.1 Classical single-ended design
This six-port design has been frequently used in the research [36][37]. It contains a Wilkinson
divider with three quadrature couplers. This is a simple and compact design. The layout of the
design is shown in Figure 4.1. The design dimensions are 20 mm × 25 mm.
Figure 4.1 Layout design of single-ended classical six-port correlator
Port 1 is referred as LO (local oscillator) and port 2 as RF (radio frequency) port. Port 3 and 4
are used as in-phase and port 5 and 6 are used as quadrature-phase ports. Port 7 is not used and
terminated by 50 Ω termination.
From fabrication point this design could not be fabrication with available equipment because of
smaller size. To make the design feasible for fabrication, extra transmission-line lengths are
added to the interconnects, increasing the design size to 31 mm × 34 mm. The electrical length of
added transmission-line on each port is λ/4 (6 mm). The modified design layout is shown in
Figure 4.2 and fabricated design in Figure 4.3.
25
mm
20 mm
P3 P4
P6 P5
P1(LO)
P2(RF)
P7
31
Figure 4.2 Layout design of single-ended classical six-port correlator modified for fabrication. The design dimensions are 31 mm × 34 mm
Figure 4.3 Manufactured Single-ended classical six-port correlator prototype
The simulated and measured S-parameter results for this design are shown in Figure 4.4 to 4.11.
34
mm
31 mm
P3 P4
P6 P5
P1(LO)
P2(RF)
P7
P1
P3 P4
50 ΩTermination
P2
P5P6
32
Figure 4.4 Measured and simulated input reflection coefficients
Figure 4.5 Measured and simulated isolations between Port 1 and Port 2
Figure 4.6 Measured and simulated S-parameters for transmission from Port 1 to Port 3 and 4
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (GHz)
Inp
ut R
efle
ctio
n (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S11
Measured S22
Simulated S11
Simulated S22
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (GHz)
Iso
latio
n b
etw
ee
n P
1 a
nd
P2
(d
B)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated
Measured
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-3
-15
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S31
Measured S41
Simulated S31
Simulated S41
33
Figure 4.7 Measured and simulated S-parameters for transmission from Port 1 to Port 5 and 6
Figure 4.8 Measured and simulated S-parameters for transmission from Port 2 to Port 3 and 4
Figure 4.9 Measured and simulated S-parameters for transmission from Port 2 to Port 5 and 6
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-3
-15
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S61
Measured S51
Simulated S61
Simulated S51
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-3
-15
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S32
Measured S42
Simulated S32
Simulated S42
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-3
-15
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S62
Measured S52
Simulated S62
Simulated S52
34
Figure 4.10 Simulated phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 or Port 2 is used as input port
Figure 4.11 Measured phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 or Port 2 is used as
input port
Figure 4.4 shows the input reflection coefficients for Port 1 and 2. The simulated results shows
the input reflection less than -20 dB at the centre frequency but measured results are not in good
agreement with the simulated results. Measured results have input reflections coefficients higher
than -10 dB for the centre frequency. It can also be noticed in Figure 4.4 the isolation between
Port 1 and Port 2 is in simulated results is -50 dB for the centre frequency and for the extreme
sides of selected band it goes to approximately -9dB but again the measured results are not that
good as simulated. The best isolation is measured to be -22 dB on 8.5 GHz.
6.5 7.0 7.5 8.0 8.56.0 9.0
0
100
200
-100
300
Frequency (GHz)
Sim
ula
ted
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S61 – S51
6.5 7.0 7.5 8.0 8.56.0 9.0
-300
-200
-100
0
100
200
300
-400
400
Frequency (GHz)
Measure
d P
hase D
iffere
nce (
Degre
e)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
35
The forward transmission response is depicted in Figure 4.5 and 4.6. Simulated results shows the
values around -6 dB for transmission to all ports. The measured results are deviated from the
simulated results specially the transmission to Port 4 and 5 which have the transmission loss of
approximately -9 dB at centre frequency . Transmission to Port 3 and 6 have relatively better
measured response and are close to simulated values. A same behavior is visible in Figure 4.7
and 4.8 for transmission to same ports but now Port 2 as input port. The simulated results hover
around -6 dB for centre frequency (7.5 GHz) and the measured results have more loss and float
around -9 dB at centre frequency.
Figure 4.9 and 4.10 shows the phase response of simulated and measured values respectively.
The phase difference between Port 3 and 4, and Port 6 and 5 for transmission when Port 1 or 2 is
used as input port is close to 90O for both simulated and measured results at centre frequency.
The phase difference deviate smoothly as frequency is changed from the centre frequency.
The measured results have deviations from the simulated because of bad etching, substrate
errors, SMA connectors and soldering. The main problem encountered in this project is uneven
etching of copper on PCB changing the width of transmission lines. The transmission line with
different widths exhibits different impedance causing high input reflection coefficients as in this
case.
4.1.2 Single-ended design with matching stubs
The classical design presented in previous topic has compact size but exhibits a narrow band
response. The Wilkinson divider has a relatively uniform response on a larger bandwidth than
the quadrature couplers. To make the quadrature couplers wideband, matching networks with
open-circuited stubs are applied on the ports. The idea is to exploit the fact that coupling depends
upon the admittance of the ports [38]. The matching network is presented in Figure 4.11.
Figure 4.11 Matching network applied on the coupler ports to broadband the response
The optimized values for Zstub and Zline are 50 Ω and 80 Ω respectively. The open-circuited stub
is folded inward to save the space on PCB.
λ/2
λ/2
Zstub
Zline
Zo
36
To increase the bandwidth of the six-port correlator, the simple quadrature couplers are replaced
with these optimized quadrature couplers with matching networks. The layout design is shown in
Figure 4.12. The design dimensions are 52 mm × 47 mm. The port assignment is same as
specified for the previous design.
Figure 4.12 Layout design for single-ended six-port correlator with matching networks
The simulated results for this six-port correlator design are shown in Figure 4.12 to 4.15
Figure 4.12 Simulated input reflection coefficients on Port 1 and 2
47
mm
52 mm
P3
P4
P6
P5
P1(LO) P2
(RF)
P7
6.3 6.6 6.9 7.2 7.5 7.8 8.1 8.4 8.76.0 9.0
-30
-20
-10
-40
0
Frequency (GHz)
Inp
ut R
efle
ctio
n (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated S11
Simulated S22
37
Figure 4.13 Simulated S-parameters for transmission from Port 1 to Port 3, 4, 5 and 6
Figure 4.14 Simulated S-parameters for transmission from Port 2 to Port 3, 4, 5 and 6
Figure 4.16 Simulated phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 or Port 2 is used as
input port
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-15
-3
Frequency (GHz)
Forw
ard
tra
nsm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated S31
Simulated S41
Simulated S51
Simulated S61
6.5 7.0 7.5 8.0 8.56.0 9.0
-12
-9
-6
-15
-3
Frequency (GHz)
Forw
ard
tra
nsm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated S32
Simulated S42
Simulated S52
Simulated S62
6.3 6.6 6.9 7.2 7.5 7.8 8.1 8.4 8.76.0 9.0
-300
-200
-100
0
100
200
300
-400
400
Frequency (GHz)
Sim
ula
ted
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
38
Figure 4.12 presents the input reflection coefficients for Port 1 and 2. The values at centre
frequency are less than -20 dB and for the worst condition it is less than -12 dB. The response for
forward transmission to Port 3, 4, 5 and 6, when Port 1 or Port 2 is used as input port are shown
in Figure 4.13 and 4.14 respectively. The both figures show a flat response around -7dB. The
design exhibits a wideband response from 6 – 9 GHz.
The phase response is depicted in Figure 4.15. The phase difference between Port 3 and 4, and
between Port 5 and 6 is close to 90O on 6 – 9 GHz frequency band.
4.2 Differential designs
The differential designs are optimized for 100 Ω port impedance and for centre frequency of 7.5
GHz. For differential designs, the same substrate (Rogers4350B) with double thickness is used.
For manufacturing the prototypes, two single-sided PCBs are fabricated and then joined back-to-
back to make differential designs.
Two type of differential six-port correlator structures are designed.
i) Classical differential design
ii) Wideband differential design
4.2.1 Classical differential design
This design is exactly same to the classical single-ended one but it is implemented on DSPSL
(double-sided parallel strip-line) instead of microstrip structure, i.e. the ground plane is removed
and same pattern of transmission lines is printed on both sides of PCB. This design has been used
in research [8]. The design dimensions are 20 mm × 25 mm, same to its single-ended counter-
part. The 3D view of layout design is shown in Figure 4.17.
The design is then modified according to fabrication and measurement requirements. The
available measuring equipment is single-ended vector network analyzer (VNA). To make the
differential design compatible with single-ended VNA, the differential ports are converted to
single ended by inserting the ground-plane on the boundaries of the design. By inserting the
ground plane between DSPSL, a single 100 Ω differential port is converted to two single-ended
50 Ω ports. An electrical length of λ/2 (12 mm) is added on each port to make is feasible for
fabrication. The modified layout design and manufactured prototype are shown in Figure 4.18
and 4.19 respectively. The design dimensions after modifications are 35 mm × 48.5 mm.
To measure the design a single-ended wideband 180O coupler is designed (discussed in section
4.3). The purpose of this coupler is to divide the single-ended signal from VNA to two
complementary signals; these two single-ended complementary signals are used to feed the
differential design.
39
Figure 4.17 3D view of layout design for classical differential six-port correlator. The front view is exactly same as its single-ended counter-part
Figure 4.18 Layout design of classical differential six-port correlator modified for fabrication. The design dimensions are 35 mm × 48.5 mm
P3
P4
P6
P5
P1(LO)
P2(RF)
P7
Top layer
Bottomlayer
35 mm
48
.5 m
m
Top+Bottom layer(Differential signal here)
Top layer (SE signal here)
Bottom layer (SE signal here)
Ground layerbetween two PCBsjoined back to back
P1+
(LO +)
P1
(LO )
P3+
P3
P4+
P4
P5+
P5
P6+
P6
P7
40
(a) (b)
Figure 4.19 Manufactured classical differential six-port correlator prototype. (a) Top-side (b) Bottom-side
The simulated and measured S-parameter results for this design are shown in Figure 4.20 to 4.27.
Figure 4.20 Measured and simulated input reflection coefficient for Port 1 and 2
P1+
P3+
P3-
P1-
P4+
P4-
P5-
P5+
P6-
P6+
P6+
P6-
P6+
P6-
P3+
P3-P4+
P4-
P5-
P5+
P6-
P6+
P1+
P1-
(a) (b)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
0
-60
10
Frequency (GHz)
Inp
ut R
efle
ctio
n (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S11
Measured S22
Simulated S11
Simulated S22
41
Figure 4.21 Measured and simulated isolation between Port 1 and 2
Figure 4.22 Measured and simulated S-parameters response for transmission from Port 1 to Port 3 and 4
Figure 4.23 Measured and simulated S-parameters response for transmission from Port 1 to Port 5 and 6
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-60
-10
Frequency (GHz)
Iso
latio
n b
etw
ee
n P
1 a
nd
P2
(d
B)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated
Measured
6.5 7.0 7.5 8.0 8.56.0 9.0
-24
-18
-12
-6
-30
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S31
Measured S41
Simulated S31
Simulated S41
6.5 7.0 7.5 8.0 8.56.0 9.0
-24
-18
-12
-6
-30
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S61
Measured S51
Simulated S61
Simulated S51
42
Figure 4.24 Measured and simulated S-parameters response for transmission from Port 2 to Port 3 and 4
Figure 4.25 Measured and simulated S-parameters response for transmission from Port 2 to Port 5 and 6
Figure 4.26 Simulated phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 and Port 2 is used as
input port
6.5 7.0 7.5 8.0 8.56.0 9.0
-24
-18
-12
-6
-30
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S32
Measured S42
Simulated S32
Simulated S42
6.5 7.0 7.5 8.0 8.56.0 9.0
-24
-18
-12
-6
-30
0
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S62
Measured S52
Simulated S62
Simulated S52
6.5 7.0 7.5 8.0 8.56.0 9.0
0
100
200
-100
300
Frequency (GHz)
Sim
ula
ted
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
43
Figure 4.27 Measured phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 and Port 2 is used as input port
Figure 4.20 shows the input reflection coefficients on Port 1 and 2 for the simulated and
measured values. The values of input reflection coefficients for simulated response are less than -
20 dB for the centre frequency and less than -10 dB for the whole 6 – 9 GHz frequency band.
The measured response has the undesired values above -10 dB for the centre frequency showing
improper input impedance. At 8 GHz the measured values have complete mismatch. Also the
simulated response for isolation between Port 1 and 2 (Figure 4.21) have better isolation of more
than 50 dB on centre frequency compared to measured response with 27 dB isolation on centre
frequency. The best isolation for measured response is 44 dB at 6.1 GHz.
The forward transmission response from Port 1 to Port 3, 4, 5 and 6 is presented in Figure 4.22
and 4.23. Simulated design has loss close to -6 dB for transmission to all ports on the centre
frequency and values change smoothly as the frequency changes. The measured structure does
not have good response for forward transmission. At 8 GHz a huge insertion loss is due to large
input mismatching on this frequency.
Figure 4.24 and 4.25 shows the measured and simulated behaviors for transmission from Port 2
to Port 3, 4, 5 and 6. The simulated values shows -6 dB loss at the centre frequency and the loss
increases rapidly but smoothly as move on the frequency axis. The measured response is not
following any trend and changing rapidly. The measured values have high insertion loss. The
highest insertion loss occurs on 8.0 GHz due to input mismatching.
Phase response (Figure 4.26) for simulated values has 90O phase difference between transmission
to Port 3 and 4, and between Port 5 and 6. The Measured phase response (Figure 4.27) has the
phase difference between Port 3 and 4, and between Port 5 and 6 floating around 180O.
Some reasons of measured results deviations from the simulated results are discussed in single-
ended design section. The additional non-idealities in differential design are that the differential
design is produced by joining two separate PCBs back-to-back with glue. The glue and the air
6.5 7.0 7.5 8.0 8.56.0 9.0
-300
-200
-100
0
100
200
300
-400
400
Frequency (GHz)
Me
asu
red
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
44
trapped inside changes the effective dielectric constant of the substrate. While joining the PCBs,
one PCB should be exactly collocated on top of other. The human error in joining the PCBs is
also there. In addition, the coupler used for single-ended to differential conversion is not ideally
splitting the power in half with ideal 180O phase. All these problems tend to change the measured
results from the simulated results.
4.2.2 Wideband differential design
It is a novel six-port design using rat-race couplers (180O hybrid couplers) instead of quadrature
couplers. To make the six-port correlator wideband, the rat-race couplers are designed with flat
amplitude response on a wide frequency range.
A typical rat-race coupler contains six λ/4 (quadrature phase) transmission lines in its main ring
as shown in Figure 4.28a. To miniaturize and broadband the differential design two of the three
quadrature-phase transmission lines between port 2 and 4 are replaced by DSPSL phase inverter
(crossing the conductors of both layers using VIAs) [39]. Two quadrature-phase transmission
lines perform 180O phase shift and a same effect is performed by DSPSL phase inverter. Figure
4.10b shows the differential wideband rat-race coupler with layers’ crossing for phase inversion
and Figure 4.10c presents the structure of DSPSL phase inverter. VIA holes with 0.15 mm
diameter are used in DSPSL phase inverter.
Figure 4.28 (a) A typical rat-race coupler (b) Wideband miniaturized differential rat-race coupler with phase inverter (c) 3D view of a DSPSL phase-inverter
The wideband rat-race couplers are used with Wilkinson divider to get a wideband six-port
correlator depicted in Figure 4.29. Port 1 and 4 of designed couplers are joined together and port
2 and 3 are used for external interface. Unlike quadrature coupler the rat-race coupler combines
and divides power with 0O or 180
O phase difference. In six-port correlator we require 90
O phase
difference in division or combination by each coupler. Extra transmission lines are added in this
design to adjust the phase difference. The overall design dimensions are 33 mm × 23 mm.
/4
/4
/4
/8
/8
/4
/4
/4
3/4
Top layer
Bottomlayer
(b)(a) (c)
P3
P4
P1
P2
P1 P2
P3 P4
45
Figure 4.29 Layout design for Wideband differential six-port correlator. The total design dimensions are 33 mm × 23 mm
The design is then modified to be suitable for fabrication and measurement with single-ended
VNA. Same technique of converting differential port to two single-ended ports as described in
previous section is used Transmission-lines with electrical length of λ/2 are added on each port
and diameter of VIA hole is increased to 0.4 mm to make the design feasible for fabrication.
Figure 4.30 and 4.31 shows the modified layout and manufactured prototype respectively.
The simulated and measured S-parameter results for this design are shown in Figure 4.32 to 4.39.
23
mm
33 mm
P3
P4
P6
P5
P1 (LO)
P2 (RF)
P7
46
Figure 4.30 Layout design for wideband differential six-port correlator modified for fabrication
Figure 4.31 Manufactured prototype of wideband differential six-port correlator
60
mm
61 mm
P3+
P1+ (LO +) P1 (LO )
P3 P6+P6
P4+
P4
P2+ (RF +) P2 (RF )
P5+
P5
P7
P1+
P1-
P3+
P6+
P2+
P2-
P4+
P4-
P5+
P5-
P1+
P1-
P3-
P6-
P4+
P4-
P5+
P5-
P2+
P2-
(a) (b)
47
Figure 4.32 Measured and simulated input reflection coefficient for Port 1 and 2
Figure 4.33 Measured and simulated isolation between Port 1 and Port 2
Figure 4.34 Measured and simulated S-parameters response for transmission from Port 1 to Port 3 and 4
6.5 7.0 7.5 8.0 8.56.0 9.0
-25
-20
-15
-10
-5
0
-30
5
Frequency (GHz)
Input re
flectio
n (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S11
Measured S22
Simulated S11
Simulated S22
6.5 7.0 7.5 8.0 8.56.0 9.0
-40
-35
-30
-25
-45
-20
Frequency (GHz)
Iso
latio
n b
etw
ee
n P
1 a
nd
P2
(d
B)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Simulated
Measured
6.5 7.0 7.5 8.0 8.56.0 9.0
-25
-20
-15
-10
-30
-5
Frequency (GHz)
Forw
ard
Tra
nsm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S31
Measured S41
Simulated S31
Simulated S41
48
Figure 4.35 Measured and simulated S-parameters response for transmission from Port 1 to Port 5 and 6
Figure 4.36 Measured and simulated S-parameters response for transmission from Port 2 to Port 3 and 4
Figure 4.37 Measured and simulated S-parameters response for transmission from Port 2 to Port 5 and 6
6.5 7.0 7.5 8.0 8.56.0 9.0
-25
-20
-15
-10
-30
-5
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
1 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S61
Measured S51
Simulated S61
Simulated S51
6.5 7.0 7.5 8.0 8.56.0 9.0
-25
-20
-15
-10
-30
-5
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S32
Measured S42
Simulated S32
Simulated S42
6.5 7.0 7.5 8.0 8.56.0 9.0
-25
-20
-15
-10
-30
-5
Frequency (GHz)
Fo
rwa
rd T
ran
sm
issio
n fro
m P
ort
2 (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured S62
Measured S52
Simulated S62
Simulated S52
49
Figure 4.38 Simulated phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 and Port 2 is used as
input port
Figure 4.39 Measured phase difference between Port 4 and 3, and between Port 5 and 6, when Port 1 and Port 2 is used as
input port
Figure 4.32 shows the input reflection coefficients for Port 1 and 2. The simulated values have
input reflection for Port 1 less than -10 dB and for Port 2 less than -20 dB for the whole
frequency band of 6 – 9 GHz. The measured results show high reflection at input ports. The
values deviate rapidly with frequency. At 8.5 GHz the reflection coefficient is approximately 0
dB showing complete mismatching. The isolation between Port 1 and Port 2 (depicted in Figure
4.33) is more than -20 dB for the simulated design on whole 6 – 9 GHz band. The measured
values show more isolation than simulated values.
6.5 7.0 7.5 8.0 8.56.0 9.0
-200
-100
0
100
200
-300
300
Frequency (GHz)
Sim
ula
ted
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
6.5 7.0 7.5 8.0 8.56.0 9.0
-300
-200
-100
0
100
200
300
-400
400
Frequency (GHz)
Measure
d P
hase D
iffere
nce (
Degre
e)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S31 – S41
S61 – S51
S32 – S42
S62 – S52
50
Forward transmission from Port 1 to Port 3, 4, 5 and 6 is depicted in Figure 4.34 and 4.35. The
simulated values for transmission to Port 3 and 4 shows values of -6dB and transmission to Port
5 and 6 shows values of approximately -6.5 dB for whole 6 – 9 GHz frequency band. A same
behavior is visible in Figure 4.36 and 4.37 for simulated values showing a flat response around -
6.5 dB. The simulated results show the designed six-port as a wideband device. The measured
results for forward transmission are not in agreement with simulated results. The forward
transmissions show a loss of more than -10 dB for whole frequency band in all cases (Figure
4.34 – 4.37) with the response rapidly changing with the frequency. At 8.5 GHz the forward
transmission has the worst response in all cases with insertion loss more than -20 dB as the
reflection coefficient at this frequency is approximately 0 dB.
Phase response (Figure 4.38) for simulated values has flat 90O phase difference (on whole
frequency band) between transmission to Port 3 and 4, and between Port 5 and 6. The Measured
phase response (Figure 4.27) has the phase difference between Port 3 and 4, and between Port 5
and 6 floating around 180O.
The reasons of measured result deviations are explained in section 4.1.1 and 4.2.1. In addition to
all these problems this design uses PCB VIAs to connect the transmission line on one side of
PCB to other side. These VIAs were implemented by drilling the PCB and soldering a small wire
on both sides. These solder joints come very close to each other introducing an undesired
capacitance.
4.3 Wideband 180O coupler design
To measure the manufactured prototypes of differential designs with available single-ended
vector network analyzer, each 100 Ω differential port of the six-port correlator is split into two
50 Ω single-ended ports. For conversion, ground plane is inserted between the DSPSLs on the
design boundaries. These two single-ended ports represent one differential port and carry single-
ended signals equal in magnitude but out of phase with 180O.
To connect the pair of 50 Ω ports (carrying complementary signals) to one 50 Ω VNA port a
wideband 180O coupler is designed which can split the one single-ended signal from VNA to two
signals equal in magnitude and out of phase by 180O and the same operation in reverse. A simple
rat-race coupler can be used for this purpose but rat-race coupler has a very limited bandwidth.
A Wilkinson divider is a broadband device in terms of equal power splitting. For equal phase
shift on large frequency-band a broadband phase-shifter using loaded transmission lines is
designed [40]. To achieve a high bandwidth with 3db coupling and constant 180O phase shift the
Wilkinson divider in combination with the broadband phase-shifter is used. Figure 4.40 and 4.41
shows the layout design and manufactured prototype respectively.
51
Figure 4.40 Layout design of wideband 180O
coupler
Figure 4.41 Manufactured prototype of wideband 180s coupler
The simulated and measured S-parameter results for this design are shown in Figure 4.42 to 4.44.
22 m
m
33 mm
P1(Input)
P2(Output)
P3(Output)
P1P3
P2
52
Figure 4.42 Input reflection coefficients for Port 1 to 3
Figure 4.43 S-parameters for transmission between the Ports
Figure 4.44 Phase difference between the output signals
6.5 7.0 7.5 8.0 8.56.0 9.0
-40
-30
-20
-10
-50
0
Frequency (GHz)
Inp
ut
refle
ctio
ns (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S11 Measured
S33 Measured
S22 Measured
S11 Sim
S33 Sim
S22 Sim
6.5 7.0 7.5 8.0 8.56.0 9.0
-30
-25
-20
-15
-10
-5
-35
0
Frequency (GHz)
Tra
nsm
issio
n c
oe
ffic
ien
ts (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
S21 Measured
S23 Measured
S31 Measured
S21 Sim
S23 Sim
S31 Sim
6.5 7.0 7.5 8.0 8.56.0 9.0
-100
0
100
200
-200
300
Frequency (GHz)
Ph
ase
Diffe
ren
ce
(D
eg
ree
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
Measured
Simulated
53
The input reflection coefficients (Figure 4.42) for simulated values shows input reflection of less
than -10 dB on Port 1 and less than -20 dB on Port 2 and 3. The measured values show a higher
reflection on all ports. The worst reflection occurs at 8.2 GHz with -3dB of reflection.
The forward transmission from Port 1 to 2 and 3 are shown in Figure 4.43. The simulated results
shows a wideband response from 6 – 9 GHz. Measured results have deviation from the simulated
results. The maximum deviation occurs at 8.2 GHz. Both simulated and measured results show
more than 15 dB of isolation between Port 2 and 3.
Figure 4.44 shows the simulated and measured phase difference between the outputs on Port 2
and 3 when Port 1 is used as input port. The simulated values show a maximum of 5O phase error
on 7 GHz. The measured response has 28O of maximum error on 6 GHz. At the centre frequency
(7.5 GHz) simulated results have 4O and measured results have 12
O of phase error.
The reasons for differences between the simulated and measured results are already discussed as
bad etching, substrate errors, SMA connector losses, soldering and bad grounding. Here the
uneven etching can be easily noticed in Figure 4.41. An example is the transmission line
connecting Wilkinson divider to Port 2. The transmission line gets thicker as it goes from left to
right.
54
5 Six-Port modulator and demodulator design
A six-port correlator can be used as a modulator or demodulator. Its theory and mathematics is
explained in detail in chapter 3. In this chapter the implementation of modulator and
demodulator using the designed six-port correlators is presented. The simulated electromagnetic
models of the designs are used to implement the modulator and demodulator.
The modulation scheme used is 8-PSK and a mixed analog-DSP designing is used. The baseband
signal processing is done by DSP algorithms. The digital baseband signal is then converted to
analog signal and modulated, transmitted and demodulated using six-port correlators.
5.1 Six-port modulator implementation
A six-port modulator utilizes the reflections of LO signals from port 3-6 by changing reflection
coefficients (Γ3 to Γ6) to modulate the signal. For this purpose variable impedance loads are
used which are controlled by the baseband signal to be transmitted. To check the performance of
the designs, initially 16-QAM modulation is implemented in analog environment because 16-
QAM can be easily implemented in analog environment unlike 8-PSK which need some DSP
processing. 16-QAM can be implemented by switching Γ3 and Γ6 between +1/-1 and Γ4 and Γ5
between +0.5/-0.5 [41]. Figure 5.1 depicts a simple analog six-port 16-QAM modulator.
Figure 5.1 A six-port QAM modulator. Z1 and Z2 are selected to get reflection coefficients of +0.5 and -0.5
LO RF out
Z1Z2
Z1
Z2
55
The designed six-port correlators are then optimized for modulation by minimizing the I-Q
imbalances and carrier leakage.
5.1.1 Variable port impedances
The reflection coefficients on the port 3 to 6 are altered (controlled) by the baseband data for
modulation. To obtain variable reflection coefficients, voltage controlled impedances are
required. Different design approaches have been used in research including hetero-junction FET
as variable impedance [33] and Schottky Diode as High-Speed Variable Impedance [42].
In this project an attempt was made to use the PIN diode as RF-switch or variable voltage-
controlled impedance. A PIN diode can behave like voltage-controlled impedance and PIN
diodes are also widely used as RF-switches [43]. The impedance of PIN diode decrease as
applied voltage increases, this phenomenon was exploited to switch the PIN-diode impedance
between two values generating two reflection coefficient values. Various circuits with different
models of PIN diode were simulated and some noticeable results were obtained but an optimized
design with accurate results could not be achieved because of limited time.
The task of PIN diode based variable impedance is left for future work and ideal switch model in
ADS is used as voltage controlled variable impedance, depicted in Figure 5.2. Each out of four
ports of six-port (Port 3 to 6) is terminated by an ideal switch. The switch is controlled by the
baseband input bits and it terminates the port with two different impedances. When the switch is
in open state, input impedance is Z1, and when the switch is closed, Z2 comes in parallel with Z1.
Figure 5.2 Ideal switch in ADS used as voltage controlled variable impedance. Z1 and Z2 are adjusted to get two desired input impedance values
5.1.2 8-PSK modulation using mixed analog-DSP designing
8-PSK modulation uses three baseband data bits to encode one symbol. Three baseband data-
sources are required for 8-PSK modulation or a single data-source can be divided into three
parallel bit-streams with 1/3 data rate as of original source.
56
The proposed encoding scheme uses two data-sources to map the symbol in I-axis and third data-
source to map the symbol in Q-axis. Switches on Port 3 and 6 are toggled between reflection
coefficients of +1/-1 and switches on Port 4 and 5 are toggled between reflection coefficients of
+0.414/-0.414. Two baseband sources are used to operate switches of Port 3 and 4 (toggling Γ3
& Γ4) mapping in-phase data symbol in the signal space (shown in Figure 5.3a). The third
baseband source operates the switch on Port 6 (toggling Γ6) and shifts the mapped symbol along
Q-axis (shown in Figure 5.3b), upward or downward depending on the value of bit.
Figure 5.3 Stepwise constellation diagrams for 8-PSK proposed scheme. (a) Symbols are mapped on I-axis using data-bits from two baseband source (b) Third baseband data source is used to move the symbol in Q-axis (c) Symbols without
proper 8-PSK positions (d) Symbols moved to proper positions using baseband processing
Port 5 is used to further move the symbol along Q-axis to give the constellation proper shape
(shown in Figure 5.3c-d). If the mapped symbol on I-axis is on the edges, port 5 moves it
vertically inwards by generating signal opposite to the Port 6. If the mapped symbol on I-axis
was not on the edge, Port 5 generates same signal as on port 6 supporting it to take the symbol
further away from the origin on Q-axis.
To decide about Port 5 first Port 3 and Port 4 data bits are compared, if the bits are same then
inverted bit of Port 6 is fed to Port 5 and viceversa. DSP baseband processing is used to conduct
this operation. A simple combination of XOR and XNOR gate (shown in Figure 5.4) can do this
job. Table 5.1 shows the possible values for Port 5.
57
Table 5.1 Possible combinations of data for Port 5
Port
3
Port
4
Port
6
Port
5
0 0 0 1
0 0 1 0
1 1 0 1
1 1 1 0
0 1 0 0
0 1 1 1
1 0 0 0
1 0 1 1
The complete 8-PSK modulator based on analog-digital mixed design is shown in Figure 5.4
Figure 5.4 Six-port 8-PSK modulator based on analog-DSP mixed designing
58
Three bit-sources generates baseband data. The data is processed by the combination of XOR
and XNOR gates for Port 5. The digital signal is converted to analog and applied to analog part
of six-port modulator to control the port switches.
The resultant constellation diagrams and signal spectrums are presented in Figure 5.5
(a)
(b)
6.4 6.6 6.8 7.0 7.2 7.4 7.6 7.8 8.0 8.2 8.4 8.66.2 8.8
-100
-80
-60
-40
-120
-20
Frequency (GHz)
Mo
du
late
d S
ign
al P
ow
er
(dB
m)
6.4 6.6 6.8 7.0 7.2 7.4 7.6 7.8 8.0 8.2 8.4 8.66.2 8.8
-90
-80
-70
-60
-50
-40
-100
-30
Frequency (GHz)
Mo
du
late
d S
ign
al P
ow
er
(dB
m)
59
(c)
Figure 5.5 Constellation diagrams and spectrum of 8-PSK modulated signal with data rate of 500 Mbps and modulator LO power of 0 dBm for (a) Single-ended design (b) Differential design (c) Wideband Differential design with crossed
conductors
5.2 Six-port demodulator implementation
The six-port demodulator is implemented according to the demodulation process explained in
Chapter 3. The LO signal is applied on Port 1 and received RF signal is applied on Port 2. The
output signals on Port 4 to 6 are squared, filtered by a LPF and signal on Port 3 and Port 6 are
subtracted from signal on Port 4 and 5 respectively. The schematic diagram of six-port receiver
is shown in Figure 5.6
Figure 5.6 Schematic diagram of a Six-port demodulator
7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.97.0 8.0
-80
-70
-60
-50
-40
-30
-90
-20
Frequency (GHz)M
od
ula
ted
Sig
na
l P
ow
er
(dB
m)
60
Zero-biased Schottky diodes are used for squaring the signals. As an alternative of LPF, open-
circuited radial stubs are used as notch filters on first and second harmonics of the carrier
frequencies. Operational-amplifiers are used as signal subtracting device. The detail of each
device is presented in the following sections.
5.2.1 Diode modeling
To square the signals, zero-biased Schottky diodes are used. Schottky diodes have fast switching
speed and low cut-in voltage [44]. The diode model HSMS-286B from Avago Technologies is
used. The diode is modeled in ADS along with the package-parasitic effects provided in the
datasheet. The ADS model of the diode is shown in Figure 5.7.
Figure 5.7 ADS model for HSMS-286B Schottky diode according to parameters provided in the datasheet
Along with the package-parasitic effects the input impedance of the diode model is measured to
be 49.84-j8.09 Ω, so no matching network is used between each port of correlator and diode as
the value is nearly equal to 50 Ω. The data for input impedance values for the diode model is
shown in Figure 5.8.
Figure 5.8 Input impedance of the modeled Schottky diode
7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.97.0 8.0
0
20
40
-20
60
Frequency (GHz)
Inp
ut Im
pe
da
nce
(O
hm
)
Readout
m1
Readout
m2
m1freq=real(Zin1)=49.85
7.50GHz
m2freq=imag(Zin1)=-8.09
7.50GHz
7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.97.0 8.0
0
20
40
-20
60
Frequency (GHz)
Input Im
pedance (
Ohm
)
Readout
m1
Readout
m2
m1f req=real(Zin1)=49.85
7.50GHz
m2f req=imag(Zin1)=-8.09
7.50GHz
real(Zin1)
imag(Zin1)
Real
Imag
61
5.2.2 Notch filter
The squaring device (Schottky diode) produces the harmonics of the fundamental frequency of
the carrier wave. To remove the harmonics, LPF can be used. Here notch filters instead of LPF
are used. The notch filter can be implemented by an open-circuited quarter-wave stub. To
broadband the filter, radial-stubs are used. Notch filters for first-order and second-order
harmonics are implemented and higher order harmonics are neglected as they have quite weak
power. The response of the implemented notch filter is shown in Figure 5.9. The insertion loss is
more than 50 dB at 7.5 GHz and 15 GHz.
Figure 5.9 Response of implemented notch filter for demodulation.
5.2.3 Digital judgment circuit
The demodulated signal represents each transmitted symbol by some voltage level. To decide
about the received data a judgment circuit is used to compare the demodulated signal with
threshold values and adjust the voltage to the accurate levels. The judgment circuit can be an
analog or a digital circuit. In analog processing, judgment circuit can be implemented by
instrumentation amplifier [45]. In this project digital technique is used. ADS Quantizer block in
DSP processing is used. This block takes the demodulated signal as input, compares it with the
preset thresholds to decide about the transmitted symbol and gives the baseband bits as its
output.
5.3 Six-port transmitter-receiver system
A full six-port transmitter-receiver system is implemented using behavioral modeling of
modulator and demodulator. The full transmitter-receiver system block diagram is shown in
Figure 5.10.
2 4 6 8 10 12 14 16 180 20
10
20
30
40
50
60
0
70
Frequency (GHz)
Inse
rtio
n L
oss (
dB
)
62
Figure 5.10 Complete six-port 8-PSK transmitter-receiver system implemented using mixed analog-DSP design
Transmitter-side baseband processing
Three binary bit-sources with same data-rate are used as input baseband signals. The input data
sources are processed to produce Port 5 data by the operation explained in section 5.1.2. The
digital signals are then converted to analog signals and applied to Port 4 – 6 of modulator six-
port. In parallel, the baseband binary signals are sampled exactly in the middle of each bit and
delayed. The output of this operation is used to be compared with the output of receiver after
demodulation.
Modulator, channel and demodulator
The modulator six-port uses 8-PSK scheme to modulate the applied 7.5 GHz LO signal
according to the baseband data. The modulator’s block diagram is shown in Figure 5.1. The
values of Z1 and Z2 are chosen to produce reflection coefficients of +0.414 and -0.414.
The modulated wave is transmitted over a line-of-sight (LOS) wireless link. The model of LOS
link provided in ADS is used.
The channel output is applied to demodulator six-port. The internal working of demodulator is
shown in Figure 5.6.
Down sampling and delay
BER calculation
Processing for Port 5
data
Down sampling
A/D conversion
Modulation, LOS link,
DemodulationD/A
conversionQuantiz-
ation
Baseband Signal
Generation
63
Receiver-side baseband processing
The demodulated symbols are converted to digital signal to be processed by DSP. The Quantizer
blocks in ADS are used to decide about the received symbol with number of preset threshold
levels 4 and 2 for in-phase and quadrature-phase data respectively. Two symbols are recovered
from in-phase data and one symbol from quadrature-phase data. The recovered binary bits are
sampled in the middle of the bit interval and compared with the sampled bits on transmitter side
to calculate bit error rate (BER).
64
6 Designed Six-port Transceiver System Evaluation
In this chapter the system level evaluation of single-ended, simple differential and differential
with cross-conductors (wideband) Six-port transceiver designs is discussed. ADS simulation
results of the Six-port transceiver system comprising the electromagnetic models of single-ended
and differential designs are presented and different system level parameters such as Noise figure,
BER (Bit Error Rate) and Dynamic Range are evaluated and compared.
6.1 Noise Figure Comparison
Noise figure of a component is defined as the ratio of input SNR to output SNR. The noise figure
of the whole receiver system is dominated by the noise figure of first active component. The
sensitivity of a wireless receiver depends on the receiver’s noise figure [19].
For single-ended and differential six-port designs the noise figures are obtained using S-
parameter simulations with “noisecalc” option enabled. For the six-port modulator, Port 3 to Port
6 are terminated with voltage-controlled variable impedances. Noise figure simulations are
performed with Port 1 as input (LO signal) and Port 2 as output port (RF output signal). For the
six-port demodulator, Port 1 is used as LO port, Port 3 to Port 6 are attached with diodes, notch
filters and differential amplifiers to recover baseband signals and modulated RF signal is applied
on Port 2. The simulations are performed to obtain noise figure between the input modulated
signal (Port 2) and output demodulated signal.
Differential systems are claimed to have the advantage of better noise rejection [12]. The
simulated results showed the same phenomena. The compared results of all three designs for
transmitter-side or modulator-side are shown in Figure 6.1.
In Figure 6.1 noise figure for Differential designs is less than that for Single-ended design. The
differential system does not depend on ground reference-plane for working so it does not
experience ground noise. For single-ended design, the NF was relatively low until 7.0 GHz but
with increase in frequency, it started to increase. For differential design the maximum noise
figure is 3.3 dB at 7.3 GHz.
65
Figure 6.1 Transmitter-side Noise Figure comparison of Single-ended, Differential and Differential with crossed conductors
Figure 6.2 Receiver-side Noise Figure comparison of Single-ended, Differential and Differential with crossed conductors
Figure 6.2 shows the receiver noise figure for all three systems. The differential design has the
best noise figure with value less than 2 dB for 6 – 8.5 GHz. The differential design with crossed
conductors has more noise than simple differential. The reason for this is that signal is passed
through VIA holes several times.
6.2 BER and Dynamic Range Comparison
The dynamic range of a receiver is the range of RF input power over which the receiver can
correctly demodulate the received signal. The lower boundary of the range is restricted by the
sensitivity of the receiver. A more sensitive receiver can sense a minor signal increasing dynamic
6.5 7.0 7.5 8.0 8.56.0 9.0
2
3
4
5
1
6
Frequency (GHz)
No
ise
Fig
ure
(d
B)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))Diff with VIAsDifferentialSingle-ended
6.5 7.0 7.5 8.0 8.56.0 9.0
0.5
1.0
1.5
2.0
2.5
3.0
0.0
3.5
freq, GHz
Tra
nsm
itte
r N
ois
e F
igu
re (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
2
3
4
5
6
7
8
9
1
10
Frequency (GHz)
Re
ce
ive
r N
ois
e F
igu
re (
dB
)
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2))
6.5 7.0 7.5 8.0 8.56.0 9.0
-50
-40
-30
-20
-10
-60
0
Frequency (dB)
Inp
ut R
efle
ctio
n (
dB
)
dB(S(1,1))
dB(S(2,2))
dB(_2_SixportSE..S(1,1))
dB(_2_SixportSE..S(2,2)) Diff with VIAsDifferentialSingle-ended
66
range. The upper boundary of dynamic range is usually restricted by the 1-dB compression point
of the power amplifier used in the receiver [19].
For the designed six-port receiver the BER and Dynamic Range comparison is performed by
varying the RF power at the receiver side and recording the corresponding BER while keeping
all other system parameters constant. For that purpose the transmitter and receiver are connected
through an ideal amplifier model provided in ADS. The amplifier gain is first decreased below 0
dB in steps, using amplifier as an attenuator. The RF power and BER are recorded for each
simulation. Then the gain is increased in steps above 0 dB (amplifying the signal) and again RF
power and BER are recorded. The same simulation setup is used for all three six-port
transceivers. The results are plotted in Figure 6.3 as obtained BER vs. receiver input RF power.
Figure 6.3 Dynamic range comparison of single-ended, differential and differential with crossed conductors design
As can be seen in Figure 6.3, generally for all three designs there is a lower and an upper limit
beyond which the BER is very high. The lower limit is regarded as the Sensitivity of the receiver
and is a certain threshold value of RF signal power below which it can’t interpret the signal. The
upper limit is dependent on the threshold power levels of the Quantizer above which the digital
decision making circuitry is unable to decode the signal properly.
The results show the similarity in behavior among the Single-ended and Simple Differential
designs whereas the Differential with cross conductors require more power to interpret the signal
properly and has less dynamic range.
The sensitivity of the single-ended and differential are found approximately the same. The proper
signal reception starts at approximately 0 dBm which is a high value as compared to
commercially available systems. The values recorded are without using low noise amplifier
(LNA). A LNA can be embedded in the system to increase the sensitivity.
67
6.3 BER and Data rate comparison
BER of the single-ended design is also compared with differential design by varying the data-
rate of the system. Symbol rate of the transceiver system is varied from 100 Msymbols/s to 500
Msymbols/s and the corresponding BER is recorded. The results are shown for the Single-ended
and simple differential design in Figure 6.4
Figure 6.4 BER vs Datarate comparison for Single-ended and Simple differential design
The BER for differential design is less than single ended for symbol rate upto 250 MSymbol/s.
From 300 to 400 MSymbol/s single-ended performs better. Although the noise figure for
differential is better but for certain data-rates its BER is more than single-ended. The reason is
that bit-error rate not only depends on noise conditions but also on the designing, I/Q mismatches
and detecting circuitry.
6.4 Modulated Signal Constellation Diagrams and Power Spectrum Comparison
The modulated signal I/Q constellation diagrams and power spectrum is compared for all three
designs. The analysis is done on data-rate of 500 Mbit/s and LO signals power of 0 dBm. For
I/Q constellation the ADS sub-circuit block “TkConstellation” is used which is present in the
DSP components library “Interactive Controls and Displays”. For viewing Signal power
spectrum, “Spectrum Analyzer” is used.
68
(a)
(b)
6.4 6.6 6.8 7.0 7.2 7.4 7.6 7.8 8.0 8.2 8.4 8.66.2 8.8
-100
-80
-60
-40
-120
-20
Frequency (GHz)M
od
ula
ted
Sig
na
l P
ow
er
(dB
m)
6.4 6.6 6.8 7.0 7.2 7.4 7.6 7.8 8.0 8.2 8.4 8.66.2 8.8
-90
-80
-70
-60
-50
-40
-100
-30
Frequency (GHz)
Mo
du
late
d S
ign
al P
ow
er
(dB
m)
69
(c)
Figure 6.5 8-PSK modulated signal constellation and spectrum for (a) Single-ended (b) Differential (c) Differential with crossed conductors
The 8-PSK constellation for single-ended design is almost same as for the differential design.
For Differential with crossed conductors the constellation diagram (Figure 6.5c) shows a
displacement from central point. This shows that LO signal leakage is present in the modulator.
Also in its signal power spectrum there is a spike at centre frequency of 7.5 GHz which is
evidence of carrier leakage. Rest of the spectrum shows lesser power compared to single ended
and simple differential design.
7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.97.0 8.0
-80
-70
-60
-50
-40
-30
-90
-20
Frequency (GHz)M
od
ula
ted
Sig
na
l P
ow
er
(dB
m)
70
7 Conclusion & Future Work
7.1 Conclusion
In this thesis work, the main focus was on the design of Six-port Direct carrier
modulator/demodulator for 8-PSK modulation/demodulation with Single ended, Differential and
Differential with crossed conductors (or phase inverter) signaling types and their performance
comparison based on system level parameters. The designing for the above mentioned three
types was done in ADS and prototypes were fabricated in PCB lab. 8-PSK modulation and
demodulation was done on simulated electromagnetic models of the three designs. For this
purpose co-simulation was performed involving Analog/RF and DSP simulations. DSP
simulation was used to produce/recover the baseband signals and to observe the modulated
signal constellation and power spectrum whereas Analog/RF simulation was done for
modulation/demodulation through Six-port correlator. After that a whole communication system
was modeled and performance comparison on the basis of different system parameters was done
in ADS. Noise Figure, BER, Dynamic range, I/Q constellation diagrams and Power spectrum
was compared for the designs.
Through noise figure comparison, we observed that both differential designs showed much better
performance than single-ended. This proves the fact that differential system offers the advantage
of better noise rejection. There was not much to choose between different designs through BER
vs. dynamic range and BER vs. data-rate comparison i.e. they were almost the same. I/Q
constellation was best for simple differential design and worst for differential with crossed
conductors due to the presence of carrier leakage. The results show that differential is better
choice in terms of noise rejection but on the other hand it has a disadvantage of complexity.
Differential signaling is harder to manage. The fabrication process demands much more
accuracy. The source should be able to provide perfect complementary signals i.e. with equal
magnitude and 180o phase apart signals. Also, the differential circuit should be able to maintain
this coupling.
7.2 Future Work
In this thesis work, the variable impedance termination was managed through ideal switches in
ADS. In order to design whole system on real-component basis, the ideal switch can be replaced
with a PIN, Schottky diode switches or a transistor. An effort was done in this work too but it
required lot more work and time so it is suggested for future work.
Also, for complete and accurate comparison between single ended and differential systems, this
system level designing can be performed on lower frequency such as 2.5 GHz so the parasitic
71
losses would not much affect the results. This can be useful in future research on the comparison
between single ended and differential systems.
72
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