IEEE PEDS 2017, Honolulu, USA

12 - 15 December 2017

Efficiency Improvement of High Frequency Inverterfor Wireless Power Transfer System Using a Series

Reactive Power Compensator

Jun OsawaGraduate School of Pure and Applied Sciences

University of Tsukuba

Ibaraki 305–8573, Japan

Takanori Isobe, Hiroshi TadanoFaculty of Pure and Applied Sciences

University of Tsukuba

Ibaraki 305–8573, Japan

Abstract—This paper proposes a wireless power transfersystem using a series compensator GCSC as a primary sidecapacitor. The GCSC is a circuit module that equivalently func-tions as a series variable capacitor by controlling semiconductorswitches. The advantage of applying the GCSC to a primaryside capacitor is giving a controllability of power factor for ahigh frequency inverter. Therefore, the optimum operation ofthe high frequency inverter can be achieved regardless of thecoil parameters. Experimental results with an 1 kW laboratoryprototype confirmed that the proposed system can achieve anoptimum operation and high efficiencies of the high frequencyinverter.

I. INTRODUCTION

In recent years, wireless power transfer (WPT) systems

are activity studied. Especially, WPT systems for Electric

Vehicles (EVs) or Plug-in Hybrid Electrical Vehicles (PHEVs)

are highly paid attention since it can improve safety and

convenience of charge operation.

A magnetic resonant coupling is the leading circuit topology

of WPT for EVs or PHEVs since it enables high-power and

high-efficiency power transfer with a large air gap. In this

topology, inductive reactance of the coils are compensated by

using resonant capacitors to reduce current or voltage applied

to the coils and the power source. However, there is one

challenge that the reactance of the coil is not fixed because the

air gap between the primary and secondary coils is expected

to change according to the parking position. So the optimum

capacitance for the resonant capacitors can vary, and fixed

resonant capacitors cannot achieve the optimum operation

constantly in that situation.

To address this problem, this paper proposes a WPT system

using an active series reactive power compensator named

GCSC (Gate Controlled Series Capacitor)[1][2] instead of

the fixed resonant capacitors. The GCSC is a circuit module

connected in series that equivalently functions as a variable

capacitor. In the proposed system, the optimum operation

can be realized by controlling the equivalent capacitance of

the GCSC according to the reactance of the coil. The coil

efficiency improvement using the GCSC on the secondary

side has been proposed in [3]. This paper proposes an inverter

Fig. 1. Circuit configuration of the gate controlled series capacitor (GCSC).

optimum operation by using the GCSC on the primary side.

This paper reports experimental verifications of the proposed

system with an 1 kW laboratory prototype and discusses the

efficiencies of the high frequency inverter and the GCSC.

II. SERIES COMPENSATOR GCSC

A. Circuit Configuration and Features

Gate controlled series capacitor (GCSC) is one of FACTS

(flexible ac transmission system) devices proposed as a series

reactive compensator applied for ac power transmission lines.

Fig. 1 shows the configuration of the GCSC. The GCSC is

consists of two full-controlled reverse conductive semiconduc-

tor switches and a capacitor. The GCSC is controlled by line

frequency switching, and the two switches share their source

potential therefore the gate drive circuit can be simple. The

GCSC can achieve soft-switching in all the operating range

and there is no conduction loss when current is conducting

the capacitor. Therefore, loss in the semiconductor switches is

comparatively low.

B. Operation Principles

Switch S1 and S2 are complementary turned on with the

line frequency and a half duty ratio. The phase angle difference

between the line current phase and the gate signals, δ, is given

for control the equivalent capacitance of the GCSC , CGCSC.

Fig. 2 shows the schematic waveforms of the GCSC and Fig. 3

shows possible current paths. The GCSC has two types of

current paths, one is conducting the capacitor as shown in (a)

(b) (d) (e), and the other is conducting semiconductor switches

as shown in (c) (f). The capacitor voltage waveform has a zero

voltage period which is achieved by modes (c) and (f), and its

978-1-5090-2364-6/17/$31.00 c©2017 IEEE

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-1

0

1

-1

0

1

0

1

Cu

rren

tC

apac

ito

rS

2 s

ign

alv

olt

age

(a)

-1

0

1

-1

0

1

0

1

Cu

rren

tC

apac

ito

rS

2 s

ign

alv

olt

age

δ

(b)

-1

0

1

-1

0

1

0

1

Cu

rren

tC

apac

ito

rS

2 s

ign

alv

olt

age

δ

(c)

Fig. 2. Three waveform modes of the GCSC with different control phase angle conditions of (a)δ = 0, (b)0 < δ < 90, and (c)δ = 90.

Fig. 3. Possible current paths of the GCSC.

duration can be controlled by δ. When δ = 0 , flowing current

conducts only the semiconductor switches and the capacitor is

not charged, that means CGCSC = ∞. By increasing δ, flowing

current conducts the capacitor and generated voltage in the

capacitor increases. Finally, when δ = 90, flowing current

conducts only the capacitor, that means CGCSC = C, where

C is the capacitance of the equipped capacitor. Therefore, the

GCSC can vary its equivalent capacitance from C to ∞ by

controlling δ.The equivalent reactance of the GCSC can be derived by

the similar way discussed in [4]. The injecting voltage, vgcsc,

within a half fundamental cycle (0 < θ < π) can be derived

as

vgcsc=

√2XcI (sin θ−cos δ)

(

π

2− δ<θ< π

2+ δ

)

0 (else) ,(1)

where the pure sinusoidal current as

i =√2I cos θ, (2)

is assumed. The rms value of the fundamental component, V1,

can be derived as

V1 =2√2π

∫

π

0

vgcsc(θ) sin θdθ

= IXc

(

2δ

π− sin 2δ

π

)

, (3)

and then, the equivalent reactance, XGCSC, can be derived as

XGCSC = V1/I

= Xc

(

2δ

π− sin 2δ

π

)

. (4)

The current conducting semiconductor switches, isw, within

a half fundamental cycle(0 < θ < π) can be derived as

isw=

0(

π

2− δ<θ< π

2+ δ

)

√2I cos θ (else) ,

(5)

The loss of the GCSC, PGCSC, is derived as

PGCSC =1

π

∫

π

0

roni2sw(θ) + VFisw(θ)dθ

=1

π(ronI

2(π − 2δ + sin (π − 2δ))

+√2VFI sin (

π

2− δ))), (6)

where ron is the on resistance of the MOSFETs, VF is the

forward voltage of the free-wheeling diodes.

III. INVERTER LOSS REDUCTION BY APPLYING GCSC

A. Optimum Operation of High Frequency Inverter

Generally, the inverter of WPT systems for EV is operated

at high switching frequency, for instance 85 kHz; therefore, the

switching loss reduction is important to improve the system

efficiency. A voltage-source type high frequency inverter with

fundamental frequency switching, which means the output

current frequency is same as the switching frequency, is used

for the purpose.

A lagging power factor at the inverter output, as shown in

Fig. 4(a), is attractive for this topology, in which the turn-

off is performed with mitigated dv/dt achieved by a snubber

capacitor and/or device parasitic capacitance, and turn-on is

performed with conducting free-wheeling diode, so that turn-

on is complete soft-switching. On the other hand, the leading

power factor, as shown in Fig. 4(b), is not attractive since

it results in shorting the capacitors and turn-off of the free-

wheeling diode that causes reverse recovery, and the switching

993

(a)

(b)

Fig. 4. Schematic waveforms of a switching device of the high frequencyinverter with (a)Lagging power factor, (b)Leading power factor.

loss increases. And same phenomena occurs if the power factor

is too high as the current crosses the zero during the dead time.

Therefore, the high frequency inverter should be operated

at lagging power factor that delays current beyond the dead

time. At the same time, in order to minimize the voltage and

current at turn-off and to reduce the turn-off losses, the power

factor should be as high as possible. Therefore, operating the

high frequency inverter at slightly lagging power factor which

the current crosses zero at the end of dead time is optimum

and can minimize the switching loss.

B. Selection of Resonant Capacitor

In the WPT system in series-series circuit topology [5]

which is shown in fig. 5, the secondary side capacitor can

improve the efficiency of the coils by compensating for the

secondary side self inductance and minimizing the primary

current to achieve the same power transfer. Efficiency of the

coils is maximized when the secondary side capacitor fully

compensates for the self inductance of the secondary coil. The

primary side capacitor improves the output power factor of

the high frequency inverter. The capacitance of the primary

capacitor is selected to operate the high frequency inverter at

the slightly lagging power factor.

When the secondary side capacitor fully compensates for

the self inductance of the secondary coil, the power factor at

L1 L2

C1 C2

RL

M

Fig. 5. An equivalent circuit diagram of the WPT system of the series-seriestopology in the fundamental frequency.

0 50 1000

0.5

1

C1=11.68 nF

po

wer

fac

tor

Mutual inductance [μH]

C1=12.79 nF

C1=13.49 nF

Fig. 6. Example of the resulting power factor with varying mutual inductance,M , and fixed primary capacitance, C1.

0 50 10011

12

13

14

pf=0.9

Mutual inductance [μH]

Pri

mar

y c

apac

itan

ce [

nF

]

pf=0.8

pf=1

Fig. 7. Required primary capacitance, C1, to achieve constant power factorsfor varying mutual inductance, M .

the inverter output can be derived as

pf = cos

(

arctan(ωL1 − 1/ωC1)RL

(ωM)2

)

, (7)

where L1 is the self inductance of the primary coil, C1 is

the capacitance of the primary side series capacitor, RL is the

load equivalent resistance, and M is the mutual inductance

between the coils. To operate the high frequency inverter at

a lagging power factor, C1 should be selected appropriately

to compensate for L1 partially. However, the power factor is

influenced by the possible mutual inductance changing in WPT

applications. Therefore, fixed capacitors can not achieve the

optimum operation for varying mutual inductance.

In conventional WPT systems using a fixed capacitor, the

994

(a) (b)

Fig. 8. Coils used in the experiment at (a)primary side (b)secondary side.

TABLE ISPECIFICATIONS OF COILS

Primary Secondary

Winding type Spiral typeSize (W / D / H) 300 mm / 300 mm / 30 mmWire specification Litz wire (φ0.08 mm × 30)Number of Parallels 30 14Number of Turns 24 45Core PC95 (ferrite)Resistance 0.21 Ω 0.59 Ω

primary capacitor should be designed to achieve a slightly

lagging power factor when mutual inductance is the largest

value, to achieve the ZVS turn-on for all the range of varying

mutual inductance.

For example, the resulting power factor with fixed L1

(300 µH), fixed RL (100 Ω), and varying M is shown in

Fig. 6. As can be seen from the figure, one capacitance can

achieve unity power factor regardless of varying M ; however,

the other values of capacitance can not achieve constant power

factor as function of M . Therefore, fixed capacitance can not

achieve a lagging constant power factor for varying M .

Fig. 7 shows required C1 to achieve constant power factors

for varying M . As can be seen from the figure, varying C1 is

needed to maintain the power factor at a certain value except

for unity power factor. This is the reason for the need of the

active compensator for the WPT applications with possible

miss alignment. In the proposed system, the power factor can

be kept constant by controlling the equivalent capacitance of

the GCSC connected in the primary side.

IV. EXPERIMENTAL VERIFICATIONS

A. Experimental Setup

To confirm the above discussion, experiments were con-

ducted. Fig. 8 shows the picture of coils used in the ex-

periments. The coil shape is a spiral winding type and the

diameter is 300 mm. Litz wires were used to reduce the loss

increased by skin effect and proximity effect, and ferrite cores

are placed on the back side of the winding to increase the

coupling coefficient and quality factor. In order to suppress

the increase of the primary copper loss, which is the dominant

loss component of the coils, the primary coil was designed to

reduce resistance by increasing the number of parallels of the

winding.

0 40 80 1200

200

400

0

600

1200

Horizontal gap [mm]

L2 [

μH

]

L1,M

[μ

H]

L1

L

Fig. 9. Inductance characteristics of the coils used in the experiments asfunction of the horizontal gap.

M

Vout

P1 P2 P3 P4

Fig. 10. The proposed circuit configuration for the WPT using the GCSCas a primary side capacitor.

TABLE IIEXPERIMENTAL CONDITIONS

Fixed cap. A Fixed cap. B with GCSC

Frequency 85 kHzRated power 1 kWDead time 800 ns

Vout 350 V

C – – 32.09 nFC1 11.28 nF 12.29 nF 17.42 nFC2 3.09 nF

Table I lists specifications of the coils. In the experiments,

the vertical gap between the primary coil and secondary coil

was set constant at 100 mm and the horizontal gap was

changed from 0 mm to 120 mm. Fig. 9 shows varying L1,

L2 and M of the coils as function of the varying horizontal

gap. It can be seen from the figure that the self inductance

L1, L2 are almost constant; on the other hand, the mutual

inductance M highly varies with the gap change.

Fig. 10 shows the circuit diagram of the proposed system

using the GCSC as a primary side capacitor, which were evalu-

ated in the experiments. Table II shows the circuit parameters

of the experiments. The power supply voltage was adjusted

to achieve a constant output power of 1 kW for varying

condition of the horizontal gap. The secondary side capacitor

was selected to achieve full compensation for L2.

Three cases with different primary side capacitors were

evaluated as shown in the table. The first condition refereed as

fixed capacitor A was given to achieve full compensation for

L1. The second condition referred as fixed capacitor B was set

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-10

0

10

0 10 20

-500

0

500

Cap

acit

or

Cu

rren

t [A

]

time [μs]

vo

ltag

e [V

]

(a)

-10

0

10

0 10 20

-500

0

500

Cap

acit

or

Cu

rren

t [A

]

time [μs]

vo

ltag

e [V

]

(b)

-10

0

10

0 10 20

-500

0

500

Cap

acit

or

Cu

rren

t [A

]

time [μs]

vo

ltag

e [V

]

(c)

Fig. 11. Measured waveforms of the current flowing through the GCSC (top), and the capacitor voltage of the GCSC (bottom) when the horizontal gap was(a)0 mm, (b)60 mm, (c)120 mm.

0 40 80 1200

1

Po

wer

fac

tor

Horizontal gap [mm]

Capacitor A

Capacitor B

GCSC

Fig. 12. Measured power factors at the output of the high frequency inverterwith varying horizontal gap .

0 40 80 12090

95

100

Eff

icie

ncy

[%

]

Horizontal gap [mm]

Capacitor A

Capacitor B

GCSC

Fig. 13. Measured efficiencies of the high frequency inverter with varyinghorizontal gap.

to achieve partial compensation for L1 to achieve a slightly

lagging power factor when the mutual inductance becomes

maximum. The third condition is using the GCSC, which is

the proposed one in this paper.

In the proposed system using the GCSC, the GCSC and

0 40 80 1200

10

Horizontal gap [mm]

Curr

ent

[A]

Capacitor A

Capacitor B

GCSC

Fig. 14. Measured output current magnitude in rms of the high frequencyinverter with varying horizontal gap.

a fixed capacitor were connected in series and divided the

voltage in order not to exceed the rated voltage of the GCSC.

The capacitance of the GCSC and the fixed capacitor were

designed to achieve full compensation for the self inductance

of the primary coil with the maximum compensation degree of

the GCSC with some margin. At the same time, the voltage

sharing of the GCSC should be high as possible within an

available voltage rating of the device in order to reduce

the conduction loss of the semiconductor switches. For the

experiments, the capacitor in the GCSC was designed to be

applied up to 800 V with considering the use of 1200 V SiC-

MOSFET. Then, the equivalent reactance in the fundamental

frequency of the GCSC was controlled to obtain a slightly

lagging power factor that delays current beyond the dead time.

In the experiment, the power flows between components,

P1, P2, P3, and P4, shown in Fig. 10, were measured and

losses were evaluated.

B. Experimental Results

Fig. 11 shows the waveforms of the voltage of the GCSC

and line current when the horizontal gap was set at 0 mm,

60 mm and 120 mm. As can be seen from the figure, the GCSC

996

0 5 10-200

0

200

400

-5

0

5

10MOSFET Voltage

Free-Wheeling Diode current

MOSFET CurrentV

olt

age

[V]

time [μs]

Cu

rren

t [A

]

(a)

0 5 10-200

0

200

400

-5

0

5

10MOSFET Voltage

Free-Wheeling Diode current

MOSFET Current

Vo

ltag

e [V

]

time [μs]C

urr

ent

[A]

(b)

0 5 10-200

0

200

400

-5

0

5

10MOSFET Voltage

Free-Wheeling Diode current

MOSFET Current

Vo

ltag

e [V

]

time [μs]

Cu

rren

t [A

]

(c)

Fig. 15. Measured current and voltage waveforms of the semiconductordevices of the inverter when the horizontal gap was 0 mm. (a)With fixedcapacitor A. (b)With fixed capacitor B. (c)GCSC was used as a primary sidecompensator.

worked as an equivalent variable capacitor since the capacitor

conduction section varied according to the horizontal gap. The

current conducted the semiconductor switches only in a short

section and the semiconductor loss was generated only in this

section.

Fig. 12 shows the output power factor of the high frequency

inverter. In the case of fixed capacitor A, output power factor

of the high frequency inverter was kept at almost unity power

factor. In the case of fixed capacitor B, a slightly lagging

power factor was achieved when the horizontal gap was

0 mm; however, the power factor was significantly decreased

according to the increase of the horizontal gap. On the other

hands, in the case of the GCSC, the slightly lagging power

factor was achieved regardless of the horizontal gap.

Fig. 13 shows the efficiencies of the high frequency inverter.

The fixed capacitor A achieved lowest efficiencies than the

others regardless of the horizontal gap. The fixed capacitor

B achieved higher efficiencies than the capacitor A; however,

0 5 10-100

0

100

200

-10

0

10

20

time [μs]

Vo

ltag

e [V

]

Cu

rren

t [A

]

MOSFET Voltage

Free-Wheeling Diode Current

MOSFET Current

(a)

0 5 10-100

0

100

200

-10

0

10

20

time [μs]

Vo

ltag

e [V

]

Cu

rren

t [A

]

MOSFET Voltage

Free-Wheeling Diode Current

MOSFET Current

(b)

0 5 10-100

0

100

200

-10

0

10

20

time [μs]

Vo

ltag

e [V

]

Cu

rren

t [A

]

MOSFET Voltage

Free-Wheeling Diode Current

MOSFET Current

(c)

Fig. 16. Measured current and voltage waveforms of the semiconductordevices of the inverter when the horizontal gap was 120 mm. (a)With fixedcapacitor A. (b)With fixed capacitor B. (c)GCSC was used as a primary sidecompensator.

the efficiency was remarkably decreased by the horizontal gap

increase. In the case of the GCSC, inverter efficiencies were

always high and the efficiency decrease in high horizontal gap

conditions was mitigated.

Fig. 14 shows the output current of the high frequency

inverter in rms. The output current of the high frequency

inverter, which is therefore same as the primary coil current,

to achieve the same output power was almost same regardless

of the primary compensation. Therefore, the conduction loss

of the MOSFETs can be same for all three conditions and the

sole difference for the inverter efficiency is switching losses.

Fig. 15 and Fig. 16 show the waveforms of current and

voltage applied to the MOSFETs and external free-wheeling

diodes when the horizontal gap was 0 mm and 120 mm,

respectively. With the fixed capacitor A, turn-on was hard-

switching and the switching loss can be though to be high

since the parasitic capacitance was charged before the turn-

on, and it was shorted by the turn-on. With the fixed capacitor

997

0 40 80 1200

100

Horizontal gap [mm]

Lo

ss [

W]

InverterTransformer

Rectifier

(a)

0 40 80 1200

100

Horizontal gap [mm]

Lo

ss [

W]

Inverter

Transformer

Rectifier

(b)

0 40 80 1200

100

Horizontal gap [mm]

Lo

ss [

W]

GCSC

Transformer

Rectifier

Inverter

(c)

Fig. 17. Loss breakdown of the overall system with (a)Fixed capacitor A, (b)Fixed capacitor B, and (c)GCSC.

0 40 80 12090

95

100

Eff

icie

ncy

[%

]

Horizontal gap [mm]

Capacitor A

Capacitor B

GCSC

Fig. 18. Measured efficiencies of the overall system including the inverter,coils, rectifier, and compensators.

B, a completely soft turn-on and a turn-off at low current

were achieved when the horizontal gap was 0 mm. However,

the current at the turn-off was increased when the horizontal

gap was 120 mm due to the lower power factor at the inverter

output. That can increase the turn-off loss. On the other hands,

in the case of the GCSC, the slightly lagging power factor

achieved complete soft-switching turn-on and turn-off, and at

the same time, the current at the turn-off could be maintained

as low regardless of the horizontal gap.

Fig. 17 shows the overall system losses. The system losses

were divided into losses of the inverter, the coil part including

compensators (capacitors and GCSC) and the rectifier by

taking differences between the measured power, P1, P2, P3,

and P4, shown in Fig. 10. Then the losses of the GCSC were

calculated by using equation 6 and separated from the coil

losses. As can be seen from the figure, the loss of the coils

was almost same regardless of the compensation at the primary

side. The possible loss increase due to harmonic components

generated by the GCSC is considered to be negligible. The

loss of the GCSC was less than 3.5% of the overall loss and

was much smaller than other loss components. The inverter

loss using the GCSC was reduced by 8.4 W compared with

the system using fixed capacitors A and B at a maximum.

Fig. 18 shows the overall system efficiencies. The proposed

system using the GCSC achieved high efficiency compare

with the system using capacitor A in the whole area. When

the horizontal gap was larger than 80 mm, the proposed

system achieved high efficiency compare with the system

using the capacitor B since the loss reduction of the inverter

was remarkable and it exceeds the additional loss of the GCSC.

V. CONCLUSION

This paper proposed the WPT system using series com-

pensator named GCSC as a primary side capacitor. The

proposed system can control the output power factor of the

high frequency inverter and minimize its switching loss. The

main advantage of using the GCSC than the other topologies

like a full-bridge converter as a series compensator is low

semiconductor losses.

Experiments were conducted with an 1 kW laboratory

prototype. The proposed system achieved the efficiency im-

provement of the high frequency inverter, and considerably low

loss of the GCSC. The loss reduction of the inverter exceeded

the loss increase by the introduce of the GCSC; therefore,

the system overall efficiency was improved when the variation

degree of the coil parameters was large. The proposed system

reduced maximum loss by 8.4 W compared with the system

using fixed capacitors as a primary side capacitor.

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[2] E. H. Watanabe, L. F. W. de Souza, F. D. de Jesus, J. E. R. Alves andA. Bianco, “GCSC-gate controlled series capacitor. a new facts devicefor series compensation of transmission lines,” IEEE/PES Transmission

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[3] T. Isobe, K. Kobayashi, K. Wakasugi, R. Shimada, ”Efficiency Improve-ment of Contactless Energy Transfer Systems Using Series CompensationDevice Named MERS,” 14th European Conference on Power Electronics

and Applications (EPE 2011), 2011[4] T. Isobe, “A full-bridge AC power flow controller with reduced capac-

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[5] Yeong H. Sohn, Bo H. Choi, Eun S. Lee, Gyu C. Lim and Gyu-Hyeong Cho, “General Unified Analyses of Two-Capacitor InductivePower Transfer Systems: Equivalence of Current-Source SS and SPCompensations”, IEEE Transactions on Power Electronics, Vol. 30,No. 11, November 2015.

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