Electronic Design eBook _ Back to Basics_ Impedance Matching

BacktoBasics:ImpedanceMatching(Part1)

ByLouFrenzel

Thetermimpedancematchingisratherstraightforward.Itssimplydefinedastheprocessofmakingoneimpedancelooklikeanother.Frequently,itbecomesnecessarytomatchaloadimpedancetothesourceorinternalimpedanceofadrivingsource.

Awidevarietyofcomponentsandcircuitscanbeusedforimpedancematching.Thisseriessummarizesthemostcommonimpedancematchingtechniques.

RationaleAndConcept

The maximum powertransfer theorem says that to transfer themaximum amount of power from a source to a load, the loadimpedance should match the source impedance. In the basiccircuit, a source may be dc or ac, and its internal resistance (Ri)or generator output impedance (Zg) drives a load resistance (RL)or impedance (ZL) (Fig. 1):RL = Ri or ZL = Zg

A plot of load power versus load resistance reveals that matchingload and source impedances will achieve maximum power (Fig. 2).

A key factor of this theorem is that when the load matches thesource, the amount of power delivered to the load is the same asthe power dissipated in the source. Therefore, transfer ofmaximum power is only 50% efficient.

The source must be able to dissipate this power. To delivermaximum power to the load, the generator has to develop twicethe desired output power.

ApplicationsDelivery of maximum power from a source to a load occursfrequently in electronic design. One example is when the speakerin an audio system receives a signal from a power amplifier (Fig.3).

Maximum power is delivered when the speaker impedancematches the output impedance of the power amplifier.

Another example involves power transfer from one stage toanother in a transmitter (Fig. 4).

http://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig1_large.pnghttp://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig2_large.png

The complex (R jX) input impedance of amplifier B should bematched to the complex output impedance of amplifier A. Itscrucial that the reactive components cancel each other. Oneother example is the delivery of maximum power to an antenna(Fig. 5). Here, the antenna impedance matches the transmitteroutput impedance.

TransmissionLineMatchingThis last example emphasizes another reason why impedancematching is essential. The transmitter output is usually connectedto the antenna via a transmission line, which is typically coaxcable. In other applications, the transmission line may be atwisted pair or some other medium.

A cable becomes a transmission line when it has a length greaterthan /8 at the operating frequency where:

= 300/fMHz


For example, the wavelength of a 433MHz frequency is:

= 300/fMHz = 300/433 = 0.7 meters or 27.5 inches

A connecting cable is a transmission line if its longer than 0.7/8 =0.0875 meters or 3.44 inches. All transmission lines have acharacteristic impedance (ZO) thats a function of the linesinductance and capacitance:

ZO = (L/C)

To achieve maximum power transfer over a transmission line, theline impedance must also match the source and load impedances(Fig. 6).

If the impedances arent matched, maximum power will not bedelivered. In addition, standing waves will develop along the line.This means the load doesnt absorb all of the power sent downthe line.

Consequently, some of that power is reflected back toward thesource and is effectively lost. The reflected power could evendamage the source. Standing waves are the distributed patternsof voltage and current along the line. Voltage and current areconstant for a matched line, but vary considerably if impedancesdo not match.

The amount of power lost due to reflection is a function of thereflection coefficient () and the standing wave ratio (SWR).These are determined by the amount of mismatch between thesource and load impedances.

The SWR is a function of the load (ZL) and line (ZO) impedances:

SWR = ZL/ZO (for ZL > ZO)

SWR = ZO/ZL (for ZO > ZL)

For a perfect match, SWR = 1. Assume ZL = 75 and ZO = 50 :

SWR = ZL/ZO = 75/50 = 1.5

The reflection coefficient is another measure of the propermatch:

= (ZL ZO)/(ZL + ZO)

For a perfect match, will be 0. You can also compute from theSWR value:

= (SWR 1)/(SWR + 1)

Back to Basics: ImpedanceMatching, Part 1



http://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig6_large.png

Calculating the above example:

= (SWR 1)/(SWR + 1) = (1.5 1)/(1.5 + 1) = 0.5/2.5 = 0.2

Looking at amount of power reflected for given values of SWR(Fig. 7), it should be noted that an SWR of 2 or less is adequatefor many applications. An SWR of 2 means that reflected power is10%. Therefore, 90% of the power will reach the load.

Keep in mind that all transmission lines like coax cable dointroduce a loss of decibels per foot. That loss must be factoredinto any calculation of power reaching the load. Coax datasheetsprovide those values for various frequencies.

Another important point to remember is that if the lineimpedance and load are matched, line length doesnt matter.However, if the line impedance and load dont match, thegenerator will see a complex impedance thats a function of theline length.

Reflected power is commonly expressed as return loss (RL). Itscalculated with the expression:

RL (in dB) = 10log (PIN/PREF)

PIN represents the input power to the line and PREF is thereflected power. The greater the dB value, the smaller thereflected power and the greater the amount of power deliveredto the load.

ImpedanceMatching

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The common problem of mismatched load and source impedancescan be corrected by connecting an impedancematching devicebetween source and load (Fig. 8). The impedance (Z) matchingdevice may be a component, circuit, or piece of equipment.

A wide range of solutions is possible in this scenario. Two of thesimplest involve the transformer and the /4 matching section. Atransformer makes one impedance look like another by using theturns ratio (Fig. 9):

N = Ns/Np = turns ratio

N is the turns ratio, Ns is the number of turns on thetransformers secondary winding, and Np is the number of turnson the transformers primary winding. N is often written as theturns ratio Ns:Ns.

The relationship to the impedances can be calculated as:

Zs/Zp = (Ns/Np)2

or:

Ns/Np = (Zs/Zp)

Zp represents the primary impedance, which is the outputimpedance of the driving source (Zg). Zs represents thesecondary, or load, impedance (ZL).

For example, a driving sources 300 output impedance istransformed into 75 by a transformer to match the 75 loadwith a turns ratio of 2:1:

Ns/Np = (Zs/Zp) = (300/75) = 4 = 2

The highly efficient transformer essentially features a widebandwidth. With modern ferrite cores, this method is useful up toabout several hundred megahertz.


An autotransformer with only a single winding and a tap can alsobe used for impedance matching. Depending on the connections,impedances can be either stepped down (Fig. 10a) or up (Fig.10b).

The same formulas used for standard transformers apply. Thetransformer winding is in an inductor and may even be part of aresonant circuit with a capacitor.

A transmissionline impedancematching solution uses a /4section of transmission line (called a Qsection) of a specificimpedance to match a load to source (Fig. 11):

ZQ = (ZOZL)

where ZQ = the characteristic impedance of the Qsection line; ZO= the characteristic impedance of the input transmission line fromthe driving source; and ZL = the load impedance.

Here, the 36 impedance of a /4 vertical groundplane antennais matched to a 75 transmitter output impedance with a 52coax cable. Its calculated as:

ZQ = (75)(36) = 2700 = 52

Assuming an operating frequency of 50 MHz, one wavelength is:

= 300/fMHz = 300/50 = 6 meters or about 20 feet

http://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig11_large.pnghttp://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig10a_large.pnghttp://ebook.electronicdesign.com/Impedance-Matching/img/impedance_fig10b_large.png

/4 = 20/4 = 5 feet

Assuming the use of 52 RG8/U coax transmission line with avelocity factor of 0.66:

/4 = 5 feet (0.66) = 3.3 feet

Several important limitations should be considered when usingthis approach. First, a cable must be available with the desiredcharacteristic impedance. This isnt always the case, though,because most cable comes in just a few basic impedances (50, 75,93,125 ). Second, the cable length must factor in the operatingfrequency to compute wavelength and velocity factor.

In particular, these limitations affect this technique when used atlower frequencies. However, the technique can be more easilyapplied at UHF and microwave frequencies when using microstripor stripline on a printedcircuit board (PCB). In this case, almostany desired characteristic impedance may be employed.

The next chapter of this book will explore more popularimpedancematching techniques.


ByLouFrenzel

Duringimpedancematching,aspecificelectronicload(RL)ismadetomatchageneratoroutputimpedance(Rg)formaximumpowertransfer.Theneedarisesinvirtuallyallelectroniccircuits,especiallyinRFcircuitdesign.ThischapterwillintroducetheLnetwork,whichisasimpleinductorcapacitor(LC)circuitthatcanbeusedtomatchawiderangeofimpedancesinRFcircuits.

LNetworkApplicationsAndConfigurations

The primary applications of Lnetworks involve impedancematching in RF circuits, transmitters, and receivers. Lnetworksare useful in matching one amplifier output to the input of afollowing stage. Another use is matching an antenna impedance toa transmitter output or a receiver input. Any RF circuitapplication covering a narrow frequency range is a candidate foran Lnetwork.

There are four basic versions of the Lnetwork, with two lowpassversions and two highpass versions (Fig. 1). The lowpass versionsare probably the most widely used since they attenuateharmonics, noise, and other undesired signals, as is usuallynecessary in RF designs. The key design criteria are themagnitudes and relative sizes of the driving generator outputimpedance and load impedance.

1. There are four basic Lnetwork configurations. The networkto be used depends on the relationship of the generator andload impedance values. Those in (a) and (b) are lowpasscircuits, and those in (c) and (d) are highpass versions.

The impedances that are being matched determine the Q of thecircuit, which cannot be specified or controlled. If it is essentialto control Q and bandwidth, a T or network is a better choice.

http://ebook.electronicdesign.com/Impedance-Matching/img/impedance2_fig1_large.png

These choices will be covered in a subsequent article.

While the Lnetwork is very versatile, it may not fit every need.There are limits to the range of impedances that it can match. Insome instances, the calculated values of inductance orcapacitance may be too large or small to be practical for a givenfrequency range. This problem can sometimes be overcome byswitching from a lowpass version to a highpass version or viceversa.

DesignExample#1The goal is to match the output impedance of a lowpower RFtransistor amplifier to a 50output load, and 50 is a universalstandard for most receiver, transmitter, and RF circuits. Mostpower amplifiers have a low output impedance, typically less than50 .

2. The RF source is a transistor amplifier with an outputimpedance of 10 that is to be matched to 50 outputimpedance load. The Lnetwork with a parallel outputcapacitor is used.

Figure 2 shows the desired circuit. Assume an amplifier output(generator) impedance of 10 at a frequency of 76 MHz.Calculate the needed inductor and capacitor values using theformulas given in Figure 1a:

Q = [(RL/Rg) 1]

Q = [(50/10) 1] = [(5) 1] = 4 = 2

XL = QRg = 2(10) = 20

L = XL/2f

L = 20/[2(3.14)(76 x 106)]

L = 42 nH

XC = RL/Q

XC = 50/2 = 25

C = 1/2fXC


C = 1/[2(3.14)(76 x 106)(25)]

C = 83.8 pF

This solution omits any output impedance reactance such astransistor amplifier output capacitance or inductance and anyload reactance that could be shunt capacitance or seriesinductance. When these factors are known, the computed valuescan be compensated.

The bandwidth (BW) of the circuit is relatively wide given the lowQ of 2:

BW = f/Q = 76 x 106/2 = 38 x 106 = 38 MHz

You can see how this matching network functions by convertingthe parallel combination of the 50 resistive load and the 25capacitive reactance into its series equivalent (Fig. 3):

Rs = Rp/(Q2 + 1)

Rs = 50/(22 + 1) = 10

Xs = Xp/[(Q2 + 1)Q2]

Xs = 25/(5/4) = 25/1.25 = 20

Note how the series equivalent capacitive reactance equals andcancels the series inductive reactance. Also the series equivalentload of 10 matches the generator resistance for maximumpower transfer.

ParallelAndSeriesCircuitEquivalentsSometimes its necessary to convert a series RC or RL circuit intoan equivalent parallel RC or RL circuit or vice versa. Suchconversions are useful in RLC circuit analysis and design (Fig. 4).


4. These are all the possible practical series and parallel RCand RL circuit equivalents. The text provides the calculationsfor RS, RP, XS, and XP.

These equivalents also can help explain how the Lnetworks andother impedancematching circuits work. The designations are:

Rs = series resistance

Rp = parallel resistance

Xs = series reactance

Xp = parallel reactance

The conversion formulas are:

Rs = Rp/(Q2 + 1)

Xs = Xp/[Q2 + 1)Q2]

Rp = Rs (Q2 + 1)

Xp = Xs [(Q2 +1)/Q2]

Q = [Rp/(Rs 1)]

Q = XL/Rs

Q = Rp/XC

If the Q is greater than 5, you can use the simplifiedapproximations:

Rp =Q2Rs


Xp = Xs

DesignExample#2Match the output impedance of 50 from a 433MHz industrialscientificmedical (ISM) band transmitter to a 5 loop antennaimpedance (Fig. 5).

5. The RF source is a transmitter at 433 MHz with an outputimpedance of 50 . The load is a loop antenna with animpedance of 5 .

Q = [(Rg/RL) 1]

Q = [(50/5) 1] = [(10) 1] = 9 = 3

XL = QRL = 3(5) = 15

L = XL/2f

L = 15/2(3.14)(433 x 106)

L = 5.52 nH

XC = Rg/Q

XC = 50/3 = 16.17

C = 1/2fXC

C = 1/2(3.14)(433 x 106)(16.67)

C = 22 pF

In this example, the capacitor, inductor, and load resistance forma parallel resonant circuit (Fig. 6).


6. The equivalent circuit of the Lnetwork and load is a parallelresonant circuit. At resonance, the parallel circuit has anequivalent resistance equal to the generator resistance of 50 for a match.

Recall that a parallel resonant circuit acts like an equivalentresistance. That resonant equivalent resistance (RR) of a parallelRLC circuit can be calculated by:

RR = L/CR

or:

RR = R(Q2 +1)

RR = L/CR = 5.52 x 109/(22 x 1012)(5) = 50.18

RR = R(Q2 +1) = 5(32 + 1) = 50

In both cases the parallel resonant load equivalent resistance is 50 and equal to the generator resistance allowing maximum powertransfer. Again, adjustments in these values should be made toinclude any load reactive component. The equivalent highpassnetworks could also be used. One benefit is that the seriescapacitor can block dc if required.

AModernApplicationIn radio communications, a common problem is matching atransmitter, receiver, or transceiver to a given antenna. Mosttransceivers are designed with a standard 50 input or outputimpedance. Antenna impedances can vary widely from a few ohmsto over a thousand ohms.

To meet the need to match a transceiver to an antenna, themodern antenna tuner has been developed. Manual versions withtunable capacitors and switched tapped inductors have beenavailable for years. Today, modern antenna tuners areautomated. When the transceiver is in the transmit mode, thetuner automatically adjusts to ensure the best impedance matchpossible for maximum power transfer.


7. The basic impedancematching circuit in the MFJ EnterprisesMFJ928 automatic antenna tuner is an Lnetwork with aswitched tapped inductor and switched capacitors. SWR,impedance, and frequency sensors provide inputs to themicrocontroller whose algorithm switches the network seekingan impedance match.

Figure 7 shows a representative tuner. It is essentially an Lnetwork that is adjusted automatically by switching differentvalues of capacitance in or out and/or switching different taps onthe inductor to vary the inductance. A microcontroller performsthe switching according to some algorithm for impedancematching.

The criterion for determining a correct match is measuring thestanding wave ratio (SWR) on the transmission line. The SWR is ameasure of the forward and reflected power on a transmissionline. If impedances are properly matched, there will be noreflected power and all generated power will be sent to theantenna. The most desirable SWR is 1:1 or 1. Anything higherindicates reflected power and a mismatch. For example, an SWRvalue of 2 indicates a reflected power of approximately 11%.

In Figure 7, a special SWR sensor circuit measures forward andreflected power and provides proportional dc values to themicrocontroller. The microcontroller has internal analogtodigitalconverters (ADCs) to provide binary values to the impedancematching algorithm. Other inputs to the microcontroller are thefrequency from a frequency counter circuit and the actualcomplex load impedance as measured by an impedancemeasuringcircuit.

One typical commercial automated antenna tuner, the MFJEnterprises MFJ928, has an operating frequency range of 1.8 to30 MHz and can handle RF power up to 200 W. It has an SWRmatching range of 8:1 for impedances less than 50 and up to32:1 for impedances greater than 50 .

The total impedancematching range is for loads in the 6 to 1600 range. The range of capacitance is 0 to 3900 pF in 256 steps,and the range inductance is 0 to 24 H in 256 steps. Note that thecapacitance may be switched in before or after the inductor. Thisprovides a total of 131,072 different L/C matching combinations.

Such automatic tuners are widely used in amateur radio and the


military where multiple antennas on different frequencies may beused. The approach is also used in highpower industrialapplications, such as matching the complex impedance of asemiconductor etch chamber in an RF plasma etcher to the 50kilowatt power amplifier output. Motors are often used to varythe capacitors and inductors in a closed loop servo feedbacksystem.

ReferencesAutomatic Lnetwork calculator:http://bwrc.eecs.berkeley.edu/research/rf/projects/60ghz/matching/impmatch.htmlFrenzel, Louis E. Jr., RF Power for Industrial Applications.Prentice Hall/Pearson, 2004,www.ece.msstate.edu/~donohoe/ece4333notes5.pdf

Back to Part 1


ByLouFrenzel

TheLnetworkisarealworkhorseimpedancematchingcircuit.Whileitfitsmanyapplications,amorecomplexcircuitwillprovidebetterperformanceorbettermeetdesiredspecificationsinsomeinstances.TheTnetworksandnetworksdescribedherewilloftenprovidetheneededimprovementwhilestillmatchingtheloadtothesource.

http://bwrc.eecs.berkeley.edu/research/rf/projects/60ghz/matching/impmatch.htmlhttp://www.amazon.com/Power-Industrial-Applications-Louis-Frenzel/dp/0130965774

RationaleForUse

The main reason to employ a Tnetwork or network is to getcontrol of the circuit Q. In designing an Lnetwork, the Q is afunction of the input and output impedances. You end up with afixed Q that may or may not meet your design specs. In mostcases the Q is very low (

1. The network matching circuit is used mostly in high tolowimpedance transformations. The basic circuit (a) is a lowpass circuit. A high pass version (b) can also be used. The network also can be considered two backtoback Lnetworkswith a virtual impedance between them (c).

You may design a network using Lnetwork procedures. The network can be considered as two Lnetworks back to back. Touse the Lnetwork procedures, you need to assume anintermediate virtual load/source resistance RV as shown in Figure1c. You can estimate RV from:

RV = RH/(Q2 + 1)

RH is the higher of the two design impedances Rg and RL. Theresulting RV will be lower than either Rg or RL depending on thedesired Q. Typical Q values are usually in the 5 to 20 range. Anexample will illustrate the process.

NetworkDesignExampleAssume you want to match a 1000 source to a 100 load atfrequency (f) of 50 MHz. You desire a bandwidth (BW) of 6 MHz.The Q must be:


Q = f/BW = 50/6 = 8.33

RV = RH/(Q2 + 1) = 1000/[(8.33) 2 + 1] = 1000/70.4 = 14.2

The design procedure for the first Lsection uses the formulasfrom Back to Basics: Impedance Matching (Part 2). Use thedesired Q of 8.33 with an RL value equal to RV.

The inductor L1 value is:

XL = QRL = 8.333(14.2) = 118.3

L = XL/2f

L1 = 118.3/2(3.14)(50 x 106)

L1 = 376.7 nH

The capacitor C1 value is:

XC1 = Rg/Q

XC1 = 1000/8.33 = 120

C1 = 1/2fXC

C1 = 1/2(3.14)(50 x 106)(120)

C1 = 26.54 pF

Now calculate the second section with L2 and C2 using an Rgvalue of RV or 14.2 with the load RL of 100 . The Q is nowdefined by the Lnetwork relationship:

Q = [(RL/Rg) 1]

Rg in this case is RV or 14.2 .

Q = [(100/14.2) 1] = [(7) 1] = 6 = 2.46

The inductance L2, then, is:

XL2 = QRg = 2.46(14.2) = 35

L2 = XL/2f

L2 = 35/[2(3.14)(50 x 106)]

L2 = 111.25 nH

The capacitance C2 is:

XC2 = RL/Q

XC = 100/2.46 = 40.65

C2 = 1/2fXC

C2 = 1/[2(3.14)(50 x 106)(40.65)]

C2 = 78.34 pF

Note that the two inductances are in series so the total is just thesum of the two or:

L1 + L2 = 376.7 + 111.25 = 487.97 nH

Figure 2 shows the final circuit.

2. The network resulting from the example problem matches a1000 generator to a 100 load at a frequency of 50 MHz with abandwidth of 6 MHz and a Q of 8.33.

Two Web tools provide the same results:

http://bwrc.eecs.berkeley.edu/research/rf/projects/60ghz/matching/impmatch.html%20

http://www.raltron.com/cust/tools/network_impedance_matching.asp%20

TNetworksAndLCCDesignExampleFigure 3 illustrates the basic Tnetworks. The basic T shown inFigure 3a is not widely used, but its variation in Figure 3b is. Thesecond network is called an LCC network.

3. There are two versions of the Tnetwork, an alternate matchingnetwork: the low pass version (a) and the more popular LCCnetwork (b).

To design these networks, you can also consider them as twocascaded Lnetworks. However, since the version in Figure 3b is socommon, you can also use some shortcut formulas. Here is theprocedure:

1. Select the desired bandwidth and calculate Q.2. Calculate XL = QRg3. Calculate XC2 = RL[Rg (Q2 + 1)/RL 1]4. Calculate XC1 = Rg (Q2 + 1)/Q[QRL/(QRL XC2)]5. Calculate the inductance L= XL/2f6. Calculate the capacitances C = 1/2fXC

Assume a source or generator resistance of 10 and a loadresistance of 50 . Let Q be 10 and the operating frequency be315 MHz.

XL = QRg = 10(10) = 100

L = XL/2f = 100/2(3.14)(315 x 106) = 50 nH

http://ebook.electronicdesign.com/Impedance-Matching/img/impedance3_fig2_large.pnghttp://ebook.electronicdesign.com/Impedance-Matching/img/impedance3_fig3_large.png

XC2 = RL[Rg (Q2 + 1)/RL 1] = 50{[10(101)/50] 1} = 219

XC1 = Rg (Q2 + 1)/Q [QRL/(QRL XC2)] = 10(101)/10[500/(500 219)] = 179

C2 = 1/2fXC = 1/2(3.14)(315 x 106)(219) = 2.31 pF

C1 = 1/2fXC = 1/2(3.14)(315 x 106)(179) = 2.82 pF

ApplicationsWithTunableNetworksImpedancematching networks must be tunable before they canbe used over a wider frequency range or match a wider range ofimpedances for lower standing wave ratio (SWR) values. One ormore of the components must be variable to make such networks.While manually variable capacitors and inductors are available,they are too large for practical modern circuits and theirvariability cannot controlled electronically. Electronic controlpermits automatic tuning and matching circuits to be built.

Different types of electronically variable capacitors are nowavailable to implement such automatic tuners and matchingcircuits. The varactor diode or voltage variable capacitor has beenavailable for years, and its continuously variable nature over awide capacitance range is desirable. However, it is nonlinear andrequires a significantly high bias voltage for control. Two otheravailable options are the digitally tunable capacitor and themicroelectromechanicalsystems (MEMS) switched capacitor. Bothare available in IC form and are ideal for making highfrequencyvariable matching networks.

4. Peregrine Semiconductors solidstate UltraCMOS DTC forwireless applications uses five MOSFET switched capacitors.

Peregrine Semiconductors PE64904 and PE64905 Digitally TunableCapacitors (DTCs) consist of five fixed capacitors switched by anarray of MOSFET switches (Fig. 4). These devices are made using aunique silicononsapphire process. The capacitance is changed by


a serial 5bit code word using either a serial peripheral interface(SPI) or an I2C interface. The capacitor may be used in a series orparallel format. Maximum shunt capacitance is 5.1 pF with aminimum of 1.1 pF. Maximum series capacitance is 4.6 pF with aminimum of 0.6 pF. The capacitors can be put in series or parallelfor larger or smaller values.

Figure 5 shows a circuit using several of these capacitors. Its botha tunable filter and impedancematching circuit. Its also areconfigurable coupled resonator topology thats widely used inbandpass filters and impedancematching networks. Usingexternal inductors, this network provides wide impedancecoverage over the cellular phone bands of 698 to 960 MHz and1710 to 2170 MHz. Such tunable networks can provide automaticadjustment when cell sites are changed or can correct for theantenna detuning when the phone is held.

5. Tunable matching networks can use UltraCMOS DTCs totarget multiband LTE/WCDMA/GSM applications on UMTSFDDBands I, II, III, IV, V, VIII, and XII.

Wisprys MEMS devices also are commercially available tunablecapacitors. The basic element is a MEMS capacitor that can beswitched to produce an incremental change of 0.125 pF per bit.Each capacitor comprises a fixed plate with a dielectric on itssurface. Above it is a mechanical flap forming the other plate.When an electrostatic charge is applied to the plates, theyreattracted to one another. The upper plate is pulled downward,significantly decreasing plate spacing and increasing capacitance.By forming an array of these tiny capacitors, larger values can becreated.

The company makes several versions of this device includingcustom circuits. Capacitance values to a maximum of 10, 20, or30 pF can be formed with a tuning range of 10:1. The capacitanceis controlled digitally with a serial word using either an SPI or theMIPI radiofrequency frontend (RFFE) interface. Capacitivedevices use external inductors.

Also, Wisprys WS2017 standard impedance matching network is avariable network matching device featuring onchip inductors.Again, the primary application is automatic tuning and impedancematching in cell phones and other small RF equipment. It operatesover the 824 to 2170MHz frequency range. An onchip dcdcconverter provides the high voltage to operate the switchablecapacitors. The device uses an SPI for control. A similar product,the WS2018, has similar specifications as well as the MIPI RFFEinterface.


Copyright 2012 Penton Media, Inc., All rights reserved.

References

Bostic, Chris, RF Circuit Design, Howard W. Sams, 1982.Frenzel, Louis E., Principles of Electronic CommunicationSystems, McGraw Hill, 2008.

AbouttheAuthor

Lou Frenzel is the CommunicationsTechnology Editor for Electronic DesignMagazine where he writes articles, columns,blogs, technology reports, and onlinematerial on the wireless, communicationsand networking sectors. Lou has been withthe magazine since 2005 and is also editorfor Mobile Dev & Design online magazine.

Formerly, Lou was professor and department head at AustinCommunity College where he taught electronics for 5 years andoccasionally teaches as Adjunct Professor. Lou has 25+ yearsexperience in the electronics industry. He held VP positions atHeathkit and McGraw Hill. He holds a bachelors degree from theUniversity of Houston and a masters degree from the Universityof Maryland. He is author of 20 books on computer and electronicsubjects.

Lou Frenzel was born in Galveston, Texas and currently lives withhis wife Joan in Austin, Texas. He is a longtime amateur radiooperator (W5LEF).

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