Analysis and Design of High-Current Constant-Current Driver for Laser Diode Bar

Dong Chuanjie School of Automation

Beijing Institute of Technology Beijing, China

dchuanjie@163.com

Huang Hong School of Automation

Beijing Institute of Technology Beijing, China

honghuang@bit.edu.cn

Abstract—High power laser diode bar is formed from P-N junctions and powered by injected electric current. In order to drive it, it is essential to have low voltage, high and steady injected current for a power supply. So a constant-current converter with low ripple is necessary. In this paper, a two-channel interleaving buck converter with synchronous rectification and peak current mode PWM control is designed to drive the laser diode bar. The Operating Principle of the converter will be introduced. The mathematical mode of the converter in peak current mode will be analyzed and built. The prototype of the converter will also be designed and discussed in this paper.

Keywords-laser diode bar driver; constant-current source; interleaving; dc-dc converter

I. INTRODUCTION In the 1990s, several big companies in Europe and America

like SDL, Inc. began to supply commercial high power diode lasers successively, which made laser revolutionized progress [1]. For other kinds of laser owing to their complex mechanism, particularly large volume, weight and power consumption limit their application. However, laser diode is extensively used because of its small volume, light weight, low power consumption and high reliability.

As we know, Laser diode is powered by injected electric current. The stability of the injected current has a direct and apparent effect on the output of the laser. A tiny variation of the current may bring about big changes on the laser’s power output. Large current transient or ns pulse width of current peak may lead to reflection mirror overheating [2] of the laser and catastrophic optical damage in the end. So the driver should be design as constant current and have low ripple. Meanwhile, soft start circuit should be used to avoid the current transients at the start.

Linear power supply has many advantages such as simple circuit, small output ripple, easy to adjust the output current. In order to meet the requirement of the laser to current, linear power supply is largely used in the field of low power [3, 4, 5]. However, low efficiency, small volume and big heat limit its application in the field of high power. Switching mode power supply overcomes those shortcomings, and obtains the widespread application. A buck-mode laser diode driver delivering up to 2.5A to laser is designed in [6]. Operation of

parallel RC (resonant converter) as a constant current source is reported in [7]. A detailed analysis and design of LCL-T RC as constant current power supply is described in [8].

In this paper, the laser diode is used as a portable pumping source in fingerprint detector. The rated current of the laser is 40A, and voltage is 0-2V. The single-channel converter approach is not suitable because of the unacceptable thermal stress on the components. Consequently, a multi-channel converter should be used. Because of the 180 phase difference between the two channel, the two inductor ripple currents tend to cancel each other in interleaving converter. A smaller ripple current flows into the output capacitor as a result. The current sharing can be easily accomplished by using peak current mode control. Hence, a two-channel interleaving buck converter with Synchronous Rectification and peak current mode PWM control comes up in this paper.

II. ANALYSIS OF THE PROPOSED POWER SUPPLY The circuit diagram of the converter with peak current

mode control is shown in Fig.1. For convenience of analysis, we record the circuit in bold in the converter in Fig.1 as “channel 1”, and another as “channel 2”.

Figure 1. Circuit diagram of the interleaving buck converter with peak current mode control

A. Operating Principle Basic buck converter is represented in [9]. The two-channel

interleaving converter will be analyzed then. To analyze the operation principles, some assumptions are made as follows:

1321978-1-4577-0321-8/11/$26.00 ©2011 IEEE

• All circuit parameters are ideal. Parameters between channel 1 and channel 2 are the same. L1=L2=L, Q1=Q2=Q3=Q4.

• Output current oI is assumed to be constant. Input voltage inV and output voltage oV are unchanged.

• Duty cycle D<0.5, cycle is T. The operation of this circuit can be briefly described by

four operation modes as shown in Fig. 2.

Mode 1: time from t0 to t1 as shown in Fig.3 and the equivalent circuit is shown in Fig. 2(a). Switches Q1, Q4 are turned on and switches Q2, Q3 are turned off. There are two current conducting loops in this mode. In the first loop, the input power is transferred to the output load oR through switch Q1 and inductor L1 sequentially. The voltage across the first inductor L1 is 0in oV V− > , increasing its current 1LI linearly as a result. In the second loop, the inductor current 2LI passes through the output load oR and the synchronous fly-wheel MOSFET Q4 sequentially. The voltage across the second inductor L2 is 0oV− < , decreasing its current 2LI linearly as a result.

The related equations can be derived as below,

11

i oL

U UI DT

L−

Δ = (1)

2

2

oL

UI DT

L

−Δ = (2)

Mode 2: time from t1 to t2 as shown in Fig.3 and the equivalent circuit is shown in Fig. 2(b). Switches Q2, Q4 are turned on and switches Q2, Q3 are turned off. The inductor currents 1LI and 2LI pass through the output load Ro

and the fly-wheel MOSFETs Q2, Q4 sequentially.

The related equations can be derived as below,

11

(1 )2

oL

U D TI

L− −Δ = (3)

22

(1 )2

oL

U D TI

L− −Δ = (4)

Mode 3: time from t2 to t3 as shown in Fig. 3 and the equivalent circuit is shown in Fig. 2(c). Switches Q2, Q3 are turned on and switches Q1, Q4 are turned off. There are also two current conducting loops in this mode. In the first loop, the inductor current 1LI passes through the output load oR and the fly-wheel MOSFET Q2 sequentially. The voltage across the second inductor L2 is 0oV− < , decreasing its

current 1LI linearly. In the second loop, the input power is transferred to the output load oR through switch Q3 and inductor L1 sequentially. The voltage across the second inductor L2 is 0in oV V− > , increasing its current 2LI as a result.

The related equations can be derived as below,

2

2

i oL

U UI DT

L−

Δ = (5)

11

oL

UI DT

L−

Δ = (6)

Mode 4: time from t3 to t4 as shown in Fig.3 and the equivalent circuit is shown in Fig. 2(c). From Fig.3, we know that mode 4 is the same to mode 2.

From (1) and (2), we can derive the output ripple current:

1 2 (1 2 )oo L L

UI I I D T

LΔ = Δ + Δ = − (7)

(a)

(b)

(c)

(d)

Figure 2. Equivalent circuit of the converter. (a)mode 1, (b)mode 2, (c)mode 3, (d)mode 4.

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Figure 3. Corresponding waveforms of the converter.

B. The Mathematical Model Of The Converter As shown in Fig.1, the control architecture of the converter

consists of two loops: an inner loop based on the peak current mode control and an outer loop based on the output current. The aim of the inner loop is to adjust the peak current in the output inductors cycle by cycle according to the reference given by EA (Error Amplifier). From [10], we know that the inner and outer loops can be analyzed individually, when the cutoff frequency is 1/10 of the value of the switching frequency. In inner loop, including inductors L1 and L2, the modulator converts the output voltage CV in EA into inductance current LI , so the inner loop can be reduced as VCCS (Voltage Controlled Current Source) which transconductance is sg . Therefore, the model of the converter in peak current mode control can be simplified as shown in Fig.4.

Figure 4. Simplified model of the converter

For each channel, transconductance:

11

Ls

C

ig

v= (8)

Because of interleaving, the total current of the inductor:

1 2 12L L L Li i i i= + = (9)

From (8) and (9), we have:

12Ls s

C

ig g

v= = (10)

We obtain the simplified transfer function (TF) between the output current oi and EA-out cv as below:

1

1( )

1 ( )o ESR O

sc O ESR L SENSE

i R C sG s g

v C R R R s+

= =+ + +

(11)

We get poles and zeros:

12 ( )p

O ESR L SENSE

fC R R Rπ

=+ +

12z

ESR O

fR Cπ

=

Feedback loop contains sampling resistance SENSER , and the TF is :

2 ( ) FB

o

vG s

i= (12)

Where FBv is input voltage of EA.

III. PREPARE YOUR PAPER BEFORE STYLING Loop compensation is the adjustment of the control loop

frequency response to assure loop stability and optimize the transient response of the power supply. From (11), there is only one pole in peak current mode control, which makes it easy to design the compensation loop. The compensation circuit is indicated by shading in Fig.4. The TF is shown as follows:

( )( ) 1 1

1 21 2 1

1 2

13( )

( ) 1

m EAg R C sW s

C CC C s R s

C C

+=

+ ++

(13)

For 1 2C C , we write (13) as follows:

( )

( )( ) 1 1

1 2 1

13( )

1m EAg R C s

W sC s C R s

+=

+ (14)

Where: ( )m EAg is transconductance of EA.

We get poles and zeros as follows:

0

1

( )

1

2p

m EA

fC

gπ

= , 11 2

12pf

R Cπ= , 1

1 1

12zf

R Cπ=

.

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Where 0pf is set to be crossover frequency, 1pf is set to

be zf , 1zf is set to be pf .

IV. SIMULATION AND EXPERIMENTAL RESULTS In this section, we represent the experimental setup in Fig.

1. The nominal converter parameters are listed in TABLE . Two power diodes in series are used as the output load. The open loop transfer function of the converter can be obtained by modeling the converter with (11), (12), (13) and the nominal parameters as shown in TABLE 1. The Bode diagrams of the open loop without compensation (lines in blue) and the open loop with compensation (lines in green) stimulated by Matlab are shown in Fig. 5. It is shown that the cutoff frequency is 20 kHz, and the phase margin is 63°, which meet the requirement of stability. The output voltage startup waveform is presented in Fig. 6 with soft-start circuit. Fig. 7 shows the 180° interleaving gate voltage waveforms on the top N-channel MOSFETs between channel 1 and channel 2. Fig. 8 shows the gate voltage Vgs1 waveform and the drain-source voltage Vds1 waveform on the top MOSFET Q1.

TABLE I. NOMINAL CONVERTER PARAMETERS The parameters The values

Input voltage Vin 12V Output voltage Vo 0-2V Output current Io 40A Switch frequency f 140kHz Inductor L1,L2 4.7uH Output capacitor Co 4*470uF Compensator capacitor C1 1nF Compensator capacitor C2 150pf Compensator resistor R1 113k

Figure 5. The Bode diagrams of the circuit

Figure 6. Output voltage startup waveform

Figure 7. Interleaving waveforms of the converter

Figure 8. Voltage waveform on Q1. Ch1:(10V/div),Ch2:(5V/div).

V. CONCLUSION In this paper, the two-channel interleaving synchronous

buck converter in peak current mode PWM control is designed. The operating principles of the converter are analyzed. Then we deduce the transfer function of the converter in peak current mode. The prototype of the converter is designed at last.

REFERENCES

[1] Guo Wenwen, “Far-field characteristics simulation analysis of blue -violet LD”, vol. 1, Electronic Test, 2011, pp. 68-72.

[2] Liao Xianbing, “Laser-diode power supplies and its influence on device performances”, vol. 15, Semiconductor Optoelectronics, 1994, pp. 229-232.

[3] Deng Jun, Shan Jiangdong, Zhang Na, Tian Xiaojian, “Research and design of high-power semiconductor laser diode driver”, vol. 24(5), Semiconductor Optoelectronics, 2003, pp. 319-320.

[4] Chang Tieyuan, Zhu Guifeng, Chen Wenjun, “Design of high stable numerically controlled constant-current source”, Second International Conference On Future Networks, 2010, pp. 173-176.

[5] Jia Wenchao, Li Juanjuan, Liu Zengjun, Cheng Quanxi, “Design of electric power control system by single chip for semiconductor laser”, vol. 5, Modern Electronics Technique, 2008, pp. 190-191.

[6] Thompson, M.T., Schlecht M.F., “High power laser diode driver based on power converter technology” vol. 12, IEEE Transactions On Power Electronics, 1997, pp. 46-52.

[7] Borage M., Tiwari S., Kotaiah S., “A parallel resonant constant-current power supply”, J. Indian Inst. Sci., 2003, pp. 117–125.

[8] Borage M., Tiwari S., Kotaiah S. “Analysis and design of LCL-T resonant converter as a constant-current power supply”, vol. 52(6), IEEE Trans. Ind. Electron., 2005, pp. 1547–1554.

[9] Wang Zhaoan, “Switching power supply technology”, Beijing: China Machine Press,2004.

[10] Wang Hongyi, ” Studies on the stability of integrated current mode DC-DC converters focused on PVTL variations”, Xian:Xidian University, 2007.

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