Design and Experimental Validation of a Silicon Carbide 100kW Battery Charger Operating at 60kHz

Alejandro Rujas, Irma Villar, Ion Etxeberria-Otadui IK-4 Ikerlan Technological Research Centre

20500 Arrasate-Mondragón, Spain arujas@ikerlan.es

Uxue Larrañaga, Txomin Nieva CAF Power & Auttomation

20271 Irura, Spain ularranaga@cafpower.com

Abstract— This paper presents the design and development of a 100kW Silicon Carbide battery charger operating at a switching frequency of 60kHz. The main aspects of the converter design such as the topology, the gate driver and the electromechanical design are described in the paper. The converter has been developed and experimentally validated at full rate.

Keywords—silicon carbide, high power density, interleaved converter, power converter design

I. INTRODUCTION Energy storage technologies are significantly evolving in recent years, going from low power mobile/portable consumption devices to increasingly higher power applications such hybrid or full electric vehicles [1], elevators [2], rail vehicles (for regeneration or free overhead line operation [3]) or stationary grid-connected systems [4]. In all these cases, the energy storage system contains a bi-directional power converter that controls the charging and discharging of the system. The topology of the charging system varies depending on the application, using a DC-DC converter in the event that the batteries are connected to a DC bus (for example in a lift, a car or in a tram), or a DC-AC converter when they are connected to an AC network (for example in grid-connected stationary systems). In all these cases, the converter must be designed to achieve the maximum possible efficiency (in order maximize the use of the stored energy), and in case of mobile applications, the minimum weight and volume. This paper presents the design and development of a DC-DC battery charger intended for mobile applications.

The integration of energy storage systems can be achieved by means of non-isolated [5] and isolated [6] power converters. The buck and boost DC-DC topologies are commonly chosen in high-power applications due to their simplicity, robustness, low cost and high power density [2] [3]. Thanks to the parallel connection of several channels and the interleaving of the modulation signals, a reduction of filter requirements can be reached. An optimal design methodology to find the optimal number of phases of an interleaving boost converter was proposed in [7]. A non-isolated bidirectional DC-DC power converter is designed and developed in the present work, focusing on the topology, gate driver and electromechanical designs.

Wide Band Gap (WBG) materials, such as Silicon Carbide (SiC), enable power electronics devices to operate over existing Silicon device limits in terms of temperature, blocking voltage and switching frequency [8]. Silicon Carbide (SiC) devices are becoming an attractive alternative to Silicon (Si) devices in various niche applications [7] [9] due to the increased availability and maturity of commercial SiC devices and their superior performances (mainly very low switching losses). Consequently, several enhancements can be expected in SiC power converters:

1. Efficiency improvement: The high switching speed of thesedevices decreases the energy losses during thecommutation and therefore improves the efficiency of theconverter.

2. Cooling system volume and weight reduction orsimplification: The high temperature operation of thesedevices and the switching losses decrease enables thereduction of the heatsink size, or at least the simplificationof the cooling system by using natural cooling systemsinstead of forced cooling systems.

3. Magnetic elements volume and weight reduction: Thedesigner can also choose to operate in a higher switchingfrequency instead of reducing losses in order to minimizethe size and volume of magnetic components, such asfilters, inductances or transformers.

The unique switching characteristics of SiC devices involve the use of adapted or even specific Gate Drivers for their optimal control. There is an extensive literature dealing with SiC discrete device Gate Drivers. Most of them are designed to be used with JFET (Junction Field-Effect Transistors), as shown in [10], where an AC-coupled gate circuit for a normally-off JFET is proposed in order to inject a peak current during on-state, or in [11], where a conventional Si driver is adapted in order to control a normally-off JFET with a cascade structure. More complex gate circuits have also been proposed in order to improve the performance of the device as in [12], where an additional Gate Driver circuit is proposed for a SiC JFET and in [13], where a direct drive JFET concept is shown (in both cases replacing the cascade structure). Concerning SiC MOSFET devices (the ones that are currently prevailing in higher power applications mainly due to the lower gate complexity that is required [8]), the literature is much more limited. Respecting commercial Gate Drivers, some

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companies, such as CONCEPT, AVAGO or ROHM, have started offering solutions for SiC discrete devices, mainly based on the adaptation of standard Silicon drivers [14].

There is a clear trade-off between the efficiency and the volume (or the weight) of the converter to be designed. The objective of this paper is to present the design of a 100kW battery charger based on Silicon Carbide technology, optimizing the volume and size of the converter and including a critical part of the converter as the gate driver.

II. CONVERTER DESIGN

A. Battery Charger Requirements Table I shows a summary of the specifications of the battery charger. The charger will be connected to a 540V DC-link (VDC) in order to control the energy flow between the DC-link and the Energy Storage System (ESS) that is connected at the low voltage side (VOUT), as it is presented in Figure 1. The ESS Voltage range goes from 200V to 400V and the output current in the whole voltage range will be ± 250A.

TABLE I SPECIFICATIONS FOR THE PROTOTYPE

Input Voltage VDC 540V

Output Voltage VOUT 200-400V

Output Power POUT 50-100kW

Output current IOUT 250A

Current Ripple ΔIOUT 15A

Figure 1: Battery Charger system.

B. Overall Design In order to comply with the specifications of the Table I, a non-isolated bidirectional DC-DC topology is selected. The power converter operates as a buck converter when it charges the energy storage system and as a boost converter when the energy flows to the dc-link. In order to minimize the volume of the DC-DC converter, a multiple channel interleaved topology has been studied, reducing the output filter requirements.

The output current ripple of the converter is defined with

(1)

where is the duty cycle of the power converter. Depending on the operation point, Figure 3.a illustrates the current ripple of one phase inductor, while Figure 3.b shows the output current ripple in terms of the number of phases of the interleaved DC-DC power converter.

(a) Phase current ripple.

(b) Output ripple for different number of phases.

Figure 3: Output ripple in terms of the duty cycle.

The optimal design, in terms of volume, weight and efficiency, is determined by the selected switching frequency and the number of phases of the converter.

Assuming the specification for the prototype and the selected topology, an interleaved converter with four phases (np) and a 60kHz switching frequency (fsw) has been obtained as a result of the optimization process, presented in Figure 2.

Figure 2: Optimization of converter volume in terms of switching

frequency and number of phases.

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The selected device to develop the converter is the SiC MOSFET CAS100H12AM1 1200V 100A from Cree [15], which fulfills with the required current and voltage. The value of the above designed filter inductor (L), is 45 H, which complies with the current ripple specifications in the worst case (2);

(2)

Figure 3 shows the selected topology for the battery charger and a simulation result presenting the worst case of the output current.

(a) Schematic.

(b) Simulated output current.

Figure 3: DC-DC interleaved Converter.

Thanks to the interleaving, a low current ripple is obtained at the output, respecting the specifications of the prototype.

C. Thermal design MOSFET conduction (Ps_cond) (3) and switching losses (Psw) (4), as well as diode conduction losses (Pd_cond) (5) of the selected device are defined by the next equations,

(3)

(4)

(5)

where Eon and Eoff are the energy losses in the switching on and in the switching off, Rdson is the on-state resistance, VSD is the diode forward voltage, Is_rms is the rms current in the MOSFET, and Id is the current of the diode.

Diode recovery energy (Erec) (6) is negligible;

(6)

Total losses (PDCDC) (7) at the rated output current at each phase (Irms, 62.5A) and at the selected switching frequency (fsw, 60kHz),

(7)

assuming a temperature difference ( T) of 50ºC between the ambient and the heatsink temperature, define the cooling system that is required (LA 17/200 24V) (8);

(8)

D. Electromechanical Design and High Switching Speed The selected high commutation frequency is achievable thanks to the high commutation speed of the SiC device and the corresponding reduction of the switching losses. In addition, regarding the diode recovery losses, the commutation speed does not increase the turn-on energy losses because, unlike the case of Si diodes, the recovery energy of SiC diodes is negligible.

Nevertheless a good electromechanical design is critical because, due to the straight inductances, theses high speed commutations can cause severe over-voltages, which can be dangerous for the semiconductor [16]. Figure 4 shows the DC-link and the electromechanical design.

(a) DC-link bus bar.

(b) Power converter module.

Figure 4: Prototype electromechanical design.

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The straight inductance placed in the commutation loop is mainly formed by the sum of the straight inductances of the SiC module, the bus capacitor, and the bus bar. The straight inductance must be as small as possible, and therefore low straight inductance capacitors are used and a bus bar is designed in order to minimize distances between semiconductors and capacitors. In addition, decoupling capacitors directly connected to semiconductor connections are used in order to reduce the effect of the straight inductances.

In addition, the effect of high voltage derivatives (dV/dt), due also to these fast commutations, must not affect the operation of the converter. These hard voltage changes inject current spikes into the heatsink through the coupling capacitance between the power module and the heatsink. In the design a defined local return path is provided by means of locating capacitors between the bus bar and the ground.

E. Output Inductance Design In order to comply with the current ripple specifications, 45 H inductance with ferromagnetic core is designed for each of the phases. The saturation of the core (9)

(9)

and the inductance value (L) (10) are defined;

(10)

where Bpk is the peak magnetic induction, μ0 is the free space permeability constant and μi is the permeability constant of the magnetic core material, N1 is the number of turns, Idc is the inductance dc current, Ipp is the peak to peak inductance current, lg is the gap distance, lm is the length of the magnetic circuit, and Ac is the magnetic core area.

Assuming a small air gap in inductors, designers can consider the same effective area in the air gap to the magnetic core in order to calculate the total magnetic reluctance, and thus, the inductance and the maximum induction values. However, this assumption is not more valid for inductors with long gaps, due to the fact that the fringing flux cannot be neglected [17]. The fringing flux decreases the total reluctance of the magnetic path, increasing the induction and also the inductance by a factor of F:

(11)

(12)

where the fringing flux factor is expressed with:

(13)

Taking into account those equations, E80/38/20 standards cores are selected, with a 9.5mm gap and 23 turns. The final developed four inductors have about 46.1 H, respecting current ripple requirements.

Moreover, different simulations in Flux are realized in order to select the conductors of the inductances. As it is shown in Figure 5, current density distribution is more homogenous in laminated conductors than in round ones, reducing total winding losses.

(a) Round conductor.

(b) Laminated conductors.

Figure 5: Conductor simulations in FLUX.

F. Gate Driver Design The unique switching characteristics of SiC devices involve the use of adapted or even specific gate drivers for their optimal control. In order to take full advantages of this semiconductor and to protect it, a specific module gate driver is designed and developed. The driver consists of two different parts; the Core Driver (CD) and the Base Driver (BD).

The BD is placed on the module and its main functionalities are; (a) the turn-on and turn-off dynamic control, (b) drain to source voltage monitoring, (c) fiber optic isolation and (d) clamp active protection (in order to assure the integrity of the device in case of over-voltage). The design of the BD varies depending on the electrical and mechanical features of the selected power module (in this case the CAS100H12AM1 1200V 100A Silicon Carbide power module).

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The CD is mainly responsible (e) for over-current protection and (f) short-circuit protection, as well as (g) for galvanic isolation. The design of the CD does not depend on the selected module, except for the isolation requirements, and consequently it can be used with any module of the same voltage isolation requirements. Figure 6 presents a block diagram of the gate driver.

Figure 6: Gate driver block diagram.

Figure 7 shows the electromechanical design of the driver. As it can be seen in this picture, the BD is divided into two parts in order to allow a suitable adaptation of the driver to the converter physical shape: one part (red continuous line) is in direct contact with the module (located over it) and a second part (blue discontinuous line), including the CD and the fiber optic connections, that can be located separately.

Figure 7: Device gate driver electromechanical design.

This separation can be achieved by using 2 different PCB (Printed Circuit Boards) or using a flexible PCB (in this case the first solution has been adopted due to its lower cost).

III. CONVERTER DEVELOPMENT AND EXPERIMENTAL VALIDATION

The designed battery charger has been constructed in order to experimentally test its operation. Figure 8 shows a picture of the developed gate driver, the Core Driver (a) and Board Driver adapted to the module (b). Figure 9 shows a picture of the complete DC-DC interleaved converter.

(a) Core Driver.

(b) Board Driver with the power electronic module.

Figure 8: Developed SiC gate driver.

Figure 9: Developed DC-DC interleaved converter prototype.

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The final dimensions of the developed whole 100kW 60kHz battery charger is 350x300x250mm, including the power module, the output filter inductors, the power supplies and the control cards. The weight of the power converter is 7.5kg, achieving a high power density value (13.3kW/kg) and reaching converter efficiency over 97%.

Firstly, the main functionalities of the Gate Driver have been validated to assure the integrity of the system against over-voltages and over-currents. The following figures show some results of this study.

Figure 10 and Figure 11 present the turn-on and turn-off behavior of the converter. As it can be seen, the device turns-on in less than 100ns and turns-off in less than 80ns, producing low switching losses (total energy losses less than 3mJ at 100A).

Figure 10: Turn-on switching.

Figure 11: Turn-off switching.

In addition, Figure 12 shows that the device is correctly protected against over-voltage conditions, as the voltage clamp active protection acts at approximately 800V. Figure 13 shows that the device is also correctly protected against short-circuit as the protection acts in 4μs, turning-off the device with a soft switching in order to avoid over-voltages.

Figure 12: Clamp active protection.

Figure 13: Short-circuit protection.

The developed interleaved DC-DC power electronic converter has been validated experimentally at nominal rating, as it is shown in Figure 14. A resistance and a 10mF capacitance are used as load in order to emulate de energy storage system.

Figure 14: Experimental electrical results.

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In order to verify the thermal behavior of the battery charger, the different temperatures are measured: heat-sinks (HA and HB), output inductors (L1, L2, L3, and L4) and ambient temperature (Amb). Figure 15 illustrates the experimental temperature evolution at nominal rating.

Figure 15: Experimental temperature evolution.

Analyzing the thermal behavior of the power converter, the expected temperature difference is reached (50ºC) in the hottest point of the heatsink, validating the theoretical calculations.

IV. CONCLUSIONS This paper presents the main design aspects of a 100kW battery charger, based on Silicon Carbide technology and operating at 60kHz. The work is focused on the topology and electromechanical design of the power converter, optimizing the volume and weight of the system and including a critical part of the converter as the gate driver. The power converter has been developed and experimentally validated at nominal rating. The unique switching characteristics of Silicon Carbide semiconductors involve the use of adapted or even specific gate drivers in order to achieve the optimal control of the active devices. A specific module gate driver is designed and developed, including protections as over-voltage and short-circuit. The driver has been experimentally validated with SiC MOSFETs, and some measurements are presented in this paper.

The volume and the weight are one of the most important objectives of the power converter design in transport applications. In the optimization of these cost functions, there is usually a trade-off between the size of passive components and that of the cooling system, which is related to the semiconductor losses. A key design parameter is the operation frequency along with the power converter topology. Regarding the specifications, the optimal design solution is found with a four phase interleaved DC-DC converter operating at 60kHz, achieving a high power density conversion system (13.3kW/kg) and an efficiency over 97%.

ACKNOWLEDGEMENT The authors would like to thank the Basque Government EMAITEK program for its funding.

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