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IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 60, NO. 1, JANUARY 2013 221

Micromachined Passive Bandpass Filters Based onGaAs Monolithic-Microwave-Integrated-Circuit

TechnologyZhiqiang Zhang and Xiaoping Liao, Member, IEEE

AbstractThis paper presents micromachined on-chip RF pas-sive bandpass filters at 18 GHz based on utilizing a three-poleLC low-pass filter and two dc-blocking capacitors, which is ac-complished with a GaAs monolithic-microwave-integrated-circuitprocess. The microwave design model of the bandpass filters thattake into account conductor losses is given and verified. Using thismodel, the RF bandpass filter with a tunable center frequency anda desirable bandwidth can be realized. Due to only a planar spiralinductor required in the design, the layout size of the filter is lessthan 700 m 400 m. Furthermore, in order to minimize the

effect of substrate losses caused by the inductor on the bandpassfilter, metal shores (MSs) and patterned ground shields (PGSs) arelocated and inserted between the inductor and the GaAs substrate,respectively, and a cavity on the backside of the inductor substrateis processed by the via-hole etching technique. Measurement re-sults demonstrate that RF bandpass filters without MS and PGSshow agreement with the design performance and those with MSand PGS have resulted in the improvement of about 24% insertionloss and a slight effect on the center frequency.

Index TermsBandpass filter, GaAs monolithic microwave in-tegrated circuits (ICs) (MMICs), metal shores (MSs), microma-chined, patterned ground shields (PGSs), planar spiral inductor.

I. INTRODUCTION

BANDPASS FILTERS play important roles in RF and

microwave integrated circuits (ICs). In RF applications,

most of commercially available RF bandpass filters like ceramic

filters or surface acoustic wave filters have good performance.

Unfortunately, these filters are bulky and off-chip discrete com-

ponents, which is difficult for the ultimate miniaturization and

integration of the system. At present, on-chip passive bandpass

filters operating in the gigahertz frequency have been exten-

sively researched through using the transmission line theory

[1][3] at the cost of the chip area but seldom reported through

using RF on-chip inductors. On-chip inductors have posed the

primary bottleneck against achieving high-performance RF on-

chip filters, and the quality factor (Q) of RF on-chip inductors

Manuscript received August 25, 2011; revised October 11, 2012; acceptedOctober 29, 2012. Date of current version December 19, 2012. This workwas supported in part by the National Natural Science Foundation of Chinaunder Grants 60976094, 61076108, and 60676043 and in part by the ScientificResearch Foundation of Graduate School of Southeast University under GrantYBPY1207. The review of this paper was arranged by Editor A. M. Ionescu.

The authors are with the Key Laboratory of MEMS of the Min-istry of Education, Southeast University, Nanjing 210096, China (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TED.2012.2228197

is one of the most important parameters and mainly limited by

various losses [4][6], such as substrate losses and metal losses.

In order to reduce the parasitic losses and increase the Q of on-

chip inductors, some RF on-chip inductors have been proposed

based on the MEMS technology [7][9] or fabricated on the

low-loss substrate [10], [11]. In the recent year, two typical

RF passive bandpass filters with on-chip inductors have been

reported based on the micromachining technology. Yook et al.

[12] presented the suspended spiral inductor and the 2.45 GHzbandpass filter on the selectively anodized aluminum. Gu and

Li [13] and Wu et al. [14] developed the solenoid inductors

and the 5.4-GHz bandpass filter based on the post-CMOS

micromachining process. However, these filters have complex

processes or additional processing steps and are not compat-

ible with traditional Si- or GaAs-based RF ICs. Moreover, it

is problematic for designers that a full-wave electromagnetic

(EM) simulation is performed to optimize their design, without

an accurate design model of bandpass filters. Moreover, most

of the on-chip bandpass filters occupy the larger chip area due

to the limit of inductors.

On the other hand, to increase the Q of planar spiral in-

ductors, patterned ground shields (PGSs) applied in the on-chip planar inductors for Si- or GaAs-based RF ICs have been

widely investigated [15][17]. These inductors have good per-

formance in the RF range, but the effect of PGS on the perfor-

mance of bandpass filters hardly has been studied. As for planar

spiral inductors, the bigger the inductance of an RF inductor is,

the more the spiral winding of the inductor is. However, RF on-

chip inductors with the bigger inductance can suffer from the

downward sloping in the outer of the spiral winding or even the

collapse if the sacrificial layer for supporting the spiral winding

is released, which affects the performance of inductors [17].

In addition, an RF on-chip inductor occupies a considerable

space compared with an on-chip metalinsulatormetal (MIM)capacitor. Therefore, the design of an RF on-chip LC bandpassfilter is usually required to use the less number and smaller

inductance of on-chip inductors, so that the bandpass filter

suffers from small parasitic losses caused by the inductors in

the RF range, and increase the integration.

The circuit configuration of the RF on-chip bandpass filter

is designed to consist of a monolithic microwave integrated

circuit (MMIC) inductor and four MIM capacitors [18]. The

microwave design model of the bandpass filter with the conduc-

tor losses is proposed based on the microwave network theory

and verified by the EM simulation and the measurement in

this paper. Using the design model, the bandpass filter with

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222 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 60, NO. 1, JANUARY 2013

Fig. 1. (a) Schematic view with structural parameters and (b) simple-network topology circuit of the bandpass filter.

a tunable center frequency and a desirable bandwidth can be

realized. Due to only utilizing an on-chip spiral inductor with

a smaller inductance and four MIM capacitors in this design,

the bandpass filter greatly reduces the chip area and parasitic

losses in the RF range. Furthermore, the effects of metal shores

(MSs), PGS, and the back cavity on micromachined RF on-chip

LC passive bandpass filters at 18 GHz are presented in thispaper, in order to minimize substrate losses. These bandpass

filters provide the compatible capability with the GaAs MMIC

technology. Measured results show that the effects of MS, PGS,

and the back cavity on the performance of the RF bandpass

filters have resulted in better improvements.

II. DESIGN AND ANALYSIS

A. Lossless Prototype

Passive bandpass filters are generally designed based on the

filter synthesis/insertion-loss method [19], [20]. In the method,

the transformation of a prototype low-pass filter to a bandpass

filter needs not only a large number of the inductors but also

much more inductance value, often reaching tens of nanohen-

ries, so the design of the bandpass filter with on-chip inductors

is compromised. In this paper, a micromachined RF on-chip

passive bandpass filter at 18 GHz is proposed by utilizing a

three-pole LC low-pass filter and two dc-blocking capacitors.Fig. 1 shows a schematic view and a simple -network topologycircuit of the bandpass filter. In Fig. 1(b), ports 1 and 2 are

input and output ports, Z0 is the characteristic impedance of thecoplanar waveguide (CPW) transmission line, C3 and C4 aredc-blocking capacitors, and C1 L1 C2 constitutes a three-pole low-pass filter. It should be noted that the bandpass filter

is designed to only utilize an on-chip spiral inductor with a

smaller inductance, in order to reduce parasitic losses in the RF

range and the chip area.

As shown in Fig. 1(b), the equivalent circuit model of the

bandpass filter is described by a lumped network and divided

into seven parts by the dashed r1 r1, s1 s1, t1 t1, r2 r2, s2 s2, and t2 t2. As for the r1 r1 side, under lossless

conditions, the ABCD matrix normalized of the lumped equiv-alent circuit between port 1 and port r1 can be given as

A BC D

1r1

=

cos j sin

j sin cos

(1)

where is the electrical length of the transmission line. Fora simple analysis, the electrical length can be defined as zero.

Therefore, (1) can be simplified asA BC D

1r1

=

1 0

0 1

. (2)

In Fig. 1(b), according to the microwave theory, the ABCDmatrix normalized of the bandpass filter between ports 1 and 2

can be obtained asA BC D

12

=

A BC D

1r1

A BC D

r1s1

A BC D

s1t1

A BC D

t1t2

A BC D

t2s2

A BC D

s2r2

A BC D

r22

=

1 0

0 1

1 1

jC3Z00 1

1 0

jC1Z0 1

1 jL1Z00 1

1 0j C2Z0 1

11

jC4Z00 1

1 00 1

(3)

where = 2f is the angular frequency and f is the frequency.In order to further simplify the analysis in (3) so that C1 = C2and C3 = C4, the ABCD matrix normalized between ports 1and 2 can be expressed as (4) shown at the bottom of the page.

According to the conversion relationship of the ABCD ma-

trix and the scatting matrix, based on (4), the S-parameters of

the bandpass filter between ports 1 and 2 can be calculated using

the MATHEMATICA software as

S11 = C3(2 2C3L1) 2C21 L1 1 + 2C23 Z20+2C1

1 2C3L1 + 2C23 Z20

/

((jC3 + C1(j + C3Z0)) (2j + C3(jL1 2Z0)

+2C1L1(j + C3Z0)

(5.1)

S21 =

2jC23 Z0

/((jC3 + C1(j + C3Z0))

(2j + C3(jL1 2Z0)+2C1L1(j + C3Z0)

. (5.2)

A BC D

12

= C32C21L1+C1(22C3L1)

C3

j(C1+C3)(2+2C1L1+

2C3L1)

C2

3Z0

jC1(2 + 2C1L1)Z0C3

2C21L1+C1(2

2C3L1)C3

(4)

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ZHANG AND LIAO: MICROMACHINED PASSIVE BANDPASS FILTERS BASED ON MMIC TECHNOLOGY 223

Fig. 2. S-parameters of theoretical results of the RF passive bandpass filter.

Using (5.2), we can calculate the center frequency f0 and3-dB-bandwidth boundary frequencies f3 dB of the band-pass filter by

|S21|2

f

f0

= 0 (6.1)

|S21|2f3 dB

= 12

. (6.2)

This paper presents a 3-GHz RF LC passive bandpass fil-ter with a 3-dB bandwidth of 0.8 GHz. Based on (5.2), ifthe given f0 and f3 dB are substituted into (6.1) and (6.2),respectively, two equations as a function of L1, C1, and C3will be obtained. Because smaller L1 is a primary choice,a set of date (C1 and C3) is solved when the characteristicimpedance Z0 of the transmission line is 50 . In order toshow the design model of the bandpass filter, the S-parameters

of the theoretical results are shown in Fig. 2. Fig. 3 shows the

insertion lossS21

of the bandpass filter for different inductance

L1, different dc-blocking capacitance C3 and C4, and differentshunt capacitance C1 and C2. L1 can be used to design thetunable center frequency of the bandpass filter [see Fig. 3(a)],

while C3 and C4 can be used to achieve a narrow or widebandwidth [see Fig. 3(b)]. C1 and C2 play an important rolein designing the center frequency and the bandwidth of the

bandpass filter [see Fig. 3(c)]. As shown in Fig. 3(b) and (c), the

rolloff rate of the bandpass filter in the stopband region becomes

much steeper as the capacitance C3 and C4 decrease or C1and C2 increase. Therefore, the passive bandpass filter with adesirable center frequency, bandwidth, and stopband rejection

can be realized by using the microwave design model.

B. Loss Modeling

In Fig. 1(a), the losses of the bandpass filter consist of con-

ductor and dielectric losses of the CPW, the capacitors, and the

inductor. Due to the GaAs substrate with a high resistivity, these

dielectric losses are smaller than the conductor losses. Further-

more, the CPW and the capacitors have shorter transmission

dimensions than the inductor. Therefore, in these losses, the

conductor loss of the inductor with a large spiral winding is a

main loss. It is mainly caused by the skin and proximity effects

(i.e., current crowding). Fig. 4 shows an improved -networktopology circuit of the bandpass filter. In Fig. 4, Rs represents

the parasitic series resistance of the inductor and symbolizesthe conductor loss due to the skin and proximity effects. As

Fig. 3. S21 of bandpass filter for different (a) inductance L1, (b) dc-blockingcapacitance C3 and C4, and (c) shunt capacitance C1 and C2.

Fig. 4. Improved-network topology circuit of the bandpass filter.

the proximity effect from the conductor turns farther than the

adjacent turns in the same plane can be neglected, the classical

1-D approximation ofRs in the inductor is given by

Rs =l

w(1 et/)(7)

with

=

f(8)

where is the skin depth, w and t in meters are the width andthe thickness of the inductor, l in meter is the total length ofall line segments in the inductor winding, and = 0r arethe resistivity and the permeability of gold where 0 and r arethe free space and relative permeability, respectively, and f in

hertz is the frequency. It should be noted that Rs increases as decreases with the increase of the frequency.

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TABLE IELECTRICAL AND STRUCTURAL PARAMETERS OF THE BANDPASS FILTER

Similarly, if C1

= C2

and C3

= C4

in Fig. 4, the

S-parameters of the bandpass filter with the conductor losses

between ports 1 and 2 can be calculated as

S11 = ((C1 + C3) (2 + (C1 + C3)(L1 + jRs))2C1C

23 (2 + C1(L1 jRs)) Z

20

/ ((C1C3Z0 j(C1 + C3))

(2j + (C1 + C3)(j L1 Rs)+C3 (2 + C1(L1 jRs)) Z0)) (9.1)

S21 =

2jC23 Z0

/ ((C1C3Z0 j(C1 + C3)) (2j + (C1 + C3)(j L1 Rs)

+C3 (2 + C1(L1 jRs)) Z0)) . (9.2)

Table I shows the electrical and structural parameters of the

bandpass filter. In order to verify the accuracy of the design

model with the conductor losses, the theoretical and simulated

results are compared. The simulation of the bandpass filter is

performed using the EM simulator (HFSS). In the simulation,

the GaAs substrate parameter is set to be 0.006 in dielectric

loss angle [7] and infinite in resistivity. Fig. 5 shows the com-

parison of the theoretical and simulated results in the 3-GHz

bandpass filter. In Fig. 5, the theoretical results calculated based

on the improved -network topology circuit model show goodagreement with the simulated results by HFSS at 18 GHz.

It shows that the S-parameters of the bandpass filter can bedesigned and optimized using the design model with the con-

ductor losses, and thus, the center frequency and the bandwidth

of the filter can be obtained using (6.1) and (6.2). Compared

with the lossless results (see Fig. 2), the theoretical results with

the conductor losses lead to the insertion loss from 0.01 dBto 3.37 dB, which shows that the microwave loss modelcan better predict the performance of the fabricated bandpass

filter. As can be observed in Fig. 5, the theoretical S21 in themodel is slightly bigger than the simulated one in HFSS at

the center frequency, which is probably due to the fact that the

loss of the bandpass filter is overestimated by utilizing the 1-D

approximation ofRs in the improved model.

The RF bandpass filter with an inductor can be regardedas a pretty LC network, with a standard resonant spectrum.

Fig. 5. Comparison S11 and S21 of theoretical and simulated results in thebandpass filter.

Fig. 6. Change of 34 GHz in the center frequency of the bandpass filter withthe widened passband and the small insertion loss.

Therefore, the designed center frequency with a minimal in-

sertion loss in ideal conditions (see Fig. 2) is a point frequency

instead of a frequency spectrum with ripples. In this case, the

bandwidth (selectivity) of the bandpass filter is narrow and apoint frequency. In some applications, the passband (3-dBbandwidth) of the bandpass filter is used to permit the selec-

tivity of the bandpass filter. In this paper, the passband of the

bandpass filter can be designed to be wide by adjusting the

capacitance of the MIM capacitors (see Fig. 3). In order to

show the wide passband and the tunable center frequency, the

change of 34 GHz in the center frequency of the bandpass

filter with the small insertion loss is shown in Fig. 6. Using

this loss model, the change of 34 GHz in the center frequency

is obtained by simply reducing the capacitance C1 and C2 ofthe shunt capacitors, in good agreement with HFSS. In the

bandpass filter with the center frequency of 4 GHz, the widths

of the capacitance C1 and C2 are reduced to be 17 m, thecorresponding C1 and C2 are 1.05 pF, and the other values arethe same as that in Table I.

C. MS and PGS

At 18 GHz, various parasitic effects and numerous loss

mechanisms have a great impact on bandpass filters, particu-

larly on-chip spiral inductors. In order to reduce the effect of

substrate electric and magnetic losses caused by the inductor,

the micromachined RF on-chip passive bandpass filter with MS

and PGS is proposed in this paper. Fig. 7 shows the top and

across-sectional views of micromachined spiral inductors withMS and PGS.

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Fig. 7. Top andacross-sectional views of micromachined spiralinductors withMS and PGS.

MSs indicate that some cubic pillars are orderly located

between the spiral winding and the dielectric layer and play

an important role in supporting the suspended windings (see

Fig. 7). The dielectric layer Si3N4 is used to separate the pillars

and the substrate. The MSs are fabricated on the dielectric

layer, and their top planes are connected to the winding of the

inductors. Due to the spiral winding with the large area andsmaller Youngs modulus in the GaAs MMIC process, most

of the inductors suffer from the collapse after the sacrificial

layer is released using a wet technique, which results in the

performance deterioration. The MSs make the inductors applied

in the filters keep suspending after the release of the sacrificial

layer, which reduces displacement current losses.

PGSs indicate that some side-by-side ribbons are densely

inserted between the GaAs substrate and the spiral windings of

the inductors and connected with the ground lines of the CPW

line (see Fig. 7). The inductors are suspended above the PGS. It

shows that a ground plane underneath the inductors is formed

by the ribbons, in order to prevent the electric and magneticfields of the inductors from penetrating the substrate. The

PGSs are made of gold and fabricated on the GaAs substrate.

The PGSs are equivalent to make the substrate underneath the

inductor a short and eliminate electric and magnetic losses of

the substrate, yet they do not generate the image current and

affect the inductance. In order to prevent the connection of the

inductor windings and the grounded ribbons, a Si3N4 dielectric

layer with the isotropic growth is used to overcover the PGS.

Furthermore, the GaAs substrate underneath all inductor

windings is removed to form the back cavity by the via-hole

etching technology (see Fig. 7). The back-etching cavity is

equivalent to make the substrate an open and eliminates the

energy dissipation. The effects of the MS, the PGS, and the back

etching on the performance of the MMIC inductors have been

reported in [17] and [21], where the quality factor improvement

can be extended to the bandpass filter in this paper for showing

the design validity of the improved filter.

III. MEASUREMENT AND DISCUSSION

In this paper, micromachined RF on-chip LC passive band-pass filters without and with MS and PGS are fabricated by

using the GaAs MMIC process [18], [21]. The performance

of these passive bandpass filters is measured by an Agilent

8719ES network analyzer and a Cascade Microtech GSG probestation. The so-called full port calibration technique is used

Fig. 8. SEM photographs of micromachined RF passive bandpass filters(a) without MS and PGS and (b) with MS and PGS.

for the realization of the measurement. Fig. 8 shows the SEM

photographs of the two kinds of RF passive bandpass filters.

To facilitate the comparison, the same layout dimensions and

electrical parameters of the bandpass filters are taken. In Fig. 8,

the inner diameter of the inductor is optimized to be 60 m,and the total length of the corresponding windings is 2025 m,

in order to compensate the offset of the center frequency causedby the collapse of the inductor without MS and PGS or the

insertion of MS and PGS. Other parameters were given in

Table I. The layout area of the fabricated bandpass filters is less

than 700 m 400 m.The S-parameter comparison of the theoretical, simulated

(HFSS), and measured results of the RF bandpass filter without

and with MS and PGS are shown in Figs. 9 and 10, which

further demonstrate the validity of the microwave design model

of the bandpass filter with the conductor losses. The same

model of Fig. 4 and values of Table I are used for both filter

configurations. In Fig. 9, the height of the suspended inductor

is 0.41 m in the simulated results. The measured insertion andreflection losses of the bandpass filter without MS and PGS are7.4 dB and 4.8 dB at the center frequency of 3.08 GHz (seeFig. 9). Compared with the theoretical and simulated results,

the measured reflection and insertion losses of the bandpass

filter are unsatisfactory but acceptable for applications. This

is probably due to the following reasons. First, the inductor

and capacitors of the bandpass filter are embedded in the 50-CPW and lead to the mismatch of the device system. At the

same distance between the two ground lines, the characteristic

impedance of CPW increases with the decrease of the signal

width (see Fig. 11). The characteristic impedances of the CPW-

based RF input and output ports in the filter are simulated to be

about 50.9 and 51.0 , respectively, with the G/S/G dimensionof 58/100/58 m for the microwave probe measurement, where

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Fig. 9. Measured S-parameters (a) S11 and (b) S21 of the bandpass filterwithout MS and PGS compared with theoretical and simulated results.

Fig. 10. Measured S-parameters (a) S11 and (b) S21 of the bandpass filterwith MS and PGS compared with theoretical and simulated results.

the G and the S correspond to the slot between the ground plane

and the signal line (G) and the signal width (S), respectively.

However, the connecting lines of the MIM capacitors and the

spiral inductor, as well as the two ground lines, constitute new

CPW transmission lines with the G/S/G of 98/20/98 m, andthe simulated characteristic impedances are about 87.1 and

88.2 at the left and right ports, respectively, but their lengthis very short. Second, the conductor losses of the inductor, the

Fig. 11. Characteristic impedances of CPW under different CPW signal linewidths at 3 GHz by HFSS. It should be noted that a constant total CPW widthof 216 m is considered for all the RF signal widths.

Fig. 12. Simulated S-parameters S11 and S21 of the bandpass filter as afunction of the height (h) between the MMIC inductor and the GaAs substrate.

losses of the CPW and the capacitors, and the deviation in

thickness of component parameters arising from the process tol-

erance result in the difference. In Fig. 9(a), we can find that the

reflection loss of the bandpass filter is much bigger at the centerfrequency. It means that the bandpass filter is in the mismatch

case, and more than 33% RF power is reflected back to the input

at the center frequency. Third, the suspended MMIC inductor

with an original height suffers from the collapse after the sac-

rificial layer is released, which results in the increase of the ca-

pacitance between the inductor and the substrate. The resulting

capacitance leads to the device mismatching, i.e., the increase

of the insertion and reflection losses, and the drop of the center

frequency in the bandpass filter (see Fig. 12). Therefore, it is

important to take into account appropriate design margins in the

center frequency by reducing the winding inductance based on

the design model and the simulation tool, which can make themeasured results reach/approximate the design requirement.

In Fig. 10, the measured insertion and reflection losses of the

bandpass filter with MS and PGS are 5.6 dB and 7.5 dBat the center frequency of 3.01 GHz, respectively, while the

simulated insertion and reflection losses are about 3.3 dB and10.1 dB at the center frequency of 3.05 GHz. Compared withthe measurement of the filter without MS and PGS in Fig. 9, the

measured bandpass filter with MS and PGS reduces more than

24% insertion loss at the center frequency, with an improvement

of 2.7 dB in the reflection loss. Because the bandpass filter is

fabricated on the semi-insulator GaAs substrate with a high

resistivity where the inductor suffers smaller substrate losses

and designed to only utilize an on-chip spiral inductor with asmaller inductance for reducing the parasitic losses, the effects

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TABLE IISUMMARY OF THEORETICAL, SIMULATED, AN D MEASURED RESULTS OF THE RF BANDPASS FILTERS

TABLE IIICOMPARISON OF RF PASSIVE BANDPASS FILTERS WITH ON-CHI P INDUCTORS

of the MS and PGS on the performance of the filter offer smaller

improvements. Compared with the HFSS simulation without

MS and PGS in Fig. 9, the simulated insertion and reflection

losses of the filter with MS and PGS have improvements of

about 0.5 and 0.9 dB, respectively. As can be observed from the

simulated and measured results in Figs. 9 and 10, the effects of

MS and PGS on the performance of the bandpass filter lead

to two advantages: reducing insertion losses and decreasingreflection losses. As for the simulation comparison in Figs. 9

and 10, the improvement of the insertion loss in Fig. 10 is

mainly from the decrease of the reflection loss that is caused

by the MS and PGS. In the simulation, the GaAs substrate

parameter is set to be 0.006 in dielectric loss angle [7] and

infinite in resistivity, so the MS and PGS have smaller impact

on the substrate loss of the filter. As for the measurement

comparison from a list of the calculation in Figs. 9 and 10,

the improvement of the insertion loss in Fig. 10 is mainly from

the decrease of the substrate loss and the reflection loss that is

caused by the MS and PGS. Furthermore, in Fig. 10, the effect

of MS and PGS on the center frequency of the bandpass filter

is slight compared with the design (f0 = 3 GHz), with lessthan 0.34% in the measurement. Compared with the simulation

without MS and PGS, in terms of the suspended inductor with a

2-m height in Fig. 12, the simulated insertion loss of the filterwith MS and PGS in Fig. 10 increases by 0.2 dB at the center

frequency, yet the center frequency moves from 3.25 GHz to

the design value (f0 = 3 GHz) and reaches 3.05 GHz. TheMS and PGS between the inductor and the substrate generate

the parasitic capacitance, and the resulting capacitance results

in the change of the center frequency. Therefore, taking intoaccount the effect of the MS and PGS on the center frequency

will make the measured results reach/approximate the design.

The increase of the simulated insertion loss in Fig. 10 compared

with the ideal filter in Fig. 12 is mainly due to the following:

1) The MS and PGS to suppress substrate loss generate an

additional loss, and 2) the GaAs substrate parameter that is

set in the simulation leads to smaller substrate losses. Table II

summarizes the theoretical, simulated, and measured results of

these bandpass filters. Table III shows the comparison of RF

passive bandpass filters with on-chip inductors. In Table III,

these referenced bandpass filters are achieved by using passive

inductors and capacitors based on different fabrication pro-

cesses. As can be seen from selectivity figures, all the bandpassfilters are point-frequency filters.

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IV. CONCLUSION

The design and fabrication of micromachined RF on-chip

LC passive bandpass filters with MS, PGS, and the back cavityare presented in this paper in order to minimize the substrate

losses. These bandpass filters provide the fully compatible ca-

pability with the GaAs MMIC technology. The design model of

the bandpass filters with the conductor losses based on utilizinga three-pole LC low-pass filter and two dc-blocking capacitorsis given by the microwave network theory and verified by

the HFSS simulation and the measurement. Furthermore, the

method of the microwave design model can be used in other

RF devices. Measurements show that the bandpass filter with

MS and PGS has resulted in the improvement of more than

24% insertion loss, with the effect on the center frequency of

less than 0.34%. The insertion loss and the layout size of the

bandpass filters have scalable improvements. These microma-

chined RF on-chip LC passive bandpass filters can be appliedtogether with RF power amplifiers to the modern personal

communication system and the radar system for achieving thefrequency selection, with a good gain of the selected signal.

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Zhiqiang Zhang received the B.S. degree fromHefei University of Technology, Hefei, China, in2006. He is currently working toward the Ph.D. de-gree in the Key Laboratory of MEMS of the Ministryof Education, Southeast University, Nanjing, China.

Xiaoping Liao (M07) received the B.S. and Ph.D.degrees from Southeast University, Nanjing, China,in 1987 and 1998, respectively.

He is currently a Full Professor with the KeyLaboratory of MEMS of the Ministry of Education,Southeast University.

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