IET Power Electronics

Research Article

Design and improvement of a soft switchinghigh step-up boost converter with voltagemultiplier

ISSN 1755-4535Received on 27th February 2017Revised 8th June 2017Accepted on 1st August 2017E-First on 19th September 2017doi: 10.1049/iet-pel.2017.0162www.ietdl.org

Sadegh Heidari Beni1, Sayyed Mohammad Mehdi Mirtalaei1 , Ardavan Kianpour1, Sara AghababaeiBeni21Department of Electrical Engineering, Najafabad Branch, Islamic Azad University, Najafabad, Iran2Department of Technology, Alzahra University, Tehran, Iran

E-mail: mirtalaei@pel.iaun.ac.ir

Abstract: A new soft switching high step-up boost converter is proposed in this study. Coupled inductors technique is usedalong with voltage multiplier in order to provide high-voltage gain and reduce voltage stress across semiconducting elements.Using an active clamp circuit, soft switching condition is provided for both the main and auxiliary switch which reduces theswitching losses and improves the circuit efficiency as well as power density. The reverse recovery problem of diodes isalleviated due to the existence of the leakage inductances of the coupled inductors in series with diodes. The circuitperformance is investigated and theoretical analysis is provided. A 200 W 40 V input to 400 V output laboratory prototype of theproposed converter is implemented, and the experimental results show the proper performance of the proposed converter.

1 IntroductionEfficient utilisation of diverse renewable sources of energy is oneof the biggest challenges in the field of power and energy. Usingpower electronic interfacing circuits is necessary for connectingrenewable sources of energy to the grid. Solar cell systems are themost desirable sources of renewable energy due to the availabilityof the sun light and flexibility in panel installation. In order toimprove the efficiency of photovoltaic (PV) panels and preventingproblems such as shadow effect on panels and mismatch betweenpanels, using modular PV systems are preferred [1, 2]. In suchsystems, in order to extend the input voltage of PV cells andconnecting them to a high-voltage bus, a DC–DC high step-upconverter with high efficiency is used [3]. In addition, theseconverters are being used in many applications from emergencypower systems, fuel cell and power supplies to high-intensitydischarge lamps [4, 5].

In isolated converters, high-voltage gain can be achieved byproper adjustment of the transformers’ turn ratio. However, theefficiency is usually low. Non-isolated converters are commonlyused in low-to-medium power range applications due to theirhigher efficiency, high power density and low price [6]. Achievinghigh-voltage gain is not possible in the basic boost converterbecause of low efficiency, high conduction loss and severe reverserecovery of the output diode. Coupled inductor-based convertersare of the favourable structures between the presented high step-upconverters. In these structures, high-voltage gain can simply beobtained by adjusting the turn ratio of the coupled inductors. Also,the voltage spikes across the main switches of the converter causedby the leakage inductance can be suppressed by incorporating alossless passive clamp circuit [7] or active clamp solution [8, 9].The main drawback of these converters is the high voltage acrossthe output diode which necessitates a snubber circuit or high-voltage diodes [10].

A three-level boost converter is presented in [11]. Comparedwith the basic boost converter, the three-level boost converter canhalve the voltage stress across the semiconducting elements, whichis suitable for high-output voltage applications. Switching lossesand electromagnetic interferences will reduce due to low-voltagestress. However, hard switching performance of thesemiconducting components and the reverse recovery problem ofthe output diode are of the most serious problems of this converter.In [12], a boost converter integrated with a fly-back converter is

presented for obtaining high step-up voltage gain. In this structure,the boost converter acts as a passive clamp circuit and recycles theleakage inductance energy.

In [13], a single switch high step-up converter based on thecoupled inductors is introduced. In this converter, the energy storedin the leakage inductance of the coupled inductors is recycled bypassive clamp circuit which reduces the voltage spike across switchand diodes. As a result, switches with small on-resistance can beemployed which contributes to efficiency improvement. However,this converter has a complex structure. A coupled inductor-basedinterleaved boost converter is introduced in [14]. High step-upvoltage gain is resulted by using voltage multiplier and coupledinductors. The input current ripple is reduced because of usingcross-coupled inductors as well as the interleaved structure. Thanksto using winding cross-coupled inductors and interleaved structure,the input current ripple is reduced in this converter. Also, theleakage inductance energy is recycled and soft switching conditionis provided for switches using active clamp circuit. High-voltagestress of the output diodes is of the shortcomings of this converterwhich make the use of high-voltage diodes indispensable. A boostconverter based on coupled inductors is presented in [15]. Theclamp capacitor is placed at the charging path of the outputcapacitor which not only absorbs the leakage energy, but alsoincreases the voltage gain.

2 The proposed soft switching high step-upboost converterBased on the discussions mentioned in the Introduction, it can beunderstood that in a solar system, a switching converter is neededwhich, in addition to high efficiency, has low-voltage stress acrossits switches so that high-quality low-priced MOSFETs could beutilised. To achieve these purposes, the proposed high gainconverter is introduced in this section. At first, the proposedconverter is presented and its operating principles are explained.Then, the converter characteristics and the design procedure areinvestigated, after which simulation and experimental results areshown. Finally, a thorough conclusion of the mentioneddiscussions is provided. Fig. 1 shows the proposed converterschematic. This converter is made of using three techniques:voltage multiplier circuit, coupled inductors and active clamp forcreating soft switching condition. Due to high-voltage gain, the

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voltage stress across switches is low. In coupled inductor-basedconverters, high-voltage gain can also be attained by adjusting theturn ratio of the coupled inductors, but the leakage inductance isproblematic in these converters. Active clamp technique can beutilised to solve the leakage inductance problems. The leakageinductance of the coupled inductors can provide soft switchingcondition for the converter. The leakage inductance can also makesdiodes to turn off softly so the reverse recovery problem would beremoved. The proposed converter is composed of two switches S1and S2, three coupled inductors L1, L2 and L3, four diodes D1, D2,D3 and D4, four output capacitors Co1, Co2, Co3 and Co4, and aclamp capacitor Cc, as shown in Fig. 1. The anti-parallel diodes ofswitches are also considered. The switching performance ofswitches is in an interleaving manner. The converter is controlledby two pulsed with duty cycles larger than 0.5° and 180° phaseshift.

In order to explain the operating modes of the proposedconverter, the coupled inductors are modelled which can be seen inFig. 2. Each switching cycle consists of five operating modes. Thekey waveforms of the proposed converter are illustrated in Fig. 3,and the equivalent circuit of the proposed converter in eachoperating mode is shown in Figs. 4 and 5. The followingassumptions are made to explain the operating modes of theproposed converter:

(1) The capacitors are considered ideal. Also, Co1 = Co2 = Co3 = Co4.(2) The coupled inductors are considered as shown in Fig. 2, andtheir other associated parasitic elements are omitted.(3) All switches and diodes are ideal and the voltage drop acrossthem is neglected.

Mode 1 (t0, t1): In this mode, switch S1 and diodes D2 and D4are conducting. Switch S2 is turned off, and diode D1 is off due toreverse voltage VCo1 + nVin across it. Magnetising inductance Lmand leakage inductance Llk are charged by the input voltage source.During this mode, D3 is not conducting because of reverse voltageVCo3 + VCo4 across it. The output capacitors Co2 and Co4 arecharged by the secondary windings of the coupled inductors, andCo1 and Co3 provide the load required energy. This mode endswhen S1 is turned off. The equivalent circuit during this mode isshown in Fig. 4. The important equations in this mode are asfollows:

ILm(t) = ILm(t0) +VCo4

nLm(t − t0) (1)

ILlk(t) = ILlk(t0) +nVin − VCo4

nLlk(t − t0) (2)

Mode 2 (t1, t2): At t = t1 switch S1 turns off, and the anti-paralleldiode DS2 starts to conduct because of resonance between Llk andparallel capacitor of S2, so S2 can be turned on under the zerovoltage switching (ZVS) condition. Also, the voltage across S1becomes limited to VCc. D2 is still conducting, but D1 is in off statebecause of the reverse voltage VCo1 + VCo2 − VCc. Also, voltage(VCo4/n) is applied across the magnetising inductance and makes itscurrent to be increased. Also, since D1 is not conducting, thevoltage across the leakage inductance is reversed and the largevoltage −VCc − (VCo4/n) + Vin is applied across it which make n12and n13 windings current to promptly reduce. Moreover, D3 is stilloff due to the reverse voltage VCo3 + VCo4. At the end of this mode,the leakage inductance current Ilk equals with the magnetisinginductance current ILm, which in turn makes the currents flowingthrough n12 and n13 windings to become zero, and D2 along withD4 turn off at zero current. In this mode, the leakage inductancecurrent equation is as follows:

ILlk(t) = ILlk(t1) −VCc + (VCo4/n) − Vin

nLlk(t − t1) (3)

Mode 3 (t2, t3): D3 turns on at the start of this mode, and thereverse voltage VCo1 + VCo2 − VCc is applied across D2 and thereverse voltage VCo3 + VCo4 is applied across D4. Also, due to theshift in n12 winding voltage polarity, D1 starts to conduct. In thismode, the magnetising inductance voltage is reversed and bothmagnetising and leakage inductance currents reduce. The outputcapacitors Co1 and Co3 are charged via the secondary windings ofthe coupled inductors, and the load is supplied by the outputcapacitors Co2 and Co4. At the end of this mode, as n12 currentreaches to ILlk, the anti-parallel diode DS2 turns off at zero current.The important equations in this mode are as follows:

ILm(t) = ILm(t2) −VCo3

nLm(t − t2) (4)

ILlk(t) = ILlk(t2) −nVCc − nVin − VCo3

nLlk(t − t2) (5)

Mode 4 (t3, t4): DS2 is turned off at zero current at the beginning ofthis mode, and clamp switch S2 along with D1 and D3 isconducting. Also, due to the reverse voltage VCo3 + VCo4 across D4,

Fig. 1  The proposed soft switching high step-up boost converter

Fig. 2  The equivalent schematic of the proposed converter

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and reverse voltage VCo1 + VCo2 − VCc across D2, these diodes arestill reverse biased. The negative voltage −(VCo3/n) is appliedacross the magnetising inductance during this mode, as a result, Lmis being discharged. Also, the negative voltage across the leakageinductance Llk is −VCo1 + (1 + (1/n))VCo3 + Vin, so Llk current isalso being decreased. The leakage inductance current becomes zeroat the end of this mode and increases at the negative direction. Inthis mode, the leakage inductance current equation is as follows:

ILlk(t) = ILlk(t3) −nVCo1 − (1 + n)VCo3 − nVin

nLlk(t − t3) (6)

Mode 5 (t4, t5): S2 is turned off at the start of this mode, and theleakage inductance total current makes DS1 to conduct. In thissituation, S1 can be turned on in a ZVS manner. The reversevoltage VCo1 − VCo3 is applied across the output diode D1 and makeit off. Also, the large voltage Vin + (VCo3/n) is applied across theprimary leakage inductance and its current increases with a sharpslope and the output diode D3 current is reduced fast. Also, due tothe reverse voltages VCo3 + VCo4 and VCo2 + VCo3 − VCc which areapplied across D4 and D2, respectively, these diodes are stillreverse biased. The output capacitor Co3 is charged by thesecondary winding of the coupled inductors, n13, and the outputcapacitors Co1, Co2 and Co4 provide the required load energy. Asthe leakage inductance current ILlk reaches to the magnetisinginductance current ILm, the output diode D3 turns off under the zerocurrent switching (ZCS) condition and this mode ends. The leakageinductance current equation in this mode is as follows:

ILlk(t) = ILlk(t3) +nVin + VCo3

nLlk(t − t3) (7)

3 Steady-state analysisIn this section, specifications of the proposed converter are stated,and its advantages and disadvantages are investigated.

3.1 Voltage gain of the proposed converter

When S1 is turned on (mode 1), the output capacitor Co4 is chargedby n13 winding. Also, the output capacitor Co2 is charged by n12winding and capacitor Cc. So, Co2 and Co4 voltages can beexpressed as follows:

VCo4 ≅ nVin (8)

VCo2 ≅ nVin + VCc (9)

where n is the turn ratio of the coupled inductors which is definedas follows:

n = n13

n11= n12

n11(10)

Applying voltage-second balance to magnetising inductance Lmwould result in voltage equations of Co1 and Cc as follows:

VCo1 = 1 + nD1 − D ⋅ Vin (11)

VCc = Vin1 − D (12)

During the third and fourth modes, the output diode D4 isconducting and Co3 is charged by n13 winding. So, by writingKirchhoff's Voltage Law (KVL) in the output loop and using aboveequations, the capacitor Co3 voltage equation is calculated asfollows:

VCo3 = n(VCc − Vin) = nD1 − D ⋅ Vin (13)

The output voltage of the proposed converter is the summation ofCo1, Co2, Co3 and Co4 voltages. So, the output voltage and thevoltage gain of the proposed converter can be obtained as follows:

Vo = 2(1 + n)1 − D Vin (14)

Gv = VoVin

= 2(1 + n)1 − D (15)

In Fig. 6, the voltage gain of the proposed converter againstdifferent duty cycles is shown. The turn ratio of the coupledinductors is considered to be 2. As can be seen, the proposedconverter can provide large voltage gain with moderate dutycycles, so extreme duty cycles can be prevented.

3.2 Voltage stress of stress across switches and diodes

Regarding equations obtained for capacitor voltages, the voltagestress across switches and diodes is achieved as follows.

The voltage stress across switch S and diode D1

VS = VD1 = VCc = Vin1 − D (16)

The voltage stress across D2 and D3

VD2 = VD3 = n + 11 − D ⋅ Vin (17)

The voltage stress across D4 and D5

VD4 = VD5 = VCo3 + VCo4 = n1 − D ⋅ Vin (18)

Fig. 3  The key waveforms of the proposed converter

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In Fig. 7, the voltage stress variations of the main switch againstdifferent duty cycles are plotted. As can be observed, the voltagestress across the main switch is low even.

3.3 Condition of ZVS realisation

In order to realise the ZVS condition, the leakage inductanceshould satisfy the following condition (it is assumed Cs = Cs1 = Cs2):

Fig. 4  The equivalent circuit of the proposed converter in each operating mode(a) Mode 1, (b) Mode 2, (c) Mode 3

Fig. 5  The equivalent circuit of the proposed converter in each operating mode(a) Mode 4, (b) Mode 5

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Llk ≥VCc

2 ⋅ CsVo ⋅ Io

η ⋅ Vi ⋅ D + 12 ⋅ Δi

2 (19)

in which Δi is defined as follows:

Δi = Vi ⋅ Lm ⋅ D ⋅ T(Lm + Llk)2 (20)

To design the snubber capacitor, the following equation can beused which is the basic equation for designing the snubbercapacitor in power electric circuits:

Cs > iswtf2 ⋅ k ⋅ vsw

(21)

where tf is the voltage across the switch in off time, Vsw is theswitch current before turn-off instant, isw is the guaranteecoefficient for creating soft switching condition, which is

considered about 0.2. k is the switch current deterioration time. Cs1and Cs2 are designed based on (21).

3.4 Performance comparison

The properties of the proposed converter are compared with thebasic boost converter and some recent high step-up converters. Theresults are shown in Table 1. As can be seen, the proposedconverter has the highest voltage gain among all the comparedconverters, so high duty ratios can be prevented. The voltage stressacross the main switch of the proposed converter is the leastbetween other converters; as a result, high-quality low-voltageswitches can be employed. Also, the voltage across the outputdiode is lower than those of [1–3], so the reverse recovery problemhas been improved in the proposed converter. There is anadditional switch in the proposed converter compared with thebasic boost converter and the converter presented in [1], but softswitching condition is created for the main switch which improvescircuit efficiency and power density of the proposed converter. Itcan be concluded that compared with the presented converters in[6, 8–10], the proposed converter is a better option to be used in therenewable energy-based high-voltage gain application.

4 Experimental results of the proposed converterIn order to prove the discussions mentioned in the previous section,the proposed converter is implemented as a laboratory prototype.The input voltage is 40 V and the output voltage is 400 V for 200 W output power. The switching frequency is selected as 50 kHz.Other specifications of the implemented converter are reported inTable 2. The experimental results are depicted in Figs. 8 and 9. Thevoltage and current waveforms of the main switch are shown inFig. 8a. As can be seen, the voltage spikes across switch caused bythe leakage inductance of the coupled inductors are well absorbedand the voltage stress across switch is clamped at almost a quarterof the output voltage. Also, as it is shown, at turn-on instance, theswitch current is negative which shows the current flow throughthe switch anti-parallel diode and proves the ZVS operation of theswitch. As it is shown in Fig. 8b, one can see that the auxiliary

switch performs under the ZVS condition because the negativeswitch current shows that the switch anti-parallel diode isconducting. Also, the voltage stress across the auxiliary switch is aquarter of the output voltage. As a result, low-voltage switcheswith small on-state resistance could be selected for both the mainand auxiliary switch. The voltage and current waveforms of diodesD1 and D2 are shown in Figs. 8c and d, respectively. The soft turn-on of D1 can be observed, and ZCS turn-off of D2 is clear whichalleviates its associated reverse recovery problem. Also, thevoltage stress across D2 is half of the output voltage. The voltageand current waveforms of diodes D3 and D4 are shown in Figs. 9a)and b, respectively. The voltage stress across these diodes is aquarter of the output voltage which helps improve their associatedreverse recovery problem. It can be seen that due to the existenceof the leakage inductance of the coupled inductors in series withdiodes, currents flowing through diodes decrease softly, and thereverse recovery problem of these diodes is alleviated. The

Fig. 6  The voltage gain variations of the proposed converter againstdifferent duty cycles

Fig. 7  The voltage stress across the main switch against different dutycycles

Table 1 Performance comparison of the proposed converter with the basic boost converter and converters presented in [6, 8–10]

Conventional boostconverter

Converterpresented in [6]

Converterpresented in [9]

Converterpresented in [10]

Converterpresented in [8]

The proposedconverter

voltage gain 11 − D

2 + n + nD1 − D

1 + 2n − nD1 − D

1 + 2n1 − D

2 1 + n1 − D

2 1 + n1 − D

stress voltage onswitches

Vo Vo2 + n + nD

Vo1 + 2n − nD

VE1 + 2n

Vo2 1 + n

Vo2 1 + n

stress voltage onoutput diode

11 − DVin

1 + n1 − DVin

1 + n1 − DVin

1 + n1 − DVin

1 + n1 − DVin

n1 − DVin

number of switches 2 1 2 2 2 2number of diodes 1 4 3 4 4 4soft switching no no yes yes yes yes

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proposed converter dynamic response of the proposed converter istested under a step load change condition (full load to no-loadtransient condition) and its result is provided in Fig. 10. Theefficiency of the proposed converter for different output powers isplotted in Fig. 11. The maximum efficiency achieved is 97% at80% of the nominal load. As the output power increases, theconduction losses of elements increase and the efficiency drops.

5 ConclusionIn this paper, a soft switching high step-up boost converter ispresented and its operating principles are explained. Then theproperties of the proposed converter are investigated and the mainequations are extracted. To verify the exactness of theoreticalanalysis, a laboratory prototype of the proposed converter is

Fig. 8  The experimental waveforms of the implemented converter(a) VDS1, Is1, (b) VDS2, Is2, (c) VD1, ID1, (d) VD2, ID2

Fig. 9  The experimental waveforms of the implemented converter (time scale 5us/div)(a) VD3, ID3, (b) VD4, ID4

Fig. 10  Dynamic response of the proposed converter (time scale 25ms/div)(a) AC Component of the output voltage and output current under step load change condition, (b) output voltage and output current under step load change condition

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implemented and tested. The results show that the proposedconverter can successfully provide large voltage gain and low-voltage stress across the main switch. ZVS turn-on and turn-offcondition is provided for the main switch, and the auxiliary switchperforms under the ZVS condition, too. Reverse recovery problemof diodes is alleviated via the leakage inductance of the coupledinductors. The experimental results prove the proper performanceof the proposed converter. So, the proposed converter has high

efficiency and good performance and is suitable to be used inapplications based on renewable sources of energy.

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[3] Tofoli, F.L., Pereira, D.d.C., Paula, W.J.d., et al.: ‘Survey on non-isolatedhigh-voltage step-up dc-dc topologies based on the boost converter’, IETPower Electron., 2015, 8, pp. 2044–2057

[4] Seong, H.W., Kim, H.S., Park, K.B., et al.: ‘High step-up DC-DC convertersusing zero-voltage switching boost integration technique and light-loadfrequency modulation control’, IEEE Trans. Power Electron., 2012, 27, pp.1383–1400

[5] Li, W., He, X.: ‘Review of nonisolated high-step-up DC/DC converters inphotovoltaic grid-Connected applications’, IEEE Trans. Ind. Electron., 2011,58, pp. 1239–1250

[6] Ajami, A., Ardi, H., Farakhor, A.: ‘A novel high step-up DC/DC converterbased on integrating coupled inductor and switched-capacitor techniques forrenewable energy applications’, IEEE Trans. Power Electron., 2015, 30, pp.4255–4263

[7] Hu, Y., Deng, Y., Long, J., et al.: ‘High step-up passive absorption circuitused in non-isolated high step-up converter’, IET Power Electron., 2014, 7,pp. 1945–1953

[8] Molavi, N., Adib, E., Farzanehfard, H.: ‘Soft-switched non-isolated high step-up DC-DC converter with reduced voltage stress’, IET Power Electron., 2016,9, pp. 1711–1718

[9] Sathyan, S., Suryawanshi, H.M., Ballal, M.S., et al.: ‘Soft-switching DC-DCconverter for distributed energy sources with high step-up voltage capability’,IEEE Trans. Ind. Electron., 2015, 62, pp. 7039–7050

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Fig. 11  The efficiency of the proposed converter for different outputpowers

Table 2 Specifications of the laboratory prototype of theproposed converterSymbol Parameter ValueP output power 200 WVin input voltage 40 VVo output voltage 400 VFS1 main switch switching frequency 50 kHzFS2 auxiliary switch switching frequency 50 kHzLm magnetising inductance 100 μHLlk leakage inductance 5 μHN turn ratio of the coupled inductors 1S1, S2 MOSFET switches IRFP260D1–D4 diode MUR460Cc capacitor 2 μFCo1–Co4 output capacitors 47 μF

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