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PHOTONIC INTEGRATED WIRELESS RECONFIGURABLE OPTICAL ADD-DROP
MULTIPLEXER (WROADM) FOR 5G COMMUNICATIONS
by
Haris Khan Niazi
____________________________ Copyright © Haris Khan Niazi 2019
A Master's Report Submitted to the Faculty of the
JAMES C. WYANT COLLEGE OF OPTICAL SCIENCES
In Partial Fulfillment of the Requirements
For the Degree of
MASTER OF SCIENCE
In the Graduate College
THE UNIVERSITY OF ARIZONA
2019
2
ACKNOWLEDGEMENTS
I would like to express my deepest gratitude to my advisor, Dr. Dan Kilper for inviting me into
his research group and introducing me to a field that I grew to love, despite all its difficulties.
Mentoring me and providing me with my own project has allowed me to grow and acquire skills
I did not think possible merely a year ago.
I also extend my thanks to Dr. Norwood who provided me with access to the Lumerical
software, without which this report would not be possible and for agreeing to be part of the
committee on such short notice, as well as Professor Stefan Preble at the Rochester Institute of
Technology for teaching me most, if not all, of what I know about the Lumerical software and
photonic chip design. I would also like to express my gratitude to Dr. Peyghambarian for kindly
taking out the time to be a part of my committee. I sincerely appreciate it.
I wish to thank my fiancée for her continued love and support, no matter the hour, and for
forgiving me my endless distractions with work. Lastly, but also most importantly, all of this
would not be possible without my parents and family, who gave away everything just so I could
be here. This milestone goes out to you folks.
3
Contents 1. Abstract ................................................................................................................................................. 7
2. Introduction ........................................................................................................................................... 8
3. mm-Wave Generation Techniques ...................................................................................................... 13
4. PIC WROADM Core Technologies Overview ................................................................................... 19
4.1. Orthogonal Frequency Division Multiplexing (OFDM) ............................................................. 21
4.2. Radio-over-Fiber (RoF) .............................................................................................................. 23
4.3. PIC WROADM Block Diagram ................................................................................................. 24
5. Simulation Setup and Results ............................................................................................................. 26
5.1. Finite-Difference Time-Domain (FDTD) ................................................................................... 27
5.2. S-Parameters and Compact Model Creation ............................................................................... 28
5.3. Waveguide Design ...................................................................................................................... 31
5.4. Y-Branch (Splitter/Combiner) Design ........................................................................................ 34
5.5. Mach-Zehnder Modulator (MZM) Design.................................................................................. 36
5.6. NRZ-OOK Transceiver Simulation ............................................................................................ 44
5.7. Filtering Stage Design ................................................................................................................. 46
5.8. System Eye Diagrams ................................................................................................................. 53
6. DRC Clean Chip Layout ..................................................................................................................... 63
7. Conclusion .......................................................................................................................................... 69
8. References ........................................................................................................................................... 71
4
List of Figures
Figure 2.1: Expected Yearly Data Traffic Projections [1] .............................................................. 8
Figure 2.2: Channel Transmission Capacity Trends [4] ................................................................. 8
Figure 2.3: 5G Primary Objectives [8] ......................................................................................... 10
Figure 2.4: Attenuation in the mm-Wave Regime [9] .................................................................. 11
Figure 2.5: Fronthaul Network Comprising PIC mm-Wave WROADMs ................................... 12
Figure 3.1: mm-Wave Generation by Filtering Spectra of Directly Modulated DFB Laser [19] 13
Figure 3.2: mm-Wave Generation using DWFL and PCF [22] .................................................... 14
Figure 3.3: DSB Modulation (Top); SSB Modulation (Middle); OCS Modulation (Bottom) ..... 15
Figure 3.4: Arbitrary mm-Wave Generation using MLL and Programmable Filter [17] ............. 17
Figure 3.5: OFC Generation using Gain Switched DFB(Left) and Resulting OFC Spectrum
(Right) [25] ................................................................................................................................... 18
Figure 4.1: Block Diagram of Traditional ROADM Node [28] ................................................... 19
Figure 4.2: Functional Overview of PIC WROADM ................................................................... 20
Figure 4.3: Frequency Division Multiplexing (Left) and Orthogonal Frequency Division
Multiplexing (Right) ..................................................................................................................... 22
Figure 4.4: Conceptual Diagram of OFDM Generation/Detection [29] ....................................... 22
Figure 4.5: Basic RoF Link [30] ................................................................................................... 23
Figure 4.6: PIC WROADM Block Diagram................................................................................. 24
Figure 4.7: RF Carrier Switching Frequency Space Diagram ...................................................... 25
Figure 5.1: Lumerical Products Inter-operability [31] .................................................................. 26
Figure 5.2: Response of a Linear Device [34] .............................................................................. 29
Figure 5.3: S-Matrix of a Two-Port Device .................................................................................. 29
Figure 5.4: Three-Port Y Splitter .................................................................................................. 30
Figure 5.5: Geometry of Strip Waveguide with Cladding ............................................................ 31
Figure 5.6: Effective Index and Multimode Behavior vs. Silicon Waveguide Thickness and
Width [35] ..................................................................................................................................... 32
Figure 5.7: (a) Cross-Section of Simulation Region with Silicon Waveguide Surrounded by SiO2
Cladding (b) Mesh with Finer Resolution Closer to Silicon Waveguide (c) Spatial Profile of TE
Mode (d) Spatial Profile of TM Mode (e) Group Index vs. Wavelength of Operation ................ 33
5
Figure 5.8: Y-Splitter Port Definition and Input Pulse Settings ................................................... 34
Figure 5.9: Optical Network Analyzer Setup and Frequency Response of Y-Splitter ................ 35
Figure 5.10: Injection Based Modulator (Left); Depletion Based Modulator (Right) ................. 37
Figure 5.11: Undoped Rib Waveguide Geometry (Left); Effective Index vs. Wavelength for
Undoped Rib Waveguide (Right) ................................................................................................. 38
Figure 5.12: Silicon PN Phase Shifter in Lumerical DEVICE (Left); PN Phase Shifter Cross-
Section(Right) ............................................................................................................................... 39
Figure 5.13: (a)-(b) N=P=1E17 cm-3
Depletion Region Widths at Forward and Reverse Bias; (c)-
(d) N=P=1E18 cm-3
Depletion Region Widths at Forward and Reverse Bias; ............................. 40
Figure 5.14: Change in Effective Index vs. Voltage for 1E17 cm-3
and 1E18 cm-3
Doping
Concentrations .............................................................................................................................. 41
Figure 5.15: Lumerical INTERCONNECT Setup for MZM Incorporating N = P =1E18 cm-3
Silicon Phase Shifters ................................................................................................................... 42
Figure 5.16: (a) Slice of MZM Transmission Spectrum at 1552.71 nm; (b) Slice of MZM
Transmission Spectrum at 1553.39 nm ......................................................................................... 43
Figure 5.17: Silicon Based PN Phase Shifter as a Voltage Dependent RC Circuit ...................... 44
Figure 5.18: MZM Based NRZ-OOK Transceiver Simulation Setup .......................................... 45
Figure 5.19: Eye Diagram at 25 Gbps .......................................................................................... 45
Figure 5.20: Eye Diagram at 10 Gbps .......................................................................................... 46
Figure 5.21: (Left) All-Pass Ring Resonator; (Right) Add-Drop Ring Resonator [39] ............... 47
Figure 5.22: Transmission Spectra of All-Pass (Red) and Add-Drop Ring(Blue) Resonators [39]
....................................................................................................................................................... 48
Figure 5.23: (Left) A Parallel Coupled Ring Resonator Filter; (Right) Transmission (dB) versus
Frequency for Parallel Coupled Ring Resonator Filter of Order 2 to 5 and FSR/Bandwidth = 10
[40] ................................................................................................................................................ 49
Figure 5.24: AIM PDK C+L Band Tunable Silicon Microring Resonator Filter [41] ................. 50
Figure 5.25: Cascaded AIM PDK MRR Filter Transmission Measurement Setup ...................... 50
Figure 5.26: Drop Port Spectrum for 3 Ring Cascaded MRRs..................................................... 51
Figure 5.27: INTERCONNECT Simulation Setup for Primary and Secondary 3 Ring Cascaded
MRR Filter Transmission Spectrum Measurement ...................................................................... 52
Figure 5.28: Transmission (dB) versus Frequency Plot for Two Stage Cascaded MRR Filter .... 53
6
Figure 5.29: AIM PDK Digital C+L Band MZM (50G) [41]....................................................... 54
Figure 5.30: AIM PDK MZM Transmission (dB) Spectrum versus Phase Shifter Voltage for
1552.71 nm and 1553.39 nm......................................................................................................... 55
Figure 5.31: (a) Simulation Setup for AIM PDK MZM operated at 10/25 Gbps NRZ-OOK; (b)
PRBS Output after NRZ Pulse Shaper; (c) Modulated Output after MZM; (d) OSA Spectrum for
25 Gbps NRZ-OOK Modulation; (e) Eye Diagram for 25 Gbps NRZ-OOK; (f) Eye Diagram for
10 Gbps NRZ-OOK ...................................................................................................................... 57
Figure 5.32: PIC WROADM System Layout in Lumerical INTERCONNECT .......................... 59
Figure 5.33: (a)-(b) Modulated Output Signals; (c)-(d) Log(BER) vs. Power (dBm) Plots Using
Signals Directly from Modulator Outputs; (e)-(f) Eye Diagrams at 0 dB Attenuation ................ 60
Figure 5.34: (a) 14 GHz Signal with 1553.39 nm Optical Carrier; (b) 28 GHz Signal with
1552.71 nm Optical Carrier; (c) Combined Signal Plus New Carrier; (d) Filtered Signal with
Additional Harmonic Term; (e) RF Spectrum Analyzer Output Showing Switched Carriers ..... 62
Figure 6.1: Chip Floorplan Showing Edge Couplers and Dicing Channels ................................. 64
Figure 6.2: AIM Waveguide, Etch and Doping Layers ................................................................ 65
Figure 6.3: AIM Metal Layers ...................................................................................................... 66
Figure 6.4: PIC WROADM DRC Clean Chip .............................................................................. 67
List of Tables
Table 1: Carrier Switching Loss Budget ....................................................................................... 67
Table 2: Optical Add Loss Budget................................................................................................ 68
Table 3: Optical Drop Loss Budget .............................................................................................. 68
7
1. Abstract
Millimeter waves are a key technology being considered for 5G communications by offering a
new swath of spectrum to meet growing bandwidth requirements, and due to their attenuation in
weather conditions such as rain/fog as well as by foliage and buildings, they rely on a dense
proliferation of small cells to transmit data from one point to another. These small cells need to
be compact, low energy consumption and cost effective in order to be scalable and hence
photonic integrated circuits are a prime candidate for these devices. This report proposes a
Wireless Reconfigurable Optical Add Drop Multiplexer (WROADM) that is a photonic
integrated circuit approximately 1.75 mm2 in size, and that is capable of accepting mm-Wave
signals and routing them across a chip modulated on optical carriers, adding/dropping signals
to/from fiber, as well as performing optical signal processing e.g. filtering in the optical domain
to switch the RF carriers at the output. It achieves this via using silicon Mach-Zehnder
Modulators with extinction ratios (ER) of -17 dB and -20 dB and product of 0.647 V•cm,
as well as a tunable microring resonator filter with a 3-dB bandwidth of 65 GHz. The design of
the WROADM is outlined starting with waveguide geometry FDTD simulations and culminating
in overall system S-matrix based simulation to observe the carrier switching, followed by DRC
(Design Rule Checking) clean chip layout and definition of an optical loss budget that varies
from 10.85 dB to 17 dB depending on whether the add, drop or carrier switching function route
is taken by the optical signal.
8
2. Introduction
Internet traffic is projected to increase by a factor of two every two years (equating to
approximately 40% growth per year) [1], as can be seen in Figure 2.1. This increase is heralded
by high fidelity content delivery services e.g. high definition video streaming, P2P file sharing,
Fog computing which takes storage and compute applications to the edge of a network [2], as
well as latency sensitive applications such as e-commerce and other financial services. Similarly,
cloud gaming is on the rise, as is evident from the recent ambitious announcement of Google's
Project Stadia [3], which promises widespread accessibility of games streamed directly via users'
web browsers, together with seamless transition to handheld devices while on the move.
Consequently, the demand for greater bandwidth in communication systems continues to
increase at an alarming rate, with capacities of 100 Tbps projected to be required by 2020 [4].
Figure 2.2 visually summarizes the trends and breakthroughs that have led to increases in
transmission capacity over the years [4]. The increase from the single-channel days of time
division multiplexing with capacities of a few gigabits-per-second to hundreds of gigabits-per-
second was brought on by the advent of Wavelength Division Multiplexing (WDM) whereby
Figure 2.2: Channel Transmission Capacity Trends [4] Figure 2.1: Expected Yearly Data Traffic Projections [1]
9
densely spaced multiple wavelength channels could be simultaneously transmitted over the same
link. This was supplemented by low loss Silica fiber (attenuation of approximately 0.2 dB/km),
as well as the development of Erbium Doped Fiber Amplifiers (EDFAs) which provided
wideband amplification within the low-loss transmission window. WDM has singlehandedly
accounted for a thousand fold increase in transmission capacity over a decade [5]. More recently,
the advent of coherent communications (whereby carrier phase and amplitude are both encoded
with information) has further increased transmission capacity by making use of high speed
Digital Signal Processing (DSP) in order to compensate for non-linear transmission effects e.g.
Polarization Mode Dispersion (PMD). Coherent communication, in tandem with advanced
modulation formats used in "superchannels" where closely spaced modulated carriers are
orthogonally transmitted via Orthogonal Frequency Division Multiplexing (OFDM), has
increased transmission capacities to the domain of terabits-per-second.
Increased bandwidth demands, however, show no signs of slowing down with 22 % of
the global population owning a smartphone in 2013 [6]. With 5G communication expected to be
widely rolled out by 2020 in order to bolster applications such as Augmented Reality (AR)/
Virtual Reality (VR), the Internet of Things (IoT) as well as communication between
autonomous vehicles, a new set of tools with different building blocks has to be developed to
work with the existing optical infrastructure in order to meet the bandwidth demands. The core
tenets that have been formulated for 5G communication are enhanced Mobile Broadband
(eMBB) leading to transmission capacities of several gigabits-per-second for high speed
streaming and downloads, Ultra-Reliable Low Latency Communication (URLLC) for latencies
on the order of 1 millisecond and packet drop rates on the order of 10-5
to ensure "mission
critical" data transmission e.g. for remote surgery, autonomous vehicle communication etc, and
10
finally massive Machine-Type Communications (mMTC) for a huge number of IoT devices to
communicate effectively e.g. smart cities [7]. This is summarized in Figure 2.3. below [8].
Among the key technologies being considered for 5G communication are the millimeter
waves (mm-Waves) that enable the leveraging of an unused swath of spectrum in the ~30 GHz-
300 GHz range where the wavelengths range from 1 mm-10 mm (hence the name mm-Wave).
One possible drawback of millimeter waves is that these frequencies suffer high attenuation
when met with physical obstacles such as buildings, foliage as well as in weather conditions such
as rain and fog [9]. As such, they are good for line-of-sight, point-to-pint links. This can also be
advantageous in the sense that the same frequencies can be re-used after a short distance with
minimal interference. Thus, 5G will require a dense deployment of "small cells" that act like
relays, essentially receiving signals from other base stations and redirecting them to users at
another location [10]. These small cells, combined with massive MIMO (multiple
transmit/receive antennas per cell) and adaptive beamforming (phased-array-like operation) can
lead to a focused beam of data that is aimed at the user with minimal interference. Due to the
dense deployment as well as the small size of these cells, they will be potentially mounted atop
lamp posts, traffic lights, kiosks etc. and these location constraints also place restrictions on the
Figure 2.3: 5G Primary Objectives [8]
11
device energy usage, footprint and cost. In order to mitigate these concerns, Photonic Integrated
Circuit (PIC) technology is a frontrunner as a scalable, low cost, high bandwidth and energy
efficient solution to this dilemma.
The ultra-dense nature of these small cells, paired with the massive volume of data
generated, makes the transportation of said data from the small cells to the core network with
minimal latency a challenge. This has led to the development of an architecture known as C-
RAN (Centralized /Cloud Radio Access Network) [11] with an optical 'fronthaul' network. In C-
RAN, a centrally located Base Band Unit (BBU) (or a few of them pooled together) controls
radio signals from one (or a few) Remote Radio Heads (RRH) which are connected via the
fronthaul network [12]. The optical network is a prime candidate for the fronthaul case due to its
low latency, scalability and high capacity. One proposed candidate for the mobile fronthaul
physical link in 5G networks is Radio-over-Fiber (RoF) which allows for high throughput
wireless services together with the coexistence of multiple radio access technologies [13],[14].
RoF also supports high bandwidth and multiplexed signals without the need for multi-gigabit-
per-second digital transceivers as is the case for Common Public Radio Interface (CPRI) [15].
The fiber-optic infrastructure of RoF also provides a suitable space for photonic aided processing
[16].
Figure 2.4: Attenuation in the mm-Wave Regime [9]
12
This fronthaul infrastructure will require a series of new building blocks for 5G access
points, aggregation points etc. These building blocks comprise of mm-Wave repeaters, mm-
Wave/Fiber Add/Drop as well as switches and multiplexers. The focus of this report shall be on
the basic building block that will assume some (if not most) of these functions, the Photonic
Integrated Circuit Wireless Reconfigurable Optical Add-Drop Multiplexer (PIC WROADM).
.
Figure 2.5: Fronthaul Network Comprising PIC mm-Wave WROADMs
13
3. mm-Wave Generation Techniques
This section comprises a broad (yet not exhaustive) review of techniques that have been used to
generate millimeter-waves. Traditional methods for mm-Wave generation primarily consist of
direct intensity modulation, external modulation and optical heterodyning [17]. More recent
efforts have also used non-linear effects in waveguide based devices to achieve mm-Wave
generation.
Direct Intensity Modulation involves using an electrical signal of required frequency to
directly modulate the current of a laser source followed by a photodetector as in [18], or by
optically filtering (using a Mach-Zehnder Interferometer in this case) the generated directly
modulated spectrum of a laser source and mixing two modes separated by the desired frequency
at a photodetector as in [19]. That being said, due to the very limited modulation bandwidth of
commercially available lasers, it is inefficient to generate high mm-Wave frequencies (e.g. in the
60 GHz band) in this manner [20].
Optical heterodyning consists of two (or more) optical signals that are simultaneously
mixed at a photodetector to obtain the desired difference frequency. This technique, however, is
very sensitive to laser phase noise and thus requires a complex laser assembly such as an optical
injection phase-lock-loop demonstrated in [21]. This drives up the cost and complexity of the
system. A more recent endeavor in this regard [22] uses a frequency stabilized Dual Wavelength
Figure 3.1: mm-Wave Generation by Filtering Spectra of Directly Modulated DFB Laser [19]
14
Fiber Laser (DWFL) together with a Tunable Band-Pass Filter (TBPF) and a spliced section of
Photonic Crystal Fiber (PCF) acting as a Mach-Zehnder Interferometer and frequency selective
element in the laser feedback loop in order to pick off two wavelengths with the requisite mm-
Wave frequency spacing (65 GHz in this case) as well as a narrow linewidth.
External modulation is an efficient way of generating mm-Waves due to the wide
modulation bandwidth, high input saturation power and small power consumption of available
modulators [17]. Depending on the modulator bias point Single Side Band (SSB), Double Side
Band (DSB) and Optical Carrier Suppressed (OCS) modulation can be achieved [23]. DSB
modulation is achieved by biasing one arm of a traditional Lithium Niobate (LN) Mach-Zehnder
Modulator (MZM) at the
point to obtain Intensity Modulation (IM), with the driving signal at
the mm-Wave frequency. SSB modulation is also achieved by biasing at the
point except in
this case, both arms of the LN-MZM are driven with the mm-Wave frequency and with an
additional phase shift of 90° in the lower arm. OCS modulation is achieved by biasing the LN-
MZM at the null point (or equivalently, ) and using a drive voltage of 2 . Thus, if the driving
signal frequency is
, and the carrier frequency is , then the output will have a
Figure 3.2: mm-Wave Generation using DWFL and PCF [22]
15
suppressed carrier, as well as two strong signals placed at
. Therefore, the
generated signals will be separated in frequency by the desired mm-Wave frequency. Of the
aforementioned techniques (DSB, SSB and OCS), OCS is the most efficient and resilient since in
case of DSB all components have a high bandwidth of operation requirement and the two
generated sidebands travel at different velocities and experience frequency selective fading
whereas in case of SSB the receiver sensitivity is lower [23]. Thus mm-Wave generation by OCS
has the highest spectral efficiency, receiver sensitivity and smaller power penalty than DSB and
SSB [17]. Also, the need for high spectral purity lasers is diminished since the optical sidebands
are derived from the same source and therefore the phase noise on the sidebands is totally
correlated at the source [24].
Figure 3.3: DSB Modulation (Top); SSB Modulation (Middle); OCS Modulation (Bottom)
16
Another viable technique for mm-Wave generation is by using an Optical Frequency
Comb (OFC). An OFC has a spectrum that consists of a series of discrete frequency lines that are
equally spaced apart by a difference known as the Free Spectral Range (FSR). A common
method of generating an OFC is via a Mode Locked Laser (MLL). Mode-locking occurs when
the different longitudinal modes of a laser are made to have a fixed phase relationship, or a
constant phase difference (thus being locked in phase). This results in the modes periodically
interfering constructively in order to give a pulsed output, with the pulses separated in time by
the cavity round-trip-time,
. This, in turn, dictates that the frequency spectrum will also be
a train of impulses separated by the inverse of the period,
, which equates to the mode
spacing of the laser. Mode-locking can be achieved either actively or passively. Active mode-
locking is achieved via placing an electro-optic modulator within the cavity which is driven at a
frequency equivalent to the mode spacing of the laser. This results in the modulated side bands
coinciding with the longitudinal modes one FSR away from the central mode, and this operation
continues on the neighboring modes until all the modes (within the gain bandwidth) of the laser
are phase locked. Passive mode-locking is commonly achieved via a saturable absorber, which is
an intracavity element that has an optical intensity dependent transmission. It attenuates low
intensity light, whereas high intensity pulses due to periodic constructive interference obtains
preferential transmission. Over time, this leads to the laser operating in pulsed mode. The
advantage of an OFC is that since the comb frequencies have a fixed phase relationship, they
exhibit a high degree of mutual coherence and as a result, low phase noise. The output spectrum
of an OFC can then be filtered to give frequency lines separated by the requisite mm-Wave
frequency (or an arbitrary multiple thereof, depending on the setup). This is implemented in [17]
where the MLL output frequencies are modulated, and then passed on to a rapid reconfigurable
17
ring-resonator based filter in order to pick off frequency lines with a frequency spacing equal to
an arbitrary integer multiple of the base frequency.
However, a disadvantage of Mode-Locked Lasers is that they suffer from a large
linewidth, mode partition noise that can be imparted to the heterodyned signal as well as a fixed
FSR [25]. Another method that is therefore employed for OFC is gain switching of a laser. It is
advantageous in the sense that it can be used to produce optical pulses with a high repetition rate
and high peak power using laser diodes of any structure [26]. This technique involves
modulating the drive current of a laser diode just below threshold with a base frequency of which
the mm-Wave frequency is a multiple. At a certain level of current injection, the number of
carriers in the active region of the device exceeds the threshold and stimulated emission occurs,
which in turn depletes the carriers faster (to below threshold) than the current injection and
therefore termination of the output occurs. This results in a pulsed output dictated by the driving
source. An example of such a system that incorporates a gain switched DFB laser with a tunable
FSR is provided in [25].
Figure 3.4: Arbitrary mm-Wave Generation using MLL and Programmable Filter [17]
18
In all of the aforementioned techniques, the desired mm-Wave is generated by feeding
the optical signal to a photodetector where two coherent optical tones are heterodyned to
generate the mm-Wave equivalent to the frequency offset between them. Consider two optical
fields with amplitudes and , frequencies and , and instantaneous phases and ,
respectively:
(3.1)
(3.2)
The generated photocurrent at the photodetector (of limited bandwidth) is proportional to the
intensity of the summation of the two fields:
(3.3)
Therefore, provided there is low phase noise and the bandwidth of the photodetector is sufficient,
desired mm-Wave signals can be generated if the frequency offset between and is
carefully selected.
Figure 3.5: OFC Generation using Gain Switched DFB(Left) and Resulting OFC Spectrum (Right) [25]
19
4. PIC WROADM Core Technologies Overview
A Reconfigurable Optical Add-Drop Multiplexer (ROADM) is a device that is employed in a
Wavelength Division Multiplexing (WDM) optical network as a means of remotely adding or
dropping wavelength traffic, or allowing it through (also known as "cut-through") without
resorting to Optical-Electrical-Optical (O-E-O) conversion. This is achieved via the use of
Wavelength Selective Switches (WSS) in order to route wavelengths to different optical fibers
(or lightpaths). A WSS is an essential building block of a wavelength routing architecture and it
can consist of a , , or configuration. The architecture consists of a
common input port from which a wavelength can be assigned to any of the output ports,
whereas the configuration consists of an and configuration in a back-to-
back manner [27]. The technologies commonly used in WSSs are Liquid Crystal on Silicon
(LCoS) and Micro-Electro-Mechanical Systems (MEMS). In a configuration, the input
composite optical signal is passed on to an optical demultiplexer to separate the individual
wavelength channels, which are then spatially redirected to the designated output port via the LC
pixels or the micromirrors on the MEMS chip [27]. In addition, groups of micromirrors or LC
pixels can be used to dynamically adjust the spectral width of a channel [27]. A traditional two-
degree ROADM Node is shown below [28].
Figure 4.1: Block Diagram of Traditional ROADM Node [28]
20
The subject of this report is a Wireless Reconfigurable Optical Add-Drop Multiplexer
(WROADM) that will purportedly be a key building block in the fronthaul infrastructure for
future 5G networks, as displayed in Figure 2.5. In order to meet the device footprint restrictions
of being deployed at traffic lights, lamp posts etc. while simultaneously being energy efficient
and cost-effective, the WROADM being investigated is a Silicon Photonic Integrated Circuit
(SiPIC). Silicon photonics, which uses Silicon-on-Insulator (SOI) as an optical waveguiding
medium, is a strong contender for low-cost, high bandwidth photonic microchip devices by
leveraging the already well established foundry fabrication processes for commercial electronic
microchips.
Figure 4.2 shows a functional overview of the proposed PIC WROADM device. It must
have the capability of receiving and transmitting signals at the chosen wavelengths of operation
which are 14 GHz and 28 GHz (technically below 30 GHz but they offer attenuations of ≤ 0.1
dB/km). The device must also possess signal add/drop capability either to/from fiber as a Radio-
over-Fiber optical signal, or to/from the 14 GHz or 28 GHz carrier-centered electrical OFDM
signal. Moreover, the device must also be capable of optical signal processing e.g. filtering in the
Figure 4.2: Functional Overview of PIC WROADM
21
optical domain in order to switch the OFDM data from the 14 GHz carrier to the 28 GHz carrier
and vice versa.
4.1. Orthogonal Frequency Division Multiplexing (OFDM)
Orthogonal Frequency Division Multiplexing (OFDM) is a Multi-Carrier Modulation (MCM)
scheme that is widely used for mm-Wave applications today and is rapidly gaining interest for
optical transmission as well. In this scheme a single high data-rate stream of information is split
up into multiple lower data-rate streams, which are transmitted over orthogonal subcarriers [27].
As a consequence of this orthogonality, the OFDM subcarriers overlap in the frequency domain
(as opposed to having "guard bands" between them as in Frequency Modulation) and so can be
packed closer together in a given bandwidth, thus exhibiting high spectral efficiency since the
subcarriers are typically modulated via 2D signaling constellation schemes e.g. Quadrature
Amplitude Modulation (QAM). This orthogonality results from each subcarrier having an integer
number of cycles in a given symbol duration , and number of cycles between adjacent
subcarriers differs by an integer multiple (usually one). Therefore, if denotes the frequency of
the i-th subcarrier, then the following holds true for two adjacent subcarriers (k and l) [27]:
(4.1)
where n is an integer. The waveform for the k-th subcarrier is then:
where
(4.2)
When Equation 4.1 holds true, we see that the orthogonality between subcarriers, which is a
correlation function, is valid i.e. the correlation equates to zero:
(4.3)
22
OFDM is advantageous because it exhibits robustness to chromatic dispersion,
Polarization Mode Dispersion (PMD) and Polarization Dependent Loss (PDL), as well as
adaptivity to time-varying channel conditions, flexibility for software-defined optical transport,
straightforward channel estimation, and low complexity compared to conventional equalizer
schemes [27]. Also, since the individual subcarriers can be considered as orthogonal basis
functions, an arbitrary OFDM signal can be generated via a superposition of these orthonormal
basis functions, just like a discrete Fourier Transform. Since the subcarriers are orthogonal and
hence do not interfere, the inverse transform is implemented at the receiver side in order to
uniquely detect them. A conceptual diagram of OFDM generation and detection [29] is shown
below:
Even though neighboring subcarrier pulses overlap to some extent at an observed OFDM
subcarrier location (as can be seen in Figure 4.3), the neighboring subcarrier pulse shapes cross
zero where the peak of the detected individual subcarrier lies [27]. For an OFDM signal with N
Figure 4.3: Frequency Division Multiplexing (Left) and Orthogonal Frequency Division Multiplexing (Right)
Figure 4.4: Conceptual Diagram of OFDM Generation/Detection [29]
23
subcarriers, each spaced apart by a frequency (which is equivalent to the baud rate in
frequency), the bandwidth is equal to N .
4.2. Radio-over-Fiber (RoF)
Radio-over-Fiber (RoF) is a technique that may be used in conjunction with mm-Waves in order
to reduce spectral congestion and enhance the provision of broadband, interactive and
multimedia services in both mobile and fixed cellular networks [30]. RoF involves modulating
an optical carrier with a Radio Frequency (RF) subcarrier, followed by transmission over an
optical fiber. The advantages of this technique are leveraged on the properties of optical fibers
being high bandwidth, low loss, robust to Electro-Magnetic Interference (EMI) as well as being
cost-effective as opposed to all-electrical transmission.
RoF is essentially an example of an analog transmission system whereby the radio
waveform is transmitted directly on the radio carrier frequency from a central Control Station
(CS) to a Base Station (BS), while the data signal itself can be digital e.g. OFDM or QAM [30].
Figure 4.5 demonstrates a basic RoF link whereby an external modulator (driven by the
modulated RF signal) is used in tandem with a continuous wave (CW) laser to generate the
signal, followed by an optical fiber and a photodiode at the receiver to recover the data signal
[30]. This technique integrates well with the idea of optical fronthaul mentioned in Chapter 2,
with a Base Station only taking part in Electrical/Optical conversion while a central Control
Station (shared by several BSs) handles routing and resource management etc.
Figure 4.5: Basic RoF Link [30]
24
4.3. PIC WROADM Block Diagram
As mentioned earlier in this chapter and depicted in Figure 4.2, the proposed PIC WROADM
device must possess the capability of:
receiving and transmitting mm-Wave signals at 14 GHz and 28 GHz
add/drop to/from fiber
add/drop to/from mm-Wave
filtering and switching mm-Wave carriers
Figure 4.6 depicts a possible block diagram configuration for such a PIC WROADM device.
Optical signals are displayed in red, whereas electrical signals are displayed in blue. Incoming
RF OFDM signals are input to the drive arms of Mach-Zehnder Modulators (MZMs) to generate
RoF signals. These signals can either be dropped to optical fibers (dotted lines), passed straight
through to the photodetector to obtain the baseband OFDM signal (dashed lines), or passed
through a filtering stage whereby a new optical carrier is added such that it switches over the
OFDM data signals to the other mm-Wave carrier (solid lines). In addition, RoF signals can be
added from fiber and directed to the photodetector instead (dash-dotted lines).
Figure 4.6: PIC WROADM Block Diagram
25
The optical wavelengths of operation lie within the C-band and hence face an attenuation
of ~0.2 dB/km in Standard Single Mode Fiber (SSMF). They have been carefully chosen such
that the new optical carrier, , lies exactly midway between and . This means that the
OFDM sideband spaced at 14 GHz from 192.992 THz ( ) is now spaced at 28 GHz from
193.035 THz ( ), whereas the OFDM sideband spaced at 28 GHz from 193.077 THz ( ) is
now spaced at 14 GHz from 193.035 THz ( ). If this signal were filtered and routed to the
photodetector, the beating would result in the 14 GHz signal switched to 28 GHz carrier and vice
versa. This is shown in a frequency space diagram below, where the electrical OFDM signals are
represented in blue and the optical carriers are shown in red, with the new carrier ( ) shown
with a dotted outline.
Figure 4.7: RF Carrier Switching Frequency Space Diagram
Possible
Filter
Locations
26
5. Simulation Setup and Results
Simulations for this report were carried out in Lumerical software. This encompasses Lumerical
FDTD for solving 3D/2D Maxwell's equations for nanophotonic devices as well as materials,
Lumerical MODE for waveguide design and simulation, Lumerical DEVICE for performing
charge transport simulations and finally Lumerical INTERCONNECT for verifying system-level
operation of a photonic integrated circuit (PIC), using building blocks such as compact models
generated via simulation as well as a vast library of already tested models and elements,
including foundry-specific Process Design Kits (PDKs). Figure 5.1 [31] shows the Lumerical
products family and their inter-operability.
The following sub-sections proceed to further describe how FDTD and compact model
generation work in order to perform photonic simulations.
Figure 5.1: Lumerical Products Inter-operability [31]
27
5.1. Finite-Difference Time-Domain (FDTD)
The Finite-Difference Time-Domain (FDTD) method is a very useful technique for solving
Maxwell's Equations in 2D/3D for complex geometries and is equivalent to viewing 'a movie of
the fields in time and space' as the electric and magnetic fields evolve. This technique is
implemented by discretizing the Maxwell's equations in space and time via central difference
approximations, followed by the solution of the resulting finite-difference equations by marching
in time for the evolution of the procedure [32]. This 'marching in time' is done in a "leap-frog"
manner, which essentially means that initially all magnetic field components are solved for the
geometry for an instant in time, followed by all electric field components for the next instant in
time, and so on [32].
Writing down the Maxwell's equations for the electric and magnetic fields respectively:
(5.1)
(5.2)
where is the permittivity and is the permeability of the material. Rearranging these equations
gives:
(5.3)
(5.4)
Approximating the partial derivatives as a finite-difference:
(5.5)
(5.6)
28
Rearranging these equations provides us with a means to update the fields at an instance that
occurs slightly later in time, using prior knowledge of the fields (at a slightly earlier instance in
time):
(5.7)
(5.8)
Iteratively solving these equations for each point in space yields the electric and magnetic field
evolution over the target geometry. This is a relatively simplistic description of the steps required
to provide the reader with an overview, while in practice scaling of field amplitudes is also
necessary due to discrepancy between electric and magnetic field amplitudes, as well as
accounting for boundary conditions so that the fields' behavior at boundaries is correctly
accounted for. These steps are done by the software in Lumerical.
The advantage of FDTD lies in the fact that it is a time domain technique and that a
single simulation results in a solution that provides the response of the system to a broadband
range of frequencies, and a Fourier transform of the temporal waveform provides the spectral
decomposition of the solution [33]. A possible drawback of FDTD is that it is not suited to
resonant devices whereby the simulation will run for an indefinite amount of time.
5.2. S-Parameters and Compact Model Creation
S-parameters, or Scattering parameters, are a very useful tool that can be used to analyze linear
and time-invariant (LTI) systems and provide the response of such a system as a function of
frequency/wavelength. LTI systems are such that if a single frequency (sine wave) is input to the
system, the output is a sine wave of the same frequency but with a modified amplitude and a
shift in phase. S-parameters, as such, denote the complex amplitude (i.e. both amplitude and
phase) response of a system. This is depicted below in Figure 5.2 [34]:
29
S-parameters can be grouped together to form the S-matrix of a device, e.g. a two-port
device has an S-matrix with four elements that describe the reflection/transmission of the device.
For such a device, the elements are numbered such that the first number following the
denotes the output port and the second number denotes the input port e.g. refers to a measure
of the signal emerging from the output port when a stimulus is applied to the input port ( then
refers to the opposite case) [34]. If both numbers are the same ( or ), it denotes a reflection
measurement from said port.
From Figure 5.3 depicting a two-port device and it's S-matrix, the outputs and the
inputs form a system of linear equations:
(5.9)
or in matrix form:
(5.10)
As an example, the S-matrix for an ideal (no back-reflections at either input/output) waveguide
of length L is:
(5.11)
Figure 5.2: Response of a Linear Device [34]
Figure 5.3: S-Matrix of a Two-Port Device
30
where [ ] is the propagation loss, and is the propagation constant (
). The
reason that both and are identical is that an ideal waveguide is bidirectional.
In the Lumerical software, FDTD can be used to generate an S-matrix for a device by
taking the Fourier transform of a pulse that is launched into each port. For example, for a Y-
splitter, light will have to be individually launched into each of the three ports (which is a time
consuming simulation). Since physics based simulations such as FDTD are time consuming (as
they literally solve the fundamental differential equations underlying the device's function), it is
common to create Compact Models of the devices, the most typical method of which is the S-
matrix approach which can be thought of as a black box summarizing the complex transmission
and reflection coefficients of the device. Another method of creating compact models is via
fitting of empirical data of device responses (which is how foundry PDK element compact
models are generated).
Once the S-parameters of the necessary components are generated and hence the compact
models created, the individual compact models can be exported to Lumerical INTERCONNECT
and linked together in a chip layout fashion in order to simulate the overall circuit. These circuit
level simulations can be used to obtain data both in the time or frequency domain as well as
being converted to a photonic chip layout.
Figure 5.4: Three-Port Y Splitter
31
5.3. Waveguide Design
The first step towards realizing the proposed device was to model a waveguide in silicon. To this
extent, Lumerical MODE, which is an optical mode solver, was used to determine how an optical
mode would propagate through a silicon waveguide structure and obtain the effective index and
group index characteristics with respect to wavelength of operation. The waveguide structure
consists of a strip of silicon about 220nm high and 500nm wide, on top of an insulator (SOI)
approximately 1000-3000 nm high which is SiO2 (or silica), which in turn rests atop the substrate
(silicon handle wafer) that is approximately 500 µm high. The silicon strip (n ≈ 3.5) itself is clad
in SiO2 (n ≈ 1.5) that helps provide better coupling to an optical fiber, as well as provides a
reduced index contrast (as opposed to silicon and air) which helps reduce scattering and hence
propagation loss. Another advantage of the cladding layer is that it allows electrical contacts and
signals to be routed over the silicon layer without interacting with the optical mode.
The dimensions of 220 nm by 500 nm are chosen since they offer a good combination of
single-mode behavior (a height taller than ~250 nm and a width greater than ~500 nm is
sufficient to excite higher order modes), as well as high confinement of the optical mode [35]. A
smaller width and height of silicon decreases the effective index (which is essentially a spatial
average of the refractive indices of the waveguide and surrounding media) since the mode senses
less silicon compared to the surrounding oxide (or air). Figure 5.6 [35] shows plots of effective
Figure 5.5: Geometry of Strip Waveguide with Cladding
32
index versus silicon waveguide height and width, together with propagation of different modes
(at a wavelength of 1550 nm) in order to highlight single mode cutoffs.
In order to perform the simulation, the desired waveguide geometry is defined in
Lumerical MODE (which in this case is a silicon strip 220 nm by 500 nm and surrounded by
SiO2), and then the materials are configured. This configuration is accomplished in the 'Materials
Explorer' tab where a multi-coefficient model is used to fit plots of refractive index data versus
the wavelength range of the simulation for both silicon and SiO2. This is followed by defining a
2D mesh for the eigensolver, across the cross section of the waveguide and cladding while
making sure that the entire waveguide as well as cladding on all sides lies within the meshing
region. In order to maintain accuracy while decreasing simulation time, steps can be taken to
have a mesh that is finer in the vicinity of the waveguide and coarser farther away from it.
Finally, the simulation is run by defining the center wavelength and the number of trial modes to
calculate. The result is obtained as a list of modes with the effective index observed as well as
loss experienced in dB/cm and the TE polarization fraction of each mode (a value close to 100
means TE polarized). The software also provides spatial mode profiles for amplitude and phase
Figure 5.6: Effective Index and Multimode Behavior vs. Silicon Waveguide Thickness and Width [35]
33
of the electric and magnetic fields. This is followed by going to the 'Frequency Analysis' tab of
the eigenmode solver and running a frequency sweep which automatically provides a plot of the
effective index and group index versus wavelength for the calculated modes of the simulation
geometry. This frequency data can then be exported to a file, which can in turn be loaded into
INTERCONNECT as a compact model of our waveguide design.
(a) (b)
(c)
(d)
(e)
Figure 5.7: (a) Cross-Section of Simulation Region with Silicon Waveguide Surrounded by SiO2 Cladding (b) Mesh with Finer
Resolution Closer to Silicon Waveguide (c) Spatial Profile of TE Mode (d) Spatial Profile of TM Mode (e) Group Index vs.
Wavelength of Operation
34
5.4. Y-Branch (Splitter/Combiner) Design
A Y-Junction or Y-Branch is a device that splits input light into two outputs of equal power
(which is half of the input power). It can also be used to recombine two inputs into one output.
Since this device has three ports, it has an S-matrix with 9 terms, which need to be calculated in
order to generate a compact model. This is accomplished by using Lumerical FDTD to send a
short pulse individually into each port and observing the output at the remaining two ports using
a function called 'S-parameter' sweep. This automatically generates an S-matrix and hence a
compact model for the device/component/structure being simulated. The compact model is then
loaded into Lumerical INTERCONNECT in order to observe the reflection and transmission
characteristics of the Y-junction.
Initially, the polygon tool is used to define the segments that make up the Y-splitter,
followed by material configuration as in Section 5.3, as well as the definition of the FDTD
simulation region such that the entire Y-splitter is encompassed in all three dimensions (the
edges should extend slightly outside so that the simulation does not have artifacts at the
boundaries). Then, the three ports are defined under the FDTD menu, each one being
Figure 5.8: Y-Splitter Port Definition and Input Pulse Settings
35
bidirectional and situated at the extreme ends of the Y-splitter. Next, the source settings are
modified to incorporate a wavelength bandwidth (1.5-1.6 µm) which will correspond to a pulse
in the time domain that spans our frequencies of interest. Care must be taken to ensure that the
light launching direction is backward for the two output ports when evaluating the S-parameters
in order to get correct results. Finally, in the 'Optimizations and Sweeps' tab, an S-parameter
sweep is set up that detects the ports that have been defined (and numbers them in the order that
they were placed), and then automatically runs the FDTD simulation three times, once for each
port, and calculates the S-matrix parameters. This simulation takes approximately 2 hours to
complete.
The generated S-parameters are then imported into Lumerical INTERCONNECT into an
'Optical N-port S-Parameter' element, that automatically configures itself to display the requisite
number of ports. The positioning of the ports can also be edited. The transmission/reflection
characteristics of the device can be observed via using an Optical Network Analyzer which has
the option to edit the output power as well as frequency range (as broadband or narrowband as
desired) and can have an arbitrary number of inputs for multiple simultaneous measurements.
Figure 5.9: Optical Network Analyzer Setup and Frequency Response of Y-Splitter
36
As can be seen from Figure 5.9, both outputs are identical, and approximately -3 dB
down where exactly -3 dB refers to a power splitting ratio of 50:50. However, it is not exactly
50:50 since we see additional loss (-0.31 dB per port at best, and gets worse as wavelength
decreases). This loss is due to reflection from the junction region which might be modeled to be
too sharp a corner and hence it leads to light reflecting back to the input port.
5.5. Mach-Zehnder Modulator (MZM) Design
With passive components out of the way (filter design will be discussed in a later section), the
most important component to be designed is the Mach-Zehnder Modulator (MZM). The MZM is
crucial for generating the RoF signal by modulating the electrical OFDM signal on mm-Wave
carriers with an optical carrier frequency to facilitate routing across the chip and in fibers. The
MZM discussed in this section is a Silicon Electro-Optic modulator. It is an active device
because it essentially consists of a PIN or PN junction located at the waveguide region. In the
case where the diode is forward biased (PIN case), carriers are injected into the intrinsic region
when it is turned on, and extracted when it is turned off. This configuration provides a large
refractive index change and is relatively easy to fabricate, however it is limited by carrier
recombination time (~500 ps) both when they are injected as well as when they are extracted, so
it tends to be slow [36]. It also has a large absorption (and hence loss) when in the "on" state due
to the large number of injected carriers, as well as having high power consumption since the
diode conducts a current in the "on" state. The silicon modulator discussed here is operated in
depletion mode, which means that the diode is reverse biased (PN case). This configuration is
modulated by just varying the depletion region width, while keeping the diode reverse biased.
This is advantageous because the modulator operates very fast since it is akin to a capacitor and
hence RC constant limited. Also, since the modulator is always operated in reverse biased mode,
it has a low power consumption. The refractive index change obtained via this technique is
37
small, however, since it is limited by how much the depletion region width varies, and also since
the waveguide itself is doped, there is greater optical loss [36].
These modulators operate based on the principle of the Free Carrier Plasma Dispersion
Effect. This effect denotes a change in the refractive index and absorption of silicon with a
change in the concentration of free carriers. The change in index subsequently corresponds to a
change in the phase of the light that passes through the waveguide, and hence, if used in a Mach-
Zehnder Interferometer configuration where the input light is split into two arms and
recombined, and the phase accumulated between the two arms varies based on this effect, the
output light can be switched "on" (constructive interference/2π phase difference) or "off"
(destructive interference/π phase difference). The Free Carrier Plasma Dispersion Effect can be
modeled by the empirical relation from Nedeljkovic, Soref and Mashanovich [37]:
(5.12)
where is the change in refractive index, is change in electron concentration [cm-3
] and
is change in hole concentration [cm-3
]. Meanwhile the depletion region width varies with the
concentration of electrons and holes as:
(5.13)
where is the permittivity of the medium, and
. Since NP is much larger than
(N+P), the depletion region width decreases with increasing electron and hole concentration.
Figure 5.10: Injection Based Modulator (Left); Depletion Based Modulator (Right)
38
Consequently, closer to forward bias, the depletion region width is very small and the optical
mode experiences a smaller index relative to undoped silicon due to the high carrier
concentrations (smaller due to the negative coefficients in Equation 5.12). As the diode is reverse
biased more, the depletion region becomes wider and the optical mode sees more undoped
silicon and as a result the refractive index increases. Sweeping between the two cases lends itself
to an effective index modulation, which in turn causes phase modulation. In an MZI
configuration, if this phase modulation is sufficient to warrant a π phase difference between the
two arms, amplitude modulation results.
The MZM simulation involves creating silicon based depletion PN phase shifters and
incorporating them into the arms of an MZI. The first step is to simulate the effective index of
refraction for an undoped waveguide using Lumerical MODE. The waveguide used here is a rib
waveguide that is 220 nm high and is etched down to 110 nm on the sides, and then goes back to
the full height silicon on the sides. The method is identical to that used in Section 5.3 and we
obtain a compact model consisting of the effective index of the rib waveguide as a function of
wavelength.
The next step is to use Lumerical DEVICE in order to solve for the charge distributions
in a doped version of the waveguide, as the bias voltage of the PN junction is varied from a small
amount of forward bias (+0.5 V) to fully reverse biased (-4 V). The corresponding effect on the
Figure 5.11: Undoped Rib Waveguide Geometry (Left); Effective Index vs. Wavelength for Undoped Rib Waveguide (Right)
39
depletion region width is observed. The charge simulations are performed for two different
doping levels i.e. 1E17 cm-3
and 1E18 cm-3
to observe the impact of carrier concentration in the
vicinity of the PN junction on the depletion region width. Figure 5.12 shows the PN phase shifter
geometry as created in Lumerical DEVICE, as well as a labeled cross section. The regions
labeled N++ and P++ have a doping off 1E20 cm-3
each for offering lowest resistance to
electrical contacts while N+ and P+ have a doping of 1E19 cm-3
each. N and P dopings are both
toggled between 1E17 cm-3
and 1E18 cm-3
.
A charge monitor is deployed in the simulation region, which will automatically display
charge distributions once the simulation is complete. The anode voltage is cycled from +0.5 V to
-4 V in decrements of -0.5 V, while the cathode is kept at ground. A steady-state simulation is
then run which self-consistently solves Poisson's equation as well as free carrier density
(drift/diffusion) equations, and the charge distributions visualized. Observing the charge
distributions vary with increased reverse bias also provides information about the depletion
region width. Figure 5.13 shows a comparison of the depletion region widths for slightly forward
biased versus fully reverse biased PN junction waveguides, for both N = P = 1E17 cm-3
and N =
P = 1E18 cm-3
:
(a)
Figure 5.12: Silicon PN Phase Shifter in Lumerical DEVICE (Left); PN Phase Shifter Cross-Section(Right)
40
We see that the varying voltage causes a varying depletion region width, which in turn
causes varying carrier concentrations in the PN junction's vicinity. This in turn causes a spatially
varying refractive index profile in that vicinity due to the Free Carrier Plasma Dispersion Effect.
However, what we are interested in is the change in effective index of the waveguide (relative to
undoped silicon), since that is the index that the optical mode sees. This change can be calculated
as a mode overlap integral (calculated in Lumerical MODE by incorporating the index
perturbations resulting from the charge simulations and subsequently subtracting the results from
the undoped waveguide simulation) between the change in spatial distribution of the refractive
index with voltage and the electric field intensity in the waveguide region (normalized by the
intensity):
(5.14)
(b)
(c)
(d) Figure 5.13: (a)-(b) N=P=1E17 cm
-3Depletion Region Widths at Forward and Reverse Bias; (c)-(d) N=P=1E18
cm-3
Depletion Region Widths at Forward and Reverse Bias;
41
Consequently, even though having a higher doping in the PN region makes it more difficult to
fully deplete the diode, the change in effective index, or effective index modulation
( ) for the case of N = P = 1E18 cm-3
is much larger (~3 times
larger) than the N = P = 1E17 cm-3
case. This makes it easier to induce a phase change of π
radians when the PN phase shifter is incorporated into an MZM configuration. The only
drawback is that the higher carrier concentrations lead to a much higher propagation loss for the
N = P = 1E18 cm-3
case (~22 dB/cm versus 2 dB/cm), however the loss tends to decrease as the
junction is depleted. The change in effective index versus voltage for the two different dopings is
shown in Figure 5.14:
Finally, the results of these simulations are loaded into Lumerical INTERCONNECT to
form a Mach-Zehnder Modulator. The changes in effective index versus voltage for the N = P =
1E18 cm-3
case are loaded into the 'Modulator (measured)' element, and these phase shifters are
incorporated into the arms of an MZM together with undoped lengths of silicon waveguides to
get an unbalanced ( ) dual-drive configuration. An Optical Network Analyzer
(ONA) is used to observe the frequency response of the system in the range of 1500-1600 nm. A
voltage sweep is created so that the phase shifter voltage is varied from +0.5 V to -4 V (reverse
Figure 5.14: Change in Effective Index vs. Voltage for 1E17 cm-3
and 1E18 cm-3
Doping Concentrations
42
bias), and the transmission modulation monitored. Slices of the transmission spectrum at the
wavelengths of operation (1552.71 nm and 1553.39 nm) for the PIC WROADM are shown
below in Figure 5.16. Maximum transmission is achieved at a forward bias of +0.5 V
(constructive interference), and minimum transmission is achieved at a reverse bias of -4 V
(destructive interference). The MZM exhibits an insertion loss of -3.35 dB at 1552.71 nm and -
3.27 dB at 1553.39 dB. The Extinction Ratio (ER) at these wavelengths is -17 dB and -23 dB
respectively. The length of the phase shifter required to induce a phase shift of π radians between
the light in the two arms is known as , where
. This value comes out to be
approximately 2.877 mm for both wavelengths of operation. Operating the MZM in push-pull
mode i.e. when the voltage applied to the top arm is +0.5 V then the voltage applied to the
bottom arm will be -4 V and vice versa, allows us to get a phase shift of 2π for the same applied
voltage range, effectively halving , or consequently allowing us to use half the length of phase
shifters for a π phase change. This gives us an of 1.438mm and together with a of 4.5 V
(0.5 V - (-4 V)), we have a product of 0.647 V•cm.
Figure 5.15: Lumerical INTERCONNECT Setup for MZM Incorporating N = P =1E18 cm-3
Silicon Phase Shifters
43
Now, our modeled MZM can also be viewed as a voltage dependent RC circuit, with the
PN junction/depletion region boundaries forming a capacitor, and the doped regions connected to
electrical contacts acting as resistors. Such devices have a frequency bandwidth of operation
known as the 3-dB bandwidth and as such behave like a low-pass filter with a 3-dB roll-off
frequency (or ). This frequency can be calculated as:
(5.15)
where is the resistance of the N-doped region, is the resistance of the P-doped region and
is the voltage dependent capacitance of the junction. The resistances are obtained via placing a
virtual ground at the edges of the waveguide at the N region and the P region respectively and
performing steady state DC analysis (normalized to ) for each case. Thus, the resistances are
where V is a small applied voltage. The capacitance is calculated by changing the
voltage by a few millivolts, at each bias voltage, and integrating the charge in a 2D cross-
sectional area to figure out how much the charge changes with the change in voltage. The
capacitance is then calculated via
. These values can be calculated using
Lumerical DEVICE's charge simulations whereby the charge monitor can automatically sum up
total charge in the simulation cross section, as well as place virtual grounds at the co-ordinate of
(a) (b)
Figure 5.16: (a) Slice of MZM Transmission Spectrum at 1552.71 nm; (b) Slice of MZM Transmission Spectrum at 1553.39 nm
44
choosing. The bandwidth of the device comes out to be approximately in the 30-40 GHz range,
however, these simulations were performed via a simplistic DC analysis by lumping the
elements. In reality the length of the device needs to be taken into account since at high
frequencies of operation, the RF wave is comparable in length to that of the modulator and hence
the voltage changes across the device. External impedance matching also becomes important at
high frequencies and as a result the actual bandwidth of the modeled MZM is lower.
5.6. NRZ-OOK Transceiver Simulation
Next up, the modeled MZM is incorporated into a transceiver simulation in Lumerical
INTERCONNECT and operated with Non-Return-to-Zero On-Off-Keying (NRZ-OOK)
modulation at data rates of 10 Gbps and 25 Gbps. This is achieved by removing the ONA and
adding a CW laser source at the wavelength of operation (1552.71 nm or 1553.39 nm), as well as
removing the DC Source as voltage input to the phase shifter. It is replaced by a Pseudo-
Random-Bit-Sequence (PRBS) generator operating at either 25 Gbps or 10 Gbps, which
generates a random bit sequence that is then fed to an NRZ-OOK pulse shaper which converts
the logical data to electrical pulses that vary from -4 V to +0.5 V in amplitude. An inverted copy
of the PRBS output is also fed to an NRZ-OOK pulse shaper, which is applied to the phase
shifter in the bottom arm of the MZM in order to ensure push-pull operation. Low pass RC filters
(25 GHz) are added before the phase shifter voltage inputs in order to model the 3 dB roll-off
frequency. Thermal tuners are also added to each arm in order to obtain maximum
transmission/extinction at the wavelength of operation. Finally electrical and optical
Figure 5.17: Silicon Based PN Phase Shifter as a Voltage Dependent RC Circuit
45
oscilloscopes are employed to look at the transmitted bit sequence, as well as a photodiode
placed at the output of the MZM to obtain an electrical signal that is fed into an eye diagram
analyzer to have a qualitative assessment of transmission quality.
Initially the simulation is run at 25 Gbps and the resulting eye diagram is shown below:
As can be seen in Figure 5.19, the eye is open and hence the modulator works, however it is
slightly skewed since there is some low pass filtering taking place. This means that the
Figure 5.18: MZM Based NRZ-OOK Transceiver Simulation Setup
Figure 5.19: Eye Diagram at 25 Gbps
46
modulation speed should be reduced to improve performance. The eye diagram for 10 Gbps
modulation speed is shown in Figure 5.20. As can be seen, the eye is more open and appears to
be more symmetric, hence 10 Gbps operation is much better for the 25 GHz modulation
bandwidth.
5.7. Filtering Stage Design
As shown in Figure 4.7, a filter stage needs to be employed in order to switch the mm-Wave
carrier frequencies for the electrical OFDM data. The conceptual frequency space diagram shows
a band-pass filter that allows the central sidebands from each of the 14 GHz and 28 GHz
modulated RoF signals to pass through, together with a new optical carrier that is spaced exactly
28 GHz from the 14 GHz modulated signal and 14 GHz from the 28 GHz modulated signal in
order to switch the carriers. It would be useful if instead of just suppressing/attenuating the
remaining optical carriers and sidebands, they could be rerouted and then transmitted over fiber,
in a Single Side Band (SSB) RoF fashion, if desired. Consequently a micro-ring resonator
(MRR) add-drop filter type approach is proposed. MRRs are also advantageous since they have a
very small component footprint and thus occupy less space on a chip, thus enabling the overall
device to be more compact.
A ring resonator consists of an optical waveguide looped back on itself, and it is akin to a
cavity whereby a resonance occurs whenever the optical path length of the resonator is equal to
Figure 5.20: Eye Diagram at 10 Gbps
47
an integer number of wavelengths (much like a folded Fabry-Pérot cavity) [39]. The transmission
spectrum of an MRR thus consists of multiple resonances separated by the Free Spectral Range
(FSR) of the device, and it therefore acts as a spectral filter with dips around the resonances. Due
to the high index contrast of silicon, very tight confinement of the optical mode can be
maintained, and consequently, very small bending radii can be achieved (as small as 5 µm),
resulting in a very small footprint [39]. MRRs also have one (or more) waveguide(s) closely
spaced next to the ring to act as a coupling mechanism such that when the round trip phase shift
through the ring is an integer multiple of 2π, the cavity is in resonance.
Figure 5.21 shows both an all-pass ring resonator and an add-drop ring resonator. The all-
pass configuration consists of a single bus waveguide in proximity to the ring resonator, and it
acts as a notch filter, whereby when an integer number of wavelengths fit into the optical path
length of the ring (i.e.
), the incoming signal is delayed via the temporary storage of
optical energy within the resonator [39]. The terms r and k are the self-coupling and cross-
coupling coefficients respectively, where r2 and k
2 represent the power splitting ratios (
indicates that there is no coupling loss), and a is the round-trip amplitude transmission
( ) [39]. When the coupled power is equal to the power loss in the ring (
), critical coupling takes place and the transmission at resonance goes to zero. For the
purpose of our device, the add-drop ring resonator is a prime candidate since it consists of
another bus waveguide above the resonator, and upon resonance the filtered signal gets coupled
Figure 5.21: (Left) All-Pass Ring Resonator; (Right) Add-Drop Ring Resonator [39]
48
to the other waveguide and can be rerouted. Critical coupling in this case occurs when the losses
match the coupling ( ) [39]. Figure 5.22 shows the transmission spectra for an all-pass
resonator (red) and an add-drop resonator (blue), where the subscripts 'n', 'p' and 'd' denote notch,
pass and drop respectively.
Melloni [40] has proposed a synthesis technique for maximally-flat (Butterworth) stop
band characteristic filters via using parallel coupled ring resonators that provide high selectivity,
high rejection out of band and flat in-band response. This technique involves cascading add-drop
ring resonators with identical lengths ( ) of waveguide that are equivalent to an odd multiple of
the quarter wavelength. More specifically, if is equivalent to an odd multiple of a half
wavelength, then the contributions from all the rings add in phase (constructively) upon
resonance, and the signal is dropped at the output port, whereas in the out of band regions the
contributions from all the rings do not add in phase and sometimes cancel out completely to
create nulls (destructive interference) [40]. Figure 5.23 depicts parallel coupled ring resonators
where is input port, is the drop port, is the through port and is the add port. N is the
filter order and the length of the rings is (where
) [40].
Figure 5.22: Transmission Spectra of All-Pass (Red) and Add-Drop Ring(Blue) Resonators [39]
49
Owing to the fact that FDTD simulation is non-ideal and indefinitely slow for resonant
devices, the simulation of the cascaded MRR filter is done via using compact models already
generated by the American Institute for Manufacturing Integrated Photonics (AIM Photonics)
and contained in a Process Design Kit (PDK) that can be loaded into Lumerical
INTERCONNECT's element library. This PDK contains a plethora of component compact
models ranging from passive components like waveguides, splitters and directional couplers to
active components such as photodetectors and tunable filters, all of which have been generated
using empirical data that has been extensively tested by AIM Photonics' state-of-the-art 300mm
wafer foundry. This ensures that the end result can be readily reproduced to a high degree of
accuracy and reliability when submitted to an AIM Photonics Multi-Project Wafer (MPW) run.
That being said, the AIM PDK components are essentially black boxes since they are the
intellectual property of the AIM foundation and as such this reliability and accuracy comes at the
cost of reduced flexibility in design since all parameters of a component cannot be modified. The
MRR chosen for simulation is the AIM Photonics C+L band tunable silicon microring resonator
filter, which has an FSR of ~26nm, 1 nm/mW thermo-optic tuning efficiency and a 3-dB
bandwidth (or FWHM) of ~80 GHz [41].
Figure 5.23: (Left) A Parallel Coupled Ring Resonator Filter; (Right) Transmission (dB) versus Frequency for
Parallel Coupled Ring Resonator Filter of Order 2 to 5 and FSR/Bandwidth = 10 [40]
50
The simulation is set up in Lumerical INTERCONNECT where the AIM PDK MRR
filters are cascaded together in a similar fashion as Figure 5.23, along with thermal tuning heater
voltage supplies. An Optical Network Analyzer (ONA) is used to provide a broadband input
from 1500 nm to 1600 nm to the input port, and the output at the through and drop ports is
monitored. In order to optimize the filter shape/passband characteristics, an optimization sweep
is set up whereby the lengths of each of the waveguide segments connecting the MRRs are
simultaneously varied from a few microns to thousands of microns and the transmitted spectrum
observed for each case. Figure 5.25 displays the simulation setup for a filter of order N = 3:
Figure 5.24: AIM PDK C+L Band Tunable Silicon Microring Resonator Filter [41]
Figure 5.25: Cascaded AIM PDK MRR Filter Transmission Measurement Setup
51
The transmission spectrum for the three cascaded MRRs is shown below in Figure 5.26.
The optimum (maximally flat/box-like) filter shape for this configuration occurs at waveguide
lengths of approximately 16 µm, however, it is observed that the Extinction Ratio (ER) at the
neighboring optical carriers (192.992 THz and 193.077 THz) is reduced due to the side lobes
from -25-30 dB up to about -7-10 dB which is not good since a sizeable portion of the
neighboring carriers' power will leak into the filtered spectrum with the new optical carrier
(193.035 THz), and will hence create extra mixing terms at the photodetector. The 3-dB
bandwidth of the filter's central lobe is ~76 GHz.
In order to obtain a smaller 3-dB bandwidth, as well as much greater out of band
rejection ratios, an additional cascade of 3 identical MRRs is created in INTERCONNECT, with
the filtered signal from the drop port of the primary cascade being fed into the input port of the
secondary cascade. Next, another optimization sweep is created which simultaneously varies the
lengths of the waveguides connecting the rings in the secondary cascade, as well as the heater
tuning voltages. This allows an identical copy of the first transmission spectrum (with varying
Figure 5.26: Drop Port Spectrum for 3 Ring Cascaded MRRs
52
filter bandwidth and FSR due to the varying connecting waveguide lengths) to be swept across
the initial plot and carve out the spectrum in order to obtain a greater out of band rejection
characteristic as well as narrower 3-dB bandwidth. This simulation setup is shown below in
Figure 5.27. The new filter 3-dB bandwidth is 65 GHz, and the out of band rejection goes from -
20 dB all the way down to -50 dB further away from the central lobe. The FSR of the filter is
~3.04 THz, and the insertion loss is ≤ -0.7 dB in the maximally flat part of the filter response.
The transmission spectrum for the drop (shown in blue) port and the through (shown in green)
port are displayed in Figure 5.28.
Figure 5.27: INTERCONNECT Simulation Setup for Primary and Secondary 3 Ring Cascaded MRR Filter Transmission Spectrum
Measurement
53
As can be seen in Figure 5.28 above, the notch-like filtering characteristic of the through-
port has a suppression of greater than -100 dB. Consequently, this filter can act as a tunable
selection mechanism for sending either mm-Wave signal and optical carrier as is to the
photodetector, or for choosing the central new carrier with neighboring sidebands and route them
to the photodetector for RF carrier switching.
5.8. System Eye Diagrams
Finally, the system level implementation of the chip is simulated in Lumerical
INTERCONNECT in order to observe the eye diagram for operation with NRZ-OOK
modulation. NRZ-OOK is used because Lumerical INTERCONNECT does not currently have
the option of creating an electrical OFDM signal in order to feed it to the MZM, or an Arbitrary
Waveform Generator (AWG) feature at that. Moreover, the chip is also implemented using AIM
PDK components since they are well tested and compact models are based on empirical data that
Figure 5.28: Transmission (dB) versus Frequency Plot for Two Stage Cascaded MRR Filter
54
guarantees accurate predictions as well as minimal process variation when submitted to an AIM
foundry MPW run. As such, the Mach-Zehnder Modulators used are AIM PDK Digital C+L
band traveling wave MZMs that boast operation at up to 50 Gbps, and a product of 1.2
V•cm for single arm operation, and 0.6 V•cm for push-pull operation [41]. However, in order to
utilize the full bandwidth and traveling wave operation of the device, on-chip or external
termination is required. In the version of the PDK used, the termination elements are not
included, and the termination ports on the MZM are disabled and cannot be connected to a
circuit element.
Initially, the MZM operation at the wavelengths of 1552.71 nm and 1553.39 nm is
characterized via using an ONA to provide a broadband spectrum (1500-1600 nm) to the input
port, while an optimization sweep varies the cathode voltage (the common anode is held at
ground) from -0.5 V (forward bias) to +7 V (reverse bias) in single arm operation mode. The
transmission spectra at each wavelength slice are then observed, and the spectrum at the
Figure 5.29: AIM PDK Digital C+L Band MZM (50G) [41]
55
wavelengths of operation are provided in Figure 5.30. It can be seen that the insertion loss for
both cases is ~ -4 dB and the ER is ~ -20 dB.
Next, the MZM performance with 25 Gbps NRZ-OOK is observed. A PRBS generator is
used to generate a sequence of order 7 (27- 1 bits), which are then fed into an NRZ pulse shaper
that is biased at -0.5 V and has an amplitude of +7.5 V. The output of the pulse shaper drives the
cathode of the top arm of the AIM PDK MZM such that a '0' bit corresponds to a slight forward
bias of +0.5 V resulting in constructive interference at the output (since the bottom arm is also at
+0.5 V), whereas a '1' corresponds to a reverse bias of -7 V which causes the optical signal in the
top arm to accumulate a phase difference of π radians with respect to the signal in the bottom
arm, resulting in destructive interference. The output signal is taken from the data-bar or
complementary output so that it matches the input bit stream, and is not inverted. The output
signal is then sent to a photodiode which converts the optical signal back to an electrical signal
which is then fed into an eye diagram analyzer that takes the output of the NRZ pulse shaper as a
reference signal. The simulation is then repeated at 10 Gbps NRZ-OOK modulation. The
simulation setup and results are shown below in Figure 5.31 (the wavelength used is 1553.39 nm
but the results are quite similar for 1552.71 nm as well).
Figure 5.30: AIM PDK MZM Transmission (dB) Spectrum versus Phase Shifter Voltage for 1552.71 nm and 1553.39 nm
56
(a)
(b) (c)
(d) (e)
57
The eye diagram for the 25 Gbps case is slightly skewed which might be due to the lack
of MZM output electrical termination. However, in both cases the eye is quite open therefore the
MZM performs well at both 25 Gbps and 10 Gbps (slightly better performance at 10 Gbps). The
eye diagram analysis tool also provides BER and Q factor information. The BER and Q factor
for the 25 Gbps case are 1e-144 (error free) and 44.6 respectively, whereas the BER and Q factor
for the 10 Gbps case are 1e-219 (error free) and 53.8 respectively (ideal transmission with no
loss other than MZM insertion loss and waveguide propagation loss).
Finally, the building blocks that have been individually tested are connected together in a
system layout shown below in Figure 5.32 (zoom in for optimal viewing) in order to observe the
RF carrier switching characteristic of the chip. The setup consists of two AIM PDK MZMs,
operating at 1553.39 nm and 1552.71 nm wavelengths which are represented by two CW lasers
with powers of 0 dBm and linewidths of 100 kHz. The CW lasers each pass through an AIM
PDK silicon edge coupler before entering the MZMs. The MZMs are driven by NRZ-OOK
modulation at 1 Gbps (which is more in line for an OFDM signal raw data rate) in the manner
discussed above. The output of the NRZ-OOK pulse generators is subsequently mixed with a 14
GHz and 28 GHz (for the 1553.39 nm and 1552.71 nm cases respectively) sine wave which
signifies the RF carriers, and the mixed signal is then sent through a high pass RC filter before
(f) Figure 5.31: (a) Simulation Setup for AIM PDK MZM operated at 10/25 Gbps NRZ-OOK; (b) PRBS Output after NRZ Pulse
Shaper; (c) Modulated Output after MZM; (d) OSA Spectrum for 25 Gbps NRZ-OOK Modulation; (e) Eye Diagram for 25 Gbps
NRZ-OOK; (f) Eye Diagram for 10 Gbps NRZ-OOK
58
being sent to the MZM driving cathode. This results in spectra with pronounced sidebands at 14
GHz from the 1553.39 nm optical carrier and 28 GHz from the 1552.71 nm optical carrier. The
output of the modulators are then combined via a PDK Y-junction, and another Y-junction is
used to add the new carrier at 1553.05 nm (193.035 THz) which is spaced equidistant in
frequency from the other two carriers. The new carrier is an identical CW laser with 0 dBm
launch power and 100 kHz linewidth. The final combined signal from the second Y-junction is
then fed to the six microring resonator twin-stage tunable filter. The filtered signal is routed to an
AIM C-band digital photodetector (1 A/W responsivity and > 45 GHz bandwidth) [41] that is
operating at a -1 V reverse bias. The electric signals from the photodetector are centered at 14
GHz and 28 GHz and so are then mixed with another sine wave of the same frequency acting as
a local oscillator to bring the signals back to baseband. The baseband signal is then passed on to
an eye diagram analysis element in order to calculate BER if required. The filter can be tuned to
coincide with the signal from each optical carrier, or moved to the center to pick off the new
carrier plus neighboring sidebands (as will be discussed shortly) and the signal at the
photodetector observed. New signals can be added via the add port of the primary filter stage
(with the passband tuned to minimize loss), and signals outside the filter passband are routed via
the through port of the primary filter stage over to fiber. In addition to this, a variable optical
attenuator can be added at the output of each MZM in order to observe the degradation of the
BER at the output of the photodetector as the passband signal is attenuated. The modulated
signal (which is incrementally attenuated) can also be sent directly to a photodetector (thus
bypassing the chip) followed by a local oscillator for downconversion and eye diagram analysis
tool which would calculate the BER for a 'back-to-back' case. This would serve as a stepping
stone for future works to characterize the optical/RF transfer function of the chip.
59
Figure 5.32: PIC WROADM System Layout in Lumerical INTERCONNECT
60
The modulated signal spectra, log(BER) vs. power as well as eye diagrams for the case
where the modulated outputs are directly sent to the photodetector, local oscillator and then eye
diagram analysis tool are shown below in Figure 5.33.
(c) (d)
(a) (b)
(e) (f)
Figure 5.33: (a)-(b) Modulated Output Signals; (c)-(d) Log(BER) vs. Power (dBm) Plots Using Signals Directly from Modulator Outputs;
(e)-(f) Eye Diagrams at 0 dB Attenuation
61
Lastly, the tunable filter is moved so that the passband encompasses the central carrier
plus the neighboring sidebands. The modulated signals are then combined with the new carrier at
1553.05 nm and routed to the filtering stage, the output of which is sent to the AIM
photodetector (connected to an RF spectrum analyzer). The results are summarized in Figure
5.34 below. It can be seen that at the output of the filter stage, the neighboring carriers are
attenuated by more than -30 dB, whereas the sidebands within the passband are only attenuated
by -0.6 to -1.5 dB. There can also be seen a harmonic term but it is > -10 dB below the
sidebands. This leaves the central carrier to mix with the neighboring sidebands at the
photodetector, resulting in the RF carriers switching as seen on the RFSA output. Qualitatively,
this depicts that the switching works, however, quantitatively RF amplification will be required
at the output of the photodetector to achieve appreciable power.
(a)
(c)
(a) (b)
(d)
Filter Region
Extra Harmonic
62
The signal at 14 GHz appears to have more power than the one at 28 GHz. This is
because the harmonic term that is a by-product of modulation is also spaced 14 GHz from the
central carrier and hence at the output of the photodetector it adds as noise power to the signal at
14 GHz.
(e) Figure 5.34: (a) 14 GHz Signal with 1553.39 nm Optical Carrier; (b) 28 GHz Signal with 1552.71 nm Optical Carrier; (c) Combined
Signal Plus New Carrier; (d) Filtered Signal with Additional Harmonic Term; (e) RF Spectrum Analyzer Output Showing Switched
Carriers
63
6. DRC Clean Chip Layout
The culmination of this report is with a Design Rule Checking (DRC) clean layout of an earlier
version of the proposed photonic chip. The reason for an earlier version is that access to the
layout software was available for a limited duration, hence this section corresponds more to a
learning opportunity as well as a stepping stone for future iterations of this project. The layout
has been implemented in Lumerical's KLayout software, which is used to create a GDSII (read
GDS-two) or Graphic Database System file that is the industry standard for integrated circuit
layouts in a hierarchical format. This file can be directly translated by a foundry in order to
fabricate a chip design.
The layout begins with a 'floorplan' which, as the name depicts, is the physical extent or
bounding box for the photonic integrated circuit. The dimensions of the floorplan used are 1866
µm by 934 µm (as part of a larger chip for a MPW run). The left and right edges of the chip are
bounded by dicing channels which will be cut by the foundry using wafer saws and these
channels define the chip edge for optical fiber coupling. For the purpose of this chip layout, edge
coupling is used since it is fairly broadband and has a lower coupling loss than grating couplers.
The edge coupler used is the silicon edge coupler in the AIM PDK, and it is essential to align the
edge coupler perfectly with the dicing channel so that no design rules are violated. For multiple
fiber inputs, a fiber array is used which consists of multiple fibers arranged in place via V-
grooves. To be able to use a fiber array, edge couplers must be spaced by exactly 127 µm. It is
also advised to place all fiber inputs on one side of the chip since a test station might not be able
to simultaneously test all fiber inputs/outputs otherwise. Figure 6.1 depicts the chip floorplan as
well as edge couplers spaced apart by 127 µm.
64
Next, the floorplan is defined as the 'top cell' which is where all components will be
added. Components are then added via placing instances onto the floorplan. These instances
consist of the AIM PDK components e.g. the MZMs, tunable ring resonators and photodetector.
Apart from these components, waveguide tapers may be necessary in switching from one
waveguide width to another if required. Signals are routed between components using
waveguides which exist on SEAM layer or GDS layer 709 in the AIM PDK. Other important
rules that must be followed are that waveguides cannot be misaligned to components and hence
they have to align perfectly as well as be of the same widths. AIM PDK components cannot have
any component placed in very close proximity to them since they are black boxes and their
performance could be inadvertently affected. Lastly, waveguides need to be a certain distance
apart which is indicated by the software since otherwise light might couple from one waveguide
to the other. These regions are indicated as waveguide keep-out (WGKOAM) regions on layer
GDS 802 and metal keep out (METKOAM) regions on layer GDS 803. For the Silicon phase
shifter based MZMs, ridge etches are also incorporated on surrounding the waveguide on the
REAM layer (GDS layer 702). The REAM layer must extend at least 250 nm on either side of
the SEAM layer to avoid misalignment issues, unless silicon ridges are required at the
extremities, in which case the SEAM layer should extend beyond the REAM layer.
Figure 6.1: Chip Floorplan Showing Edge Couplers and Dicing Channels
65
Figure 6.2 shows the SEAM and REAM layers above for a top view of the PN phase
shifter waveguide. The SEAM layer extends beyond the REAM layer in order to form ridges on
the sides for metal contacts. Next, rectangles are made onto these layers which will dictate where
the doping goes. The doping layers are NDAM (GDS 791) and PDAM (GDS 794), NNAM
(GDS 792) and PPAM (GDS 795), NNNAM (GDS 793) and PPPAM (GDS 796) which
correspond to N, P, N+, P+, N++ and P++ respectively. The doping layers should also be
extended slightly beyond the SEAM layer to circumvent misalignment tolerances. Finally, the
metal layers are then deposited for electrical signal routing. The CB layer (GDS 722) provides
contact to doped silicon (in the N++ and P++ regions), and it is required that it comprise only of
400 nm by 400 nm squares that are 800 nm apart. The CB layer is connected to higher metal
layers which are M1AM (GDS 710), M2AM (GDS 725) and MLAM (GDS 780) by use of vias.
These layers can be used together with metal escalators to bypass signals from one layer to the
other in order to make better use of the space on the chip, and so that metal wires don't cross.
MLAM is the final metal layer and it is used for metal contact pads which are squares of 60 µm
or larger. These contact pads are for connections to the outside world and are placed in a column
on the right side of the chip in order to ensure that the electrical and optical probes can test
simultaneously. Since the proposed design has multiple AIM PDK components which all require
Figure 6.2: AIM Waveguide, Etch and Doping Layers
66
electrical contacts (heater tuning voltage, photodetector connections as well as MZM
connections), multiple columns of contact pads will be required. The RF connections ([Ground
Signal Ground]) are placed close to the top and bottom edges of the chip.
The final chip layout is shown below in Figure 6.4. It is an earlier version of the proposed
PIC WROADM and consists of two MZMs for modulating the received RF signals, Y-junctions
for combining them, edge couplers for optical carrier inputs and for RoF signal drop and a three
ring resonator based filter (the six ring filter discussed in this report is yet to be incorporated)
followed by a photodetector. Electrical signals are routed via the M1 and M2 layers and metal
escalators are used to switch metal layers in order to route in regions where previous connections
already exist in one layer. Waveguide crossings are also present in order to avoid overlapping
waveguides and route optical signals across. Multiple columns of metal contact pads are used
due to the number of components that require a voltage supply, and as such packaging the device
will be required if all components have to be tested simultaneously, otherwise each column can
be tested individually. All components except for the MZMs are from the AIM PDK in this
design. The MZMs are based upon the design in Section 5.5. The chip is DRC clean which
means that the GDSII file was run through a Mentor Graphics Calibre design rule check and
came out free of errors, and hence can be fabricated with no issues.
Figure 6.3: AIM Metal Layers
67
Lastly, a breakdown of the losses for the longest route in the chip is provided below in
Table 6.1. The longest route consists of the path going from one edge coupler, through an MZM
followed by the filtering stage and on to the photodetector. The calculation has been updated
with the values for six ring resonators rather than three, as well as AIM PDK MZMs instead of
custom ones. The length of the route is ~ 5.58 mm using a generous (over)estimate of three chip
lengths for erring on the side of caution. The propagation loss is generally 3 dB/cm or 0.3
dB/mm which leads to a loss of -1.7 dB. This is the only loss that has not been accounted for in
Lumerical.
Parameter Loss (dB)
Edge Coupler Insertion Loss 3
Waveguide Crossing Insertion Loss (x2) 0.5
AIM MZM Insertion Loss 3
Y-Junction Insertion Loss (x2) 1
AIM Tunable Ring Resonator Insertion Loss (x6) 1.5
Propagation Loss 1.7
Aging 1
PDL 0.5
Margin 3
Total 15.2 Table 1: Carrier Switching Loss Budget
Figure 6.4: PIC WROADM DRC Clean Chip
68
The loss breakdown for the optical add route is given below (the propagation loss is
scaled to half of the previous case):
Parameter Loss (dB)
Edge Coupler Insertion Loss 3
Waveguide Crossing Insertion Loss (x2) 0.5
Y-Junction Insertion Loss (x1) 0.5
AIM Tunable Ring Resonator Filter Insertion
Loss (x6)
1.5
Propagation Loss 0.85
Aging 1
PDL 0.5
Margin 3
Total 10.85 Table 2: Optical Add Loss Budget
Finally, the loss breakdown for the optical drop route is as follows:
Parameter Loss (dB)
Edge Coupler Insertion Loss (x2) 6
Waveguide Crossing Insertion Loss (x3) 0.75
AIM MZM Insertion Loss 3
Y-Junction Insertion Loss (x1) 0.5
AIM Tunable Ring Resonator Filter Insertion
Loss (x6)
1.5
Propagation Loss 0.85
Aging 1
PDL 0.5
Margin 3
Total 17.1 Table 3: Optical Drop Loss Budget
Since the maximum power loss through the chip is ~ -17 dB (by a generous estimate), if a
CW laser source with +10 to +12 dBm of launch power (which is not unheard of) is used the
power at the output of the route through the chip will be -7 dBm to -5 dBm, which means that
optical amplification might not be necessary which is a big plus.
69
7. Conclusion
The era of 5G demands low cost, small device footprint and energy efficient building blocks to
enable millimeter-wave technology in order to meet the increasing bandwidth demands of the
population. This report outlines the step-by-step design, as well as investigating key design
issues and physical effects involved in realizing such a building block: the wireless
reconfigurable optical add-drop multiplexer (WROADM) using silicon photonics that leverages
the mature CMOS foundry processes already established for electronic chips in order to harness
lightwave signals. The PIC WROADM is proposed as a device that is capable of accepting mm-
Wave signals and modulating optical carriers with them in a fashion similar to Radio-over-Fiber
on a photonic chip. The WROADM is capable of optical signal processing functionality whereby
it can add/drop and filter signals as well as switch RF carriers in the optical domain. It performs
these tasks via components such as Mach-Zehnder Modulators, tunable silicon microring
resonators as well as photodetectors. The design of these components is essentially followed
from the ground-up, starting with rigorous Finite-Difference Time-Domain simulations in order
to accurately describe the behavior of electric and magnetic fields as they propagate through the
component, followed by the creation of compact models using S-matrices that enable us to
translate the FDTD simulations into individual component blocks that can be linked together in
Lumerical INTERCONNECT and rapidly simulated over a broad frequency range. In
INTERCONNECT, the components making up the WROADM are individually characterized,
yielding silicon phase shifter based depletion modulators with extinction ratios of ~ -17 and -23
dB and product of 0.647 V•cm, as well as a filtering stage that consists of two sets of 3
microring resonators cascaded together in order to provide a tunable, flat bandpass filter with a
3-dB bandwidth of 65 GHz and out of band rejection from -20 dB to as low as -50 dB. The
70
components used are obtained from the AIM Photonics foundry Process Design Kit (PDK) to
ensure manufacturability/fabrication of the chip as well as guarantee of minimal process
variation and high accuracy. Then, the RF carrier switching characteristic of the overall system is
observed using 14 GHz and 28 GHz mm-Wave signals modulated with NRZ-OOK format. Also
the Bit Error Ratio (BER) vs. Power (dBm) attenuation characteristic is observed for an ideal
case where the modulated signals are sent directly to the photodetector and bypass the chip and
also the eye diagrams for the 0 dB attenuation case are observed. This culminates in a DRC
clean, ready-to-fabricate chip layout designed in KLayout that has a floorplan of 1866 µm by
934 µm, and an optical loss budget (with margin) of 15.2 dB for the carrier switching, 10.85 dB
for the optical add function and 17.1 dB for the optical drop function.
Looking to the future, this report will serve to pave the way for further advancements in
the design of the WROADM, since there is definitely much more to study about the chip in terms
of determining its overall transfer function, both simulation-wise and experimentally. The error
free operation of the chip can be improved/maintained if optical carrier launch powers of 10
dBm are used instead of 0 dBm. Similarly, the incorporation of SOAs (semiconductor optical
amplifiers) might be necessary to offset the losses that the sidebands experience on chip. A
narrower filter passband will allow the central carrier and one neighboring sideband to be filtered
instead of both simultaneously. Also, techniques to get rid of unwanted harmonic terms must be
implemented in order to avoid noise being superimposed with the signal of interest. This report
has served not only to outline the design of the PIC WROADM and its constituents, but more
fundamentally as a guide towards familiarizing oneself with an array of different software
programs and techniques necessary to design a photonic chip, and hopefully it will continue to be
a repository of information for future generations of the WROADM chip.
71
8. References
[1].Kilper, D.C, et al. “Power Trends in Communication Networks.” IEEE Journal of
Selected Topics in Quantum Electronics, vol. 17, no. 2, 2011, pp. 275–284.,
doi:10.1109/jstqe.2010.2074187.
[2].Bonomi, Flavio, et al. “Fog Computing and Its Role in the Internet of Things.”
Proceedings of the First Edition of the MCC Workshop on Mobile Cloud Computing -
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