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Radio Frequency CMOS Transmitter
Frequency Modulation in Ultra-Wideband
Leonel Severino de Almeida
Dissertation for obtaining the degree of Master in
Electrical and Computer Engineering
Jury
President: Prof. Doutor Marcelino Bicho dos Santos
Advisor: Prof. Doutor Jorge Manuel dos Santos Ribeiro Fernandes
Co-Advisor: Doutor Miguel Andrade Martins
Members: Prof. Doutor José António Beltran Gerald
April 2012
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To my parents, girlfriend, sister and brother.
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Acknowledgements
This work represents a final step in the academic journey that not also grant me the skills and
competences to face the professional life but shaped my personality as individual, through all
the difficulties, victories and companionship that it contains. Many were the people that helped
me to reach this step and for this, I would like to show them all my gratitude and appreciation in
this work.
First of all, I would like to thank Professor Jorge Fernandes and Doutor Miguel Martins for
accepting me under their supervision along the development of this work. It was a privilege and
a pleasure to work in this scientific project under their guidance and to have at my disposal all of
their knowledge and expertise.
Also, I would like to acknowledge all the colleagues and friends, Luís Costa, David Correia,
Hugo Gonçalves,Taimur Kuntz and Rui Duarte for the great environment, companionship and
team spirit I encountered and for the opportunity to learn and being part of such skilled team as
“Circuitos Analógicos e Mistos”.
I would like to thank the INESC-ID (Instituto de Engenharia de Sistemas e Computadores –
Investigação e Desenvolvimento), IST (Instituto Superior Técnico) and FCT (Fundação para a
Ciência e Tecnologia) that provided the necessary conditions to realize this work and supported
this work through SCOMagNO: Frequency Synchronization of a CMOS RF Oscillator by a
Magnetic Nano-Oscillator Based On Spin Transfer Torque”, Apr 2011-Mar 2014, PTDC/CTM-
NAN/112672/2009.
I want to thank all my closest friends, not only the ones I met in IST but also those that always
have been my companions, Ricardo Martins, Pedro Martins, Pedro Araújo, Carlos Fernandes,
Luis Afonso, Dinis Bucho, Nuno Bernardo and Luís Ferreira for providing me support,
memorable moments and fellowship.
To my family, my parents Fernanda and Baltazar, a special thanks for all the unconditional love
and support and for being such a life example to me. To my sister Ana and my “brother” Hugo,
also a special thanks for all the love, advices and support.
And at last, but not the least, to my dearest love Corine, that always gave me strength and
always believed in me, even in the most difficult times. Without her love, unconditional support
and patience I’m certain that fulfilling this task would be much more difficult. A very special
thanks.
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Abstract
In order to allocate the growth of different types of RF devices, a continuous management and
usage improvement of the available spectrum is needed. Also, higher data rates, low power
consumption and smaller circuits are among users, and therefore industry, top demands.
The Ultra-Wideband concept is focused in sharing already allocated bands, some of them
proprietary, without causing interference to existing narrowband applications, while allowing
simultaneously high data transmission rates.
This work describes the implementation of a radio frequency CMOS transmitter operating
around 8 GHz, based in a constant-envelope frequency-domain approach called Frequency
Modulation Ultra-Wideband (FM-UWB) and implemented using a standard 0.13 µm CMOS
technology. The two main transmitter circuits are based on a relaxation oscillator topology,
since it is inductorless, it is wideband tunable and provides huge area savings. In this type of
oscillator a triangular signal is also present in the circuit which can be used advantageously in
this type of transmitter.
With the help of a simple control circuit, it is possible to obtain an FSK oscillator that generates
a triangular signal with two different oscillation frequencies, controlled by the transmitted data.
Also, using only MOS Varactors, it was implemented a VCO capable of achieving 1 GHz of
tuning range bandwidth with an operation frequency around 8 GHz.
The emitter and the receiver are studied at a high-leve,l being the transmitter designed at
transistor level and layout. Simulations results are very satisfactory, proving the potential of the
designed circuit in fulfilling the desired goals for obtaining a low-power and low-cost transmitter,
capable of achieving high data rates and sharing already occupied RF transmission bands.
Keywords FM-UWB, CMOS Transmitter, Low-Power, Low-Cost, Low-Area.
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Resumo
A fim de garantir espaço para o crescimento de diferentes tipos de dispositivos de RF, é
necessário garantir uma melhoria contínua do uso e da gestão do espectro de frequências.
Também, o maior débito de dados, o baixo consumo de energia e os circuitos com menores
dimensões estão entre as principais exigências dos utilizadores e desta forma também da
indústria.
O conceito de Ultra-Wideband está focado na utilização de bandas já alocados por outros tipos
de tecnologia, algumas reguladas, e sem causar interferências com as aplicações de banda
estreita, permitindo simultaneamente taxas de transmissão de dados a alta velocidade.
Este trabalho descreve a implementação de um transmissor de radiofrequência em tecnologia
padrão CMOS de 0.13 µm e a funcionar a 8 GHz, tendo como abordagem de base a utilização
de uma máscara no domínio da frequência. Esta técnica é denominada de Frequency
Modulation Ultra-Wideband (FM-UWB) Os dois principais circuitos do transmissor são
baseados na topologia de um oscilador de relaxação, uma vez que esta não tem elementos
indutivos, permite sintonização em banda larga e economizar área de circuito. Com este tipo de
oscilador é possível obter um sinal triangular que pode ser usado como uma vantagem neste
transmissor.
Com a ajuda de um circuito de controlo simples, é possível obter um oscilador FSK capaz de
gerar um sinal triangular com duas frequências de oscilação diferentes, controladas pelos
dados a serem transmitidos. Além disso, usando apenas varactores MOS, foi implementado um
VCO capaz de atingir 1 GHz de largura de banda e a operar a 8 GHz.
É realizado um estudo de alto nível ao emissor e ao receptor sendo o transmissor
implementado ao nível do transístor e respectivo layout. Os resultados das simulações do
transmissor foram bastante satisfatórios, provando o potencial do circuito projectado para
cumprir as metas desejadas. Assim, foi possível obter um transmissor de baixa potência e de
baixo custo, capaz de atingir altas taxas de transmissão de dados utilizando bandas de
transmissão de RF já ocupadas.
Palavras Chave FM-UWB, Transmissor CMOS, Baixa Potência, Baixo Custo, Área Reduzida.
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Table of Contents
Acknowledgements iv
Abstract vi
Resumo viii
Table of Contents x
List of Figures xiv
List of Tables xviii
List of Acronyms xx
List of Symbols xxii
1. Introduction 1
1.1. Background and Motivation 2
1.2. Thesis Organization 3
1.3. Main Contributions 4
2. Ultra-Wideband Technology 5
2.1. Introduction 6
2.2. UWB Overview 6
2.3. Most Common UWB Implementation Types 9
2.3.1. Impulse Radio UWB 9
2.3.2. Multi-Band Orthogonal Frequency Division Multiplexing 10
2.3.3. Frequency Modulation UWB 11
2.4. Frequency Modulation in UWB Perspective 12
2.5. Conclusions 17
3. FM-UWB Circuit Architecture 19
3.1. Introduction 20
3.2. Transmitter 20
3.2.1. Transmitter Characteristics 21
3.2.2. Subcarrier Modulation 22
3.2.3. Carrier Modulation 23
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3.2.4. Simulation 23
3.3. Transmission Channel Model 27
3.3.1. Attenuation 27
3.3.2. Thermal Noise 28
3.3.3. Simulation and Signal-to-Noise Ratio 31
3.4. Receiver 32
3.4.1. Preamplifier 34
3.4.2. Spectral Shaping Band-Pass Filter 35
3.4.3. Signal Rectifier 37
3.4.4. Low-Pass Filter 37
3.4.5. Comparator 38
3.5. Digital Decoder 39
3.6. Conclusions 40
4. Circuit Implementation 41
4.1. Introduction 42
4.2. Relaxation Oscillators 42
4.2.1. High-Level Model 43
4.2.2. Circuit Implementation 44
4.3. Frequency-Shift Keying Implementation 46
4.3.1. Control Circuit 48
4.3.2. Simulation Results 48
4.4. Voltage Controlled Oscillator Implementation 51
4.4.1. MOS Varactors 53
4.4.2. Simulation Results 55
4.5. Current Sources 58
4.6. Conclusions 59
5. Transmitter Simulation and Results 61
5.1. Introduction 62
5.2. Transmitter Schematic and Simulations 62
5.2.1. Circuit Schematic 62
5.2.2. Graphical Results 64
5.2.3. Simulation Results using Pads, Bonding Wires, ESD protection and
Antenna 67
5.2.4. Simulation Results with High-level Models for Transmission Channel,
Receiver and Digital Decoder 72
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5.2.5. Corners 74
5.3. Layout Design 76
5.4. Conclusions 78
6. Conclusion and Future Work 79
6.1. Conclusion 80
6.2. Future Work 81
References 83
Annex 87
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List of Figures
Figure 2.1 – Comparison between narrowband and ultra-wideband bandwidths. ........................ 6
Figure 2.2 – FCC spectral mask for indoor (a) and outdoor (b) UWB transmission ..................... 7
Figure 2.3 – UWB intended bands for communication in different regions. .................................. 8
Figure 2.4 – Example of an IR-UWB pulse shape. ....................................................................... 9
Figure 2.5 – IR-UWB impulse train.............................................................................................. 10
Figure 2.6 – Proposed MB-OFDM frequency band plan. ............................................................ 11
Figure 2.7 – Spectrum of a FM-UWB signal and an unmodulated RF carrier at 4 GHz. ............ 11
Figure 2.8 – VCO transfer function.............................................................................................. 13
Figure 2.9 – Spectral power density 𝑆𝑋𝐹𝑀(𝑓) of the cosine carrier signal. .................................. 14
Figure 2.10 – PDF for the modulating signal m(t). ...................................................................... 15
Figure 2.11 – Transmission bandwidths for narrowband and wideband FM signals .................. 16
Figure 2.12 – Power spectral density 𝑆𝑋𝐹𝑀(𝑓) of the wideband FM signal 𝑋𝐹𝑀(𝑡). ..................... 16
Figure 2.13 – Modulating signal 𝑚(𝑡), triangular waveform with period T. ................................ 17
Figure 3.1 – Time-domain view of data d(t), subcarrier m(t), and UWB signal XFM(t). ................ 20
Figure 3.2 – FM-UWB high level modulator schematic. .............................................................. 21
Figure 3.3 – Subcarrier generator circuit schematic. .................................................................. 22
Figure 3.4 – Simulation results of the digital raw data 𝑑(𝑡) and the subcarrier signal 𝑚(𝑡). ...... 23
Figure 3.5 – 𝑋𝐹𝑀(𝑡) power spectral density expression and DFT parameters in the calculator. 24
Figure 3.6 – Unmodulated carrier signal and its respective power spectral density. .................. 25
Figure 3.7 – Simulation of subcarrier signal 𝑚(𝑡), carrier signal 𝑋𝐹𝑀(𝑡) and carrier power
spectral density 𝑆𝑋𝐹𝑀(𝑓) ......................................................................................... 26
Figure 3.8 – Schematic circuit for the transmission channel model. ........................................... 27
Figure 3.9 – Noise signal and Gaussian probability density function .......................................... 29
Figure 3.10 – Simulation of noise signal 𝑁(𝑡) and respective power spectral density 𝑁0. .......... 31
Figure 3.11 – Simulation of signal attenuated 𝑋′𝐹𝑀(𝑡), noise signal 𝑁(𝑡) and signal 𝑌𝐹𝑀(𝑡). ... 32
Figure 3.12 – Preamplifier high level schematic. ........................................................................ 33
Figure 3.13 – Demodulator high level schematic. ....................................................................... 33
Figure 3.14 – Signal at the input 𝑌𝐹𝑀(𝑡) and output 𝑋′′𝐹𝑀(𝑡) of the preamplifier circuit .............. 34
Figure 3.15 – Band-pass filter response to convert FM into AM signal ...................................... 35
Figure 3.16 – Subcarrier signal 𝑚(𝑡) and AM signal 𝑋𝐴𝑀(𝑡) resulting from the filter conversion36
Figure 3.17 – Signal after rectification 𝑋𝐴𝑀 𝟐(𝑡). ........................................................................ 37
Figure 3.18 – Low-pass filter output signal 𝑌𝐴𝑀(𝑡). ..................................................................... 38
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Figure 3.19 – Comparator input signal 𝑌𝐴𝑀(𝑡) (red signal) and output signal 𝑌′𝐴𝑀(𝑡) (blue signal).
................................................................................................................................ 38
Figure 3.20 – Digital decoder circuit schematic. ......................................................................... 39
Figure 3.21 – Original data 𝑑(𝑡), BDC signal 𝑌′𝐴𝑀(𝑡) and recovered data 𝑑′(𝑡). ..................... 39
Figure 4.1 – Relaxation oscillator block diagram. ....................................................................... 43
Figure 4.2 – Oscillator output waveforms VINT and VST. .............................................................. 43
Figure 4.3 – Relaxation oscillator implementation. ..................................................................... 44
Figure 4.4 – Integrator block and respective output waveform. .................................................. 44
Figure 4.5 – Schmitt-trigger block and respective transfer function. ........................................... 45
Figure 4.6 – Relaxation oscillator waveforms. ............................................................................ 46
Figure 4.7 – Integrator waveform 𝑣𝐶0 (corresponding to a capacitance C) and 𝑣𝐶1
(corresponding to a capacitance 2C). ..................................................................... 47
Figure 4.8 – Relaxation oscillator with variable capacitance. ..................................................... 47
Figure 4.9 – Control circuit schematic. ........................................................................................ 48
Figure 4.10 – Relaxation oscillator with variable capacitance and control circuit. ...................... 49
Figure 4.11 – Simulation result for the signal 𝑣𝐶, for a data signal 𝑑(𝑡) with transitions between
“0” and “1”. .............................................................................................................. 50
Figure 4.12 – Simulation result for the signal 𝑣1 and CLK, for a data signal 𝑑(𝑡) with transitions
between “0” and “1”. ................................................................................................ 50
Figure 4.13 – Simulation result for the signal DATA and comparison with the original data signal
𝑑(𝑡) with transitions between “0” and “1”. ............................................................... 51
Figure 4.14 – Proposed topology for VCO with MOS varactors and differential tuning signals. 52
Figure 4.15 – Cross section of a conventional NMOS varactor (in depletion; left) and the
generally assumed model (right). The dashed line indicates the border of the
depletion region. ..................................................................................................... 53
Figure 4.16 – Typical measured small-signal capacitance characteristic 𝑣𝐺 of a NMOS varactor
(bottom) and the relevant lumped elements (top) at zero tuning voltage. .............. 54
Figure 4.17 – Typical measured small-signal capacitance characteristic of a conventional NMOS
varactor at various tuning voltages. ........................................................................ 54
Figure 4.18 – PMOS and NMOS varactor inversion-mode capacitance characteristic 𝐶𝐺. ........ 55
Figure 4.19 – Three terminal MOS varactor circuit schematic. ................................................... 56
Figure 4.20 – VCO frequency variation resulting from NMOS varactor excitation (1) and PMOS
varactor excitation (2). ............................................................................................ 57
Figure 4.21 – VCO output 𝑣𝑜𝑢𝑡1 with a frequency of 7.87 GHz. .................................................. 57
Figure 4.22 – Current source circuit schematic. .......................................................................... 58
Figure 5.1 – Transmitter circuit schematic. ................................................................................. 62
Figure 5.2 – Signal 𝑑(𝑡) with transitions between “0” and “1” and simulation result for signal
DATA....................................................................................................................... 64
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Figure 5.3 – Simulation result for the differential output 𝑉𝑡𝑢𝑛𝑒1 and 𝑉𝑡𝑢𝑛𝑒2 produced by the FSK
oscillator. ................................................................................................................. 65
Figure 5.4 – Simulation result for the signal 𝑣𝐶 (𝑉𝑡𝑢𝑛𝑒1 − 𝑉𝑡𝑢𝑛𝑒2). ................................................ 65
Figure 5.5 – Simulation result showing the synchronism between the signal DATA and
signal 𝑣𝐶 . ................................................................................................................ 66
Figure 5.6 – Simulation result for signal 𝑉𝑜𝑢𝑡1 (including its instantaneous frequency). ............. 66
Figure 5.7 – Common drain stage schematic. ............................................................................ 67
Figure 5.8 – ESD protection schematic. ...................................................................................... 68
Figure 5.9 – Bonding wire model schematic. .............................................................................. 69
Figure 5.10 – Antenna model schematic. .................................................................................... 69
Figure 5.11 – Simulation result for signal 𝑉𝑜𝑢𝑡1´ and Simulation result for signal 𝑉𝑎𝑛𝑡_𝑖𝑛. ............ 70
Figure 5.12 – Simulation result for signal 𝑉𝑎𝑛𝑡_𝑜𝑢𝑡. ...................................................................... 70
Figure 5.13 – Simulation result for signal 𝑉𝑎𝑛𝑡_𝑜𝑢𝑡 instantaneous frequency and DFT. ............... 71
Figure 5.14 – Simulation result for signal Vchannel and signal Vpre_amp. ......................................... 72
Figure 5.15 – Simulation result for signal Vam_pulse and signal Vbdc. ............................................ 73
Figure 5.16 – Simulation result for signal DATA_Rec and comparison with the signals 𝑑(𝑡) and
DATA....................................................................................................................... 73
Figure 5.17 – FM-UWB transmitter layout without pads, ESD protections and output buffers. .. 77
Figure 5.18 – FM-UWB transmitter layout with pads, ESD protections and output buffers. ....... 77
Figure A1 – Verilog-A code for the attenuator block. .................................................................. 88
Figure A2 – Verilog-A code for white noise Gaussian source. .................................................... 88
Figure A3 – Verilog-A code for band-pass biquadratic filter. ...................................................... 88
Figure A4 – Vout1' instantaneous frequency Vs DATA signal. ...................................................... 89
Figure A5 – Vout1' Power Spectral Density. .................................................................................. 89
Figure A6 – Circuit schematic with the transmitter core, buffers, ESD protections, pads, bonding
wires and antennas. ................................................................................................ 90
Figure A7 – FM-UWB transmitter layout. .................................................................................... 91
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List of Tables
Table 3.1 – Transmitter characteristics ....................................................................................... 21
Table 3.2 – Simulation results for maximum power spectral density for different subcarrier
frequencies, transmission bands and carrier amplitudes. ......................................... 25
Table 3.3 – Attenuation values for different distances using the carrier frequency of 7.9 GHz. . 28
Table 3.4 – Noise power values for different bandwidths. .......................................................... 30
Table 4.1 – Parameters used in the simulation of the relaxation oscillator with variable
capacitance. .............................................................................................................. 49
Table 4.2 – Parameters used in the VCO simulation. ................................................................. 56
Table 4.3 – Parameters used in the FSK current source. ........................................................... 58
Table 4.4 – Parameters used in the VCO current source. .......................................................... 59
Table 5.1 – Parameters used in the transmitter implementation and simulation. ....................... 63
Table 5.2 – Pads parameters. ..................................................................................................... 68
Table 5.3 – ESD protection parameters. ..................................................................................... 68
Table 5.4 – Bonding wire parameters. ........................................................................................ 69
Table 5.5 – Antenna parameters. ................................................................................................ 69
Table 5.6 – Transmitter simulation results. ................................................................................. 72
Table 5.7 – PVT corners simulated. ............................................................................................ 74
Table 5.8 – Corners simulation results. ....................................................................................... 75
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List of Acronyms
BDC Binary digital code BFSK Binary frequency-shift keying DFT Discrete Fourier transform EIRP Effective isotropic radiated power FDMA Frequency division multiple access FCC Federal Communication Commission FM Frequency modulation FSK Frequency-shift keying IR Impulse radio LOS Line of sight MB Multi-band MOS Metal–oxide–semiconductor OFDM Orthogonal frequency division multiplexing PDF Probability density function PLL Phase locked loop PSD Power spectral density PM Phase modulation RF Radio frequency RT Radio Technology SNR Signal-to-Noise ratio UMC United Microelectronics Corporation UWB Ultra-wideband VCO Voltage controlled oscillator WGNS White Gaussian noise source
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List of Symbols
f Frequency
C Maximum channel capacity
B Channel bandwidth Carrier amplitude
Carrier instantaneous phase Carrier angular frequency
𝑓 Carrier frequency Frequency instantaneous deviation
Phase instantaneous deviation
𝑓 Frequency modulation index
Transmission band
Speed of light
Wavelength
𝑑 Distance
Planck constant
Boltzmann constant
Mean value
Standard deviation Attenuation
Quality factor
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1
1. Introduction ____________________________________________________
2
1.1. Background and Motivation
Nowadays we live surrounded of all kind of communication devices, since the most common ones, like
televisions and radios, to mobile phones and computers, being the fast access to information now a
basic demand.
In the past decade, result of the explosive growth of technological devices, short-range wireless
networks management became a major priority. The necessity of granting communication and the
capability to share information between almost all devices created a demand for RF bandwidths and
the increase of transmission data speeds. At circuit level, battery operated, mass market devices,
require the development of compact circuits with minimum area and cost, with low power consumption
and lower voltage supply, and with high degree of integration, leading to an increase of the study,
research and development both in the academic environment and in the semiconductors and
communications industries.
Towards short-range applications, Ultra-Wideband Radio Technology can drive the potential solutions
for many of the identified problems in the areas of spectrum management and radio systems
engineering. The novel and unconventional approach underlying the use of modern UWB is based on
the optimally sharing of the existing radio spectrum resources rather than looking for still available but
possibly unsuitable new bands [1].
This thesis aims the development of a radio frequency CMOS transmitter using a 0.13 µm CMOS
technology based in a constant-envelope frequency-domain approach called Frequency Modulation
Ultra-Wideband (FM-UWB). The purpose of this transmitter is mainly portable non-critical applications
and its key features are minimum area and cost, low power consumption and low voltage supply. FM-
UWB is being proposed as a standard for biomedical applications.
The FM-UWB transmitter implemented is based in two main blocks, a frequency-shift keying oscillator,
controlled by the data, which is the key element to generate a sub-carrier signal, and a voltage
controlled oscillator used to produce the carrier signal. In order to minimize the area size and the
complexity of the transmitter, both oscillators are inductorless circuits, leading to a low-cost and a low-
power implementation.
3
1.2. Thesis Organization
This work is organized in 6 chapters. Besides the introduction, in Chapter 2, is provided a global
overview of the UWB technology and its primary characteristics, also a brief introduction to the most
relevant UWB implementation schemes is made, emphasizing the frequency modulation technique in
an UWB perspective.
In Chapter 3, a study about the concepts involving the FM-UWB transmission technique using high
level circuit blocks is performed. This high level study allows the evaluation of the link budget,
including the theoretical modulation-demodulation concepts, and the definition of each block
characteristics.
In Chapter 4, the implementation of each block composing the circuit, and respective simulations are
presented. Both blocks that compose the transmitter, are based in the single relaxation oscillator
topology, where different characteristics of this specific circuit are adapted to fulfill the transmitter
requirements. The relaxation oscillator, due to its simple architecture and operation mode are a major
advantage to minimize the overall area and reduce power consumption.
In Chapter 5, simulations results for the transmitter and also the final circuit parameters are studied
and presented. Besides the circuit core simulation, it is included an overall demonstration of the circuit
performance in extreme conditions and the simulation results, using already an output buffer, bonding
wires, pads and ESD protections, since it would likely affect the final result. As final step it is produced
and presented the circuit layout along with an evaluation of the die size area.
Finally, in Chapter 6, conclusions and future work suggestions are presented.
In the final annex further simulation results and other relevant complementary information are
presented.
4
1.3. Main Contributions
The main original contributions of the work are:
1. The use of a relaxation oscillator topology to produce an almost triangular waveform signal
with the capability of changing its fundamental frequency according with a certain data signal.
The result is a significant reduction of the circuit complexity with a strong impact on the
reduction of die area and power consumption, when comparing to the common use of a DDS to
produce the same result.
2. The use of a relaxation oscillator topology to develop a VCO (operating at 8 GHz) capable of
achieving more than 1 GHz of tuning range bandwidth based only on MOS Varactors, allowing
at the same time, a VCO control through a differential input signal. This result, besides the
pretended area size and power consumption reduction, allows a major simplification to the
circuit when comparing to other VCO’s with similar tuning ranges.
5
2. Ultra-Wideband
Technology ____________________________________________________
6
2.1. Introduction
The primary objective of this chapter is to provide a global overview of UWB technology, where its
main characteristics are discussed in section 2.2. In section 2.3, the three most common schemes of
implementation are briefly described, while section 2.4 is entirely focused in Frequency Modulation
from a UWB point of view, once this modulation scheme is the theoretical basis of this work.
2.2. UWB Overview
In radio frequency communication systems, modern Ultra-Wideband is used to describe signals with a
minimum bandwidth of 500MHz, (for operation frequencies above 3.1GHz) [2], by comparison with
traditional narrowband communication systems, with bandwidths of a few kHz, as shown in figure 2.1.
Figure 2.1 – Comparison between narrowband and ultra-wideband bandwidths.
UWB signals are mostly low-power and short-range signals (usually no more than a few meters). Due
to its very wide bandwidth, this technology can provide great robustness against interference and
frequency-selective multipath propagation conditions, leading to less signal attenuation, especially
when transmitting in indoor environments [3], [4].
7
Two other characteristics that make this technology attractive are low RF transmission power and low
power-spectral density, imposed by the regulating authorities. Therefore, it is possible to develop
circuits with very low power consumption, and to share RF bandwidths previous allocated to other
technologies, since the interference between signals is practically inexistent [1], [3].
Sharing already occupied spectrum bands, rather than moving to new wave frequencies, is a solution
that UWB can provide for RF spectrum management area, justifying the growing investigation in this
field [1].
In order to allow UWB transmission in allocated bands, the signal power level must be very low,
therefore Federal Communication Commission authorized wideband signal format with a low Effective
Isotropic Radiated Power level of -41.3 dBm/MHz, creating this way a UWB signal mask that prevents
interference with other systems. This type of power level, (-41.3 dBm/MHz), create a new concept of
“noise floor” usage, since this power level is the same as that allowed for noise emissions of electronic
devices. The FCC spectral masks for indoor and outdoor transmissions are shown in figure 2.2 [4], [5].
Figure 2.2 – FCC spectral mask for indoor (a) and outdoor (b) UWB transmission [4].
The low radiated power allowed for UWB systems enables low DC power consumption, leading this
way to extended battery life and maybe to the possibility of using power harvesting circuits as energy
sources, creating really autonomous network nodes [1]. However, the inconvenient of reducing to
much the circuits’ transmission power consumption is that the range also decreases, thus, a trade-off
between transmitted power and range must be established.
Another important aspect that makes UWB as one of the future leading wireless communication
technologies is the high speed rates that can be achieved. If we take in consideration Shannon’s
channel capacity equation, presented in (2.1):
8
𝐶 2 (1 ) (2.1)
It is possible to conclude that the extended bandwidth (B) of UWB allows a great channel capacity (C)
on one hand, or on the other hand, it is possible to trade bandwidth for SNR, what grants to UWB
robustness against interference from other sources. Low SNR can lead to low-complexity circuits and
these to the possibility of achieving low cost circuits [4].
Nowadays, UWB communication systems are playing a large role in the development of new short-
range wireless networks, such as wireless personal area networks (WPANs), wireless body area
network (WBANs) and remote health monitoring (e-health) [1], [3]. The high speed rates provided by
UWB transmissions make this technology desirable for data transmission, especially for distribution of
music, video and sensor information. It is expected that UWB become a very attractive solution for the
future wireless communications and many other applications including logistics, security and military
applications, control of home appliances, search-and-rescue, family communications and supervision
of children [6].
Due to the emerging importance of UWB systems, this technology has already received legal
adoption, in what concerns to bandwidth use, by the competent regulatory authorities in many places
around the world like United States, Europe, Canada, and several countries in Asia, like Japan, China,
South Korea and Singapore. The intended bands for UWB use in these regions can be seen in figure
2.3 [5].
Figure 2.3 – UWB intended bands for communication in different regions [5].
9
2.3. Most Common UWB Implementation Types
The UWB definition does not specify any type of air interface or modulation scheme, so it is possible
to use many different techniques to create this type of signal [2]. In the further sections are explored
the UWB most common implementation techniques, Impulse Radio UWB (IR-UWB), Multi-Band
Orthogonal Frequency Division Multiplexing (MB-OFDM) and more recently Frequency Modulation
UWB (FM-UWB).
2.3.1. Impulse Radio UWB
UWB originally started with a technique of implementation called Impulse Radio Ultra-wideband (IR-
UWB). Nowadays, it is still one of the most used implementations. In conventional spread-spectrum
techniques, the signals are continuous-wave sinusoids that are modulated with a fixed carrier
frequency, while IR-UWB consists in creating a pulse wave or by generating a baseband envelope
impulse to modulate a sinusoidal wave, shifting the signal spectrum to higher frequencies (RF
frequencies) [2], [5], [6].
The short-duration pulses, (usually nanoseconds or picoseconds), will lead to wide energy spreading
acting like information carriers. Signals are intentionally made to have wider band than the necessary
to make them more noise-like, taking advantage of the UWB bandwidth available and respecting the
PSD limits imposed by FCC masks [4], [6], [7]. An example of IR-UWB pulse is illustrated in figure 2.4.
Figure 2.4 – Example of an IR-UWB pulse shape [4].
10
Transmitted data information is modulated into a sequence of pulses called pulse train, as illustrated in
figure 2.5. Pulse train results, therefore, on modulating pulses. When pulses are sent in regular
intervals, due to pulse repetition period, peaks of power limit the total transmit power. One method to
make the spectrum more noise like is to dither the signal, by adding a small random offset to each
pulse [4].
Figure 2.5 – IR-UWB impulse train [4].
One of the most important characteristics of IR is the fact that it is power efficient, what is a great
advantage, considering the context of the UWB technology. [6].
2.3.2. Multi-Band Orthogonal Frequency Division Multiplexing
Multi-band orthogonal frequency-division multiplexing is another method to implement UWB
transmission. This technique combines OFDM techniques with multi-band (MB) approach and it
consists on the use of multiple signals across the UWB spectrum with non-overlapping frequencies.
This way, it is possible to use the entire 7.5 GHz of spectrum allocated for UWB transmissions, using
several sub-bands with a -10 dB bandwidth of at least 500 MHz each. The information is distributed
into these several sub-bands and then transmitted, this also allows, that sub-bands can be treated
independently, bringing some advantages like less interference with other systems [4].
One of the proposals for the physical standard of future high speed WPAN uses MB-OFDM technique.
In this proposal, the spectrum between 3.1 and 10.6 GHz is divided into 14 bands with 528 MHz
bandwidth each that may be added or dropped depending upon the interference with other systems,
where only 13 bands are used to avoid interference between UWB and the existing IEEE 802.11a
signals. The three lower bands are used for the standard operation, which is mandatory, and the rest
of the bands are allocated for optional use or future expansions. In figure 2.6 is presented this
frequency band plan [4].
11
Figure 2.6 – Proposed MB-OFDM frequency band plan [4].
MB-OFDM continues to be the standard approach for high data-rate UWB transmissions and it is also
used in the most common narrowband communication systems. MB-OFDM systems can support bit
rates up to 480 Mbps, however this high-performance comes at the expense of great circuit complexity
and high power consumption [6], [8], [9].
2.3.3. Frequency Modulation UWB
This work will explore a technique of implementation for UWB systems called Frequency Modulation
Ultra-Wideband (FM-UWB). It consists on creating a constant-envelope in the frequency domain using
the well-known technique of frequency modulation. The basic principle of this technique is to use a
double FM scheme. First is applied a low-modulation index, using digital frequency-shift keying (FSK),
followed by high-modulation index analog FM, thus, a constant-envelope signal with flat spectrum and
a very wide bandwidth with step spectral roll-off is produced, as shown in figure 2.7 [2].
Figure 2.7 – Spectrum of a FM-UWB signal and an unmodulated RF carrier at 4 GHz [2].
12
One attractive advantage of using this approach is the possibility of accommodating multiple users. By
applying the FDMA technique, it is possible to assigning different subcarrier frequencies to different
users, a problem that is relevant to WPAN systems [3].
The receiver does not work with carrier or pulse synchronization. Acquisition is reduced to bit
synchronization, so no local oscillator or PLL circuits are required in the receiver. From a
synchronization point of view, the system will behave like a narrowband FSK system. The modulation
techniques are well known and easy to implement allowing low voltage supply [3]. Due to all these
factors, the circuit implementation for FM-UWB systems is potentially low power and low cost.
2.4. Frequency Modulation in UWB Perspective
Applying the basic principle of moving the signal spectrum from its original base band and centering it
in the carrier frequency allows carrying signal messages, (e.g., human voice or digital data), with
frequencies that could not be transmitted by radio antennas with current technology. This principle
allowed the appearance of radio communications.
Frequency modulation is a particular case of what is called sinusoidal carrier angle modulation. Angle
modulation consists in making the carrier signal instantaneous phase change linearly with a
modulating signal. This can be made by changing the signal’s phase (PM) or frequency (FM) [10].
Let’s consider only frequency modulation. The sinusoidal carrier can be represented by equation (2.2)
𝑋𝐹𝑀(𝑡) ( (𝑡)), (2.2)
where is the carrier amplitude (in Volts) and (𝑡) is the carrier instantaneous phase, that can be
represented by (2.3):
(𝑡) 𝑡 (𝑡) 𝑓 𝑡 (𝑡), (2.3)
The carrier frequency is 𝑓 , and (𝑡) represents the phase instantaneous deviation. To perform a
change in 𝑓 , it is necessary to define the frequency instantaneous deviation (𝑡) (2.4):
(𝑡) 1
2 (𝑡)
𝑡 𝑓 𝑚(𝑡), (2.4)
13
where 𝑓 is the frequency modulation index or the frequency deviation used to describe the maximum
instantaneous difference between an FM modulated frequency and the carrier’s frequency 𝑓 , and
𝑚(𝑡) is the modulating signal [10], [11]. At this point, it is possible to define the VCO transfer function
𝑉(𝑡) (2.5):
𝑉(𝑡) 1
2 (𝑡)
𝑡 𝑓 𝑓 𝑚(𝑡), (2.5)
The result is a linear function, where 𝑓 controls the slope and is proportional to the VCO gain. The
VCO function can be observed in figure 2.8.
Figure 2.8 – VCO transfer function.
Generate a FM signal with UWB characteristics means that the signal spectral power needs to fit in
the UWB requirements, a signal with minimum bandwidth around 500 MHz for frequencies of
operation above 3.1 GHz, and according to figure 2.2, an EIRP level of -41.3 dBm/MHz.
There are two major kinds of denomination for FM signals, based in its bandwidth, narrowband FM
and wideband FM [10]. FM-UWB makes part of the second type and it is important to analyze both
types to better understand all the concepts and approximations.
One of the most important aspects is the signal’s spectral power, so we will start by analyzing the
carrier’s PSD and the modulating signal’s amplitude probability density function (PDF) [12].
The carrier signal 𝑋(𝑡) is ideal and given by (2.6), and its respective power spectral density 𝑆 (𝑓)
given by (2.7):
𝑋(𝑡) ( 𝑡), (2.6)
𝑆𝑋 (𝑓) 𝐴
[ (𝑓 𝑓 ) (𝑓 − 𝑓 )], (2.7)
14
Power spectral density 𝑆 (𝑓) is a pair of Dirac delta function’s (𝑓) with amplitude ( 𝟐)𝟐, that
represents the sine wave power, (cosine wave power in this case), condensed in a single frequency,
in this case the carrier frequency 𝑓 , as shown in figure 2.9.
Figure 2.9 – Spectral power density ( ) of the cosine carrier signal.
The signal’s amplitude PDF gives us the shape of the spectral power density of a frequency
modulated sinusoidal carrier 𝑆𝑋𝐹𝑀 (𝑓). It is more intuitive to understand if we think that the Dirac delta
function’s (𝑓), (representing the spectral power of the sine wave), are shifted to other frequencies
according to a certain probability.
To maximize the transmitted power, the spectral power density 𝑆𝑋𝐹𝑀 (𝑓) should have the same shape
as the FCC spectral mask for the operation bandwidth, meaning that the signal’s amplitude PDF
should be a continuous uniform distribution [12], (in this particular case, the Dirac delta function’s (𝑓)
are being shifted to other frequencies with equal probability).
The PDF ( ) of the signal 𝑚(𝑡), with amplitude between [− ] Volt, represented in figure 2.10,
follows a continuous uniform distribution and its integral is equal to one (∫ ( )𝑑 1).
15
Figure 2.10 – PDF for the modulating signal m(t).
The definition of narrowband FM signal is given by (2.8) [10]:
| (𝑡)| | 𝑓 ∫ 𝑚( )𝑡
𝑑 | 1, (2.8)
leading to the following transmission band (2.9) where is the bandwidth for the modulating
process PSD [10]:
, (2.9)
For wideband FM signal we have (2.10) [10]:
𝑓 𝑓 , (2.10)
In this condition we are able to apply what is known by quasi-stationary approximation leading to a
different transmission band (2.11) [10]:
𝑓 |𝑚(𝑡)| 𝑎 , (2.11)
In figure 2.11 both transmission bands are represented to better understand the difference between
them.
16
Figure 2.11 – Transmission bandwidths for narrowband and wideband FM signals
By knowing the carrier PSD 𝑆𝑋 (𝑓), the modulating signal’s PDF ( ), and the respective
transmission band it is possible to obtain the wideband FM signal’s PSD 𝑆𝑋𝐹𝑀 (𝑓)(2.12),
represented in figure 2.12 [10].
𝑆𝑋𝐹𝑀 (𝑓) 𝐴
[ (
) (
)], (2.12)
Figure 2.12 – Power spectral density 𝑆𝑋𝐹𝑀 (𝑓) of the wideband FM signal 𝑋𝐹𝑀(𝑡).
To obtain a spectral power density with a continuous uniform distribution we need to a have a signal
with a linear variation, this means that the modulating signal 𝑚(𝑡) must be a triangular waveform, as
shown in figure 2.13.
17
Figure 2.13 – Modulating signal 𝑚(𝑡), triangular waveform with period T.
2.5. Conclusions
In this chapter an overview over the main UWB technology characteristics was made. The bandwidth
usage and the spectral power limitations imposed by the regulatory authorities were referred. Also, the
main implementation types were described and a more detailed description of FM-UWB was
presented, once this was the scheme chosen to develop the transmitter.
18
19
3. FM-UWB Circuit
Architecture ____________________________________________________
20
3.1. Introduction
The circuits here explained are implemented with ideal components making this a high level circuit
description. These circuits are the first approach to validate the concepts involved in FM-UWB
transmission technique. Also, to verify the theoretical demodulation concepts and the transmitter
correct operation, it was necessary to develop the transmission channel and the receiver high level
circuits. All the circuits and simulations were developed and performed using the Cadence® program.
This chapter is divided in three main sections, concerning the transmitter, the transmission channel
and the receiver, where the functioning of the major blocks are explained, followed by the simulation
analysis. In the transmitter section, it is also presented the desired characteristics.
3.2. Transmitter
The modulator described was created to implement the previous explained FM concepts. FM-UWB
use a double FM scheme, which means that first, is necessary to modulate a subcarrier signal 𝑚(𝑡)
using the data to be transmitted 𝑑(𝑡) and then use this subcarrier signal to modulate the carrier
signal 𝑋𝐹𝑀(𝑡), as illustrated in figure 3.1 [2].
Figure 3.1 – Time-domain view of data d(t), subcarrier m(t), and UWB signal XFM(t) [2].
21
In figure 3.2 is presented the high-level modulator scheme with the two basic blocks involved, the
subcarrier generator and the carrier modulator (VCO).
Figure 3.2 – FM-UWB high level modulator schematic.
3.2.1. Transmitter Characteristics
Since the beginning of this work there were some specifications that needed to be fulfilled. The
transmission power allowed is clearly the most important, since it is a requirement to the signal
transmission. The range with line of sight (LOS), the RF centre frequency as well as the raw data rate
were chosen taking in consideration some aspects, such as the final applications and the compatibility
for transmission in different regions of the globe.
The carrier’s amplitude, the RF bandwidth and the subcarrier frequencies are other transmitter
characteristics that are not specifications, but result from the simulations made to respect those
specifications.
All the referred characteristics are presented in table 3.1.
Transmission power ≈ -42.78 dBm/MHz
Range with LOS 2 m
RF centre frequency 7.9 GHz
Raw data rate 1 Mbps
Carrier amplitude 25 mV
RF bandwidth 1 GHz
Subcarrier frequencies 2 – 4 MHz
Table 3.1 – Transmitter characteristics.
22
3.2.2. Subcarrier Modulation
With a triangular waveform modulating the carrier signal we can obtain a transmitted signal with UWB
properties such as a flat spectrum and a very wide bandwidth with steep spectral roll-off.
This subcarrier must also contain the information, so some of its properties must change according to
the modulating data. The change will be only at its frequency level, which means that we are again in
the presence of frequency modulation. We intend to transmit only digital data, and this type of discrete
frequency modulation is called frequency-shift keying. For our specific case, the modulation will be
made to transmit only binary data (BFSK) with a low-modulation index 𝑓 .
To perform this BFSK on the subcarrier signal, it is used the subcarrier generator block present in
figure 3.2 and which schematic is shown in figure 3.3.
Figure 3.3 – Subcarrier generator circuit schematic.
The subcarrier generator block uses two voltage sources to generate each triangular wave with two
different periods 1 and 0 , corresponding to 𝑓1 and 𝑓0 (the
indexes in the period and frequency, correspond to digital high (1) and low (0)). The signals generated
have amplitude of 1 peak-to-peak, with a variation between − and .
A multiplexer is used to select the correct signal in the subcarrier generator output 𝑚(𝑡),
(corresponding to entrance “A” the data “low (0)” and to entrance “B” the data “high (1)”). This
selection is made directly from the digital raw data 𝑑(𝑡).
23
3.2.3. Carrier Modulation
This carrier modulation is implemented with an ideal VCO, as shown in figure 3.2. This circuit
generates a sinusoidal carrier with certain amplitude and the sinusoidal wave instantaneous frequency
is controlled by the VCO transfer function, as described previously, (the VCO transfer function is
presented in figure 2.8).
This block allows the user to select the carrier’s amplitude , centre frequency 𝑓 and the VCO gain
that corresponds to the frequency deviation 𝑓 .
These parameters allows to control the fundamental properties of the transmitted signal, such as
transmitted power and signal bandwidth, which means that it is necessary to realize some simulations
to determine which values are needed to obtain the required specifications, as the data rate and the
power transmission.
3.2.4. Simulation
The simulations were focused in the VCO output signal 𝑋𝐹𝑀(𝑡) that corresponds to the signal that will
be sent to the antenna. It is necessary to remember that the subcarrier signal 𝑚(𝑡) was also important
to these simulations once it directly controlled the frequency instantaneous deviation (𝑡).
The simulation corresponding to the subcarrier generator, where the subcarrier signal 𝑚(𝑡) is
generated, can be observed in figure 3.4.
Figure 3.4 – Simulation results of the digital raw data d(𝑡) and the subcarrier signal 𝑚(𝑡).
24
The most important problem to be solved concern the transmitted power, (UWB spectral power must
respect the maximum limit of -41.3 dBm/MHz). Using the discrete Fourier transform (DFT) special
function, provided by the Cadence® calculator tool, it is possible to calculate the carrier spectral
power.
The equation used to obtain the signal 𝑋𝐹𝑀(𝑡) power spectral density was (3.1):
𝑆𝑋𝐹𝑀(𝑓) 10 [ (𝑋𝐹𝑀(𝑡))
0 10 ] , (3.1)
The value 1 , correspond to an impedance of multiplied by 1 to convert dB units into
dBm units. Since the PSD is measured in dBm/MHz, the time interval values for beginning and stop
are [1 1 1 ] seconds, corresponding to intervals with duration of 1 s. All the other DFT
function parameters, the number of samples considered (32768 samples), the trimming shape window
(rectangular window) and the smoothing factor (equal to 1), are the standard ones used for these
function. Equation (3.1) and DFT parameters in the calculator are shown in figure 3.5.
Figure 3.5 – 𝑋𝐹𝑀(𝑡) power spectral density expression and DFT parameters in the calculator.
The unmodulated carrier signal and its power spectral density are presented in figure 3.6.
25
Figure 3.6 – Unmodulated carrier signal and its respective power spectral density.
The DFT only plots positive frequencies and, as expected, the spectral power density is not
concentrated only in one frequency as shown in figure 2.9. It is possible to observe several harmonics
that are at least 50 dB lower, as well as the phase noise introduced by the simulator. It is clear that the
highest value (-19.059 dBm) corresponds to the carrier frequency (𝑓 ).
The next step consists in applying various triangular waves, (subcarrier signals 𝑚(𝑡)), with different
frequencies 𝑓 at the VCO input, and using two different values for the VCO gain, (corresponding to
different transmission bands ). The simulations were made for three different values of the
amplitude carrier .
The results for the maximum spectral power density can be observed in table 3.2. PSD values are in
dBm.
Ac 15 mV 25 mV 35 mV
BT
fx
500
MHz
1
GHz
500
MHz
1
GHz
500
MHz
1
GHz
2 MHz -44.33 -47.44 -40.89 -44.06 -36.97 -40.05
4 MHz -41.57 -44.50 -37.91 -41.39 -34.12 -37.18
5 MHz -40.39 -43.53 -36.06 -40.06 -33.15 -36.14
10 MHz -37.40 -40.63 -33.01 -35.95 -30.09 -33.12
Table 3.2 – Simulation results for maximum power spectral density for different subcarrier
frequencies, transmission bands and carrier amplitudes.
26
Only the highlighted values in table 3.2 fulfills the maximum spectral power density limit requirement
for UWB EIRP of (-41.3 dBm/MHz), all the others were discarded This approximation takes in account
the number of samples and the time interval (1 s corresponds to 1MHz band).Thus it is guaranteed
that the values selected are in fact lower than those obtained in reality, and a real circuit implemented
with the characteristics corresponding to those values will respect the UWB EIRP limit.
Increasing the subcarrier frequency 𝑓 leads to a higher power spectral density value, (as shown by the
results in table 3.2), however the probability of error in the demodulator decreases. Also, to minimize
the bit error rate in the demodulator, the frequency 𝑓0 should be at least twice the frequency 𝑓1. Based
on table 3.2 results, it is decided to use 𝑓1 for data high “1” and 𝑓0 for data low “0”.
The subcarrier frequency 𝑓 1 was not even considered for simulations.
To obtain a higher signal-to-noise ratio (SNR) in the receiver, it is decided to use 25 mV, (50 mV peak-
to-peak), for the carrier amplitude rather than 15 mV, as shown in figure 3.6, leading to a transmission
band 1 . To allow the transmitter operation in the regions presented in figure 2.3 with a
transmission band 1 , the carrier frequency must be 𝑓 .
The simulations for subcarrier signal 𝑚(𝑡), carrier signal 𝑋𝐹𝑀(𝑡) and carrier power spectral
density 𝑆𝑋𝐹𝑀(𝑓) are represented in figure 3.7, where it is possible to analyze the spectrum of a carrier
with 25 mV amplitude, with a maximum spectral power value of -42.78 dBm/MHz centered in the
carrier frequency 𝑓 .
Figure 3.7 – Simulation of subcarrier signal 𝑚(𝑡), carrier signal 𝑋𝐹𝑀(𝑡) and carrier power spectral
density 𝑆𝑋𝐹𝑀(𝑓)
27
3.3. Transmission Channel Model
The transmission channel model here described pretends to simulate the propagation of an RF signal
in open space and the main nonlinear effects due to attenuation and thermal noise distortion. The
signal is affected by many other effects, such as other types of noise, signal path reflections and
interferences, but these will be ignored at this stage.
It is important to understand how these phenomena affect the signal, so it can be recovered by the
receiver. It is also important to adjust the transmitter specifications, (e.g., transmitted power). In order
to simulate the transmission channel effects a high-level circuit was created using ideal blocks. The
circuit comprises a white Gaussian noise source, an attenuator and an adder. The circuit schematic
can be observed in figure 3.8.
Figure 3.8 – Schematic circuit for the transmission channel model.
3.3.1. Attenuation
Attenuation is one of the major problems concerning to RF transmission. The free-space loss on a line
of sight path is due to spherical dispersion of the radio wave. The signal loss or attenuation is given
by (3.2) [13]:
(
)2
(
)2
, (3.2)
28
As we can see the loss depends directly on the signal wavelength and the distance 𝑑, since we are
considering propagation in free-space, the radio wave propagation occurs at the speed of light
( 1 ) and therefore the equation depends only on the signal frequency 𝑓 and distance 𝑑.
It is more usual to express signal attenuation in decibels, and in this case the expression is (3.3):
10 (
) , (3.3)
The attenuator block in figure 3.8 was developed in a standard modeling language for analog circuits
named Verilog-A and its code is presented in Annex, figure A1.
With (3.3), some calculations were made for different distances assuming the carrier frequency and,
as expected, the attenuation value increases approximately 6 dB when the path length doubles. The
results are listed in table 3.3.
f d 0.5 m 1 m 2m
7.9 GHz 44.37 dB 50.39 dB 56.41 dB
Table 3.3 – Attenuation values for different distances using the carrier frequency of 7.9 GHz.
3.3.2. Thermal Noise
The term noise is usually used to refer phenomena that tend to disturb the transmission and the
processing of signals in communication systems [14]. A noise signal instantaneous value and phase
cannot be predicted at any time. Since noise sources have amplitudes that vary randomly with time,
they can only be specified by a probability density function [15]. If we ignore undesirable signal
interferences from other electromagnetic sources, the type of noise that mostly affects the signal in its
transmission is thermal noise. It is in thermal noise that we will centre our attention.
Thermal noise is also known as Johnson noise (this type of noise was first measured by John B.
Johnson at Bell Labs in 1928) and is normally characterized as being additive white Gaussian noise
(AWGN), it is generated by thermal agitation of the atomic particles in a conductor. When the heat
increases in a conductor the thermal noise also increases. Heat disrupts the electrons’ response to an
applied potential, it adds a random component to their motion. Thermal noise only stops at absolute
zero [15].
29
The thermal noise is denominated as white due to its flat or uniform spectral power density, which
means that it is independent of the operating frequency. The adjective white is used as comparison to
the white light that contains equal amounts of all frequencies within the visible band of electromagnetic
radiation [14].
Thermal noise has a Gaussian PDF, meaning that there is a mean value of amplitude, which is most
likely to occur. The probability that a noise amplitude will be higher or lower than the mean falls off in a
bell shaped curve, which is symmetrical around the centre [15]. This concept is illustrated in figure 3.9.
Figure 3.9 – Noise signal and Gaussian probability density function.
From quantum mechanics considerations, the noise power spectral density 𝑁0 (in watts per hertz)
generated in any lossy element is given by (3.4) [7]:
𝑁0 𝑓 [1
𝑒 ⁄ 1 1], (3.4)
where ( 1 ) is the Planck constant, (1 1 2 ) is Boltzmann
constant, 𝑓 is the frequency in hertz and is the absolute temperature in kelvin.
30
For frequency values of Gigahertz order or smaller, the noise power density equation can be reduced
to (3.5) [7]:
𝑁0 [W/Hz], (3.5)
The noise power 𝑁 in a certain bandwidth is given by (3.6):
𝑁 1 10( .1000) , (3.6)
Noise power 𝑁 is noise density 𝑁0 multiplied by the bandwidth under consideration. The wider the
bandwidth, the more the noise power collected.
In Table 4 are presented some values for different bandwidths with .
B [GHz] N [dBm]
0.25 -90
0.5 -87
1 -84
2 -81
Table 3.4 – Noise power values for different bandwidths.
The Gaussian probability density function for thermal noise has mean and variance 2 𝑁0
[13]. Considering 1 the standard deviation is .
The white Gaussian noise source in figure 3.8, was also developed in Verilog-A, its code is presented
in Annex, figure A2.
The noise source was adjusted to have a noise power spectral density of 𝑁0 −11 , and
in this condition the maximum amplitude value detected in this sample was around , as we can
observe in figure 3.10.
31
Figure 3.10 – Simulation of noise signal 𝑁(𝑡) and respective power spectral density 𝑁0.
3.3.3. Simulation and Signal-to-Noise Ratio
The signal-to-noise ratio (SNR) can be defined as the ratio between the signal power and the noise
power, usually referred to the receiver input, and is given by (3.7):
(
), (3.7)
It is usually expressed in decibels ( ) and 𝑆 represents the signal power and 𝑁 the noise power,
both detected at the receiver input. This ratio is very useful because it provides an indication of the
degree to which the signal has been contaminated with additive noise [13].
Due to the attenuation effect, the signal power detected at the receiver input, considering a 2m path
length and a transmission band of 1 is (3.8):
𝑆 1 10 (1 1
) − 1 , (3.8)
As we saw before, for a 1 of detection band, we have a noise power 𝑁 of − and
therefore is (3.9).
𝑆 − 𝑁 − 1 − (− ) 1 1 , (3.9)
32
When realizing experimental work it is not possible to determine the signal power 𝑆 alone, because we
cannot turn off the noise, instead we must assume superposition of signal and noise power
(𝑆 𝑁), and in this case we can determine the SNR in the following way (3.10) [13]:
(
)
( )
(
) 1 1, (3.10)
In figure 3.11 we can observe the signal attenuated 𝑋′𝐹𝑀 (𝑡), the noise signal 𝑁 (𝑡) and the
signal 𝑌𝐹𝑀 (𝑡) present at the receiver input and given by (3.11):
𝑌𝐹𝑀 (𝑡) 𝑋′𝐹𝑀 (𝑡) 𝑁 (𝑡), (3.11)
Figure 3.11 – Simulation of signal attenuated 𝑋′𝐹𝑀 (𝑡), noise signal 𝑁 (𝑡) and signal 𝑌𝐹𝑀 (𝑡).
3.4. Receiver
The receiver described next comprises two major circuits, a preamplifier and a demodulator. This high
level implementation also uses ideal components. The receiver will allow the test of the signal
reception after noise and distortion and to test the concepts involved in its demodulation.
33
One of the major characteristics in a FM-UWB system is that the RF signal is not pulsed hence no
synchronization is needed, allowing non coherent detection, and therefore a simple hardware
implementation can be realized leading to a possible low power consumption [16]. The major
problems that this receiver has to face are the detection of a very low power signal and the design of
filters with a high quality factor , of extremely difficult implementation.
Physical FM demodulators based on direct frequency demodulation simply do not exist, because there
is no physical system capable of reading the instantaneous frequency of a carrier. It is necessary to
convert the signal into a PM or AM signal, so it can be demodulated [16].
This demodulator circuit includes four different stages:
First, the signal is applied to a 4th order band-pass filter, (the fundamental block to this type of
demodulation), converting the signal from FM to AM.
Second, the AM signal will be rectified and amplified by a multiplication by itself.
Third, a low-pass filter to detect the AM envelope is used.
Forth, by using a comparator the envelope signal will be converted into binary digital data, so
it could be decoded and transformed into the original data.
The preamplifier schematic circuit is presented in figure 3.12 and the demodulator schematic circuit is
presented in figure 3.13.
Figure 3.12 – Preamplifier high level schematic.
Figure 3.13 – Demodulator high level schematic.
34
3.4.1. Preamplifier
The preamplifier circuit consists in a band-pass filter and an amplifier as shown in figure 3.12. The
band-pass filter will limit the noise distortion to the signal transmission band and the amplifier will
amplify the signal, adding the minimum possible noise, to a range possible to be used by the
demodulator.
The total gain of the receiver is around 44 dB. The band-pass filter is based in a Laplace 2nd
order
transfer function (3.12) and was implemented in Verilog-A, its code is presented in Annex, figure A3.
( )
, (3.12)
where represents the filter gain, 0 is the filter centre frequency in radians per second and 0 is the
filter quality factor, the filter passing band is given by (3.13):
, (3.13)
The filter implemented has the following values, , 0 , 1
and 0 .
In figure 3.14 it is possible to observe the signal at the input 𝑌𝐹𝑀 (𝑡) and output 𝑋′′𝐹𝑀 (𝑡) of the
preamplifier circuit.
Figure 3.14 – Signal at the input 𝑌𝐹𝑀 (𝑡) and output 𝑋′′𝐹𝑀 (𝑡) of the preamplifier circuit.
35
3.4.2. Spectral Shaping Band-Pass Filter
The filter used in this demodulator is normally called spectral shaping filter and converts an FM signal
into an AM signal. The filter passing band is approximately ten times smaller than the signal
transmission band, (narrow band response), and it has a high gain. The filter is centered in the carrier
fundamental frequency, meaning that every time the carrier instantaneous frequency is approximately
equal to the carrier fundamental frequency the signal will be amplified, otherwise it will be attenuated,
acting this way like a sensor.
In this particular case it is possible to demodulate the data just by knowing the subcarrier frequency
without obtaining the original message (triangular wave). It is a conversion from FM signal into
Gaussian shape AM signal where the filter response directly controls the envelope of the demodulated
signal [16]. In figure 3.15 the conceptual idea that allows the conversion of FM into AM signal is
illustrated.
Figure 3.15 – Band-pass filter response to convert FM into AM signal.
The filter characteristics are the following, , 0 , 1
and 0 . As mentioned, the filter high quality factor is one of the challenging problems concerning
this implementation as well as the filter tunability. It is important to tune the filter to correspond to
different FM signal bandwidths and to mitigate some process variation [16].
To minimize the filter high quality factor, a 4th order band-pass filter was implemented, by using two
band-pass biquadratic sections. The characteristics for the first section are the following, 1 1 ,
36
0 , 1 1 1 and 0 , and for the second are, 2 1 ,
0 , 2 1 1 and 0 . Both sections were based in a Laplace 2nd
order transfer function (3.12) and were implemented in Verilog-A. The code is similar to the one used
for the preamplifier filter presented in Annex, figure A3.
In figure 3.16 we can observe the subcarrier signal 𝑚(𝑡) and the AM signal resulting from the filter
conversion 𝑋𝐴𝑀 (𝑡). It is important to note that the resulting AM pulses frequency is twice the
respective subcarrier frequency, since for each subcarrier period the carrier frequency will assume two
times the fundamental frequency 𝑓 , as represented in figure 3.15.
Figure 3.16 – Subcarrier signal 𝑚(𝑡) and AM signal 𝑋𝐴𝑀 (𝑡) resulting from the filter conversion.
37
3.4.3. Signal Rectifier
The signal rectifier block allows a signal improvement and elimination of the signal negative
component, by multiplying the signal for itself. The result is shown in figure 3.17.
Figure 3.17 – Signal after rectification 𝑋𝐴𝑀 𝟐(𝑡).
3.4.4. Low-Pass Filter
The low-pass filter acts as a signal envelope detector removing this way the carrier signal. The low-
pass filter is also based in a Laplace 2nd
order transfer function (3.14) and it was implemented in
Verilog-A. The code used is the same shown in Annex figure A3, but with the following low-pass filter
Laplace function.
( )
, (3.14)
The filter characteristics are, , 0 𝑑 and 0 1. The AM envelope signal
𝑌𝐴𝑀 (𝑡) detected by the low-pass filter is presented in figure 3.18.
38
Figure 3.18 – Low-pass filter output signal 𝑌𝐴𝑀 (𝑡).
3.4.5. Comparator
The comparator used here is a basic converter that compares the signal pulses amplitude with a
certain reference value and converts the signal into a digital binary code. The reference value used
is 𝑉 𝑒 , if the signal amplitude is above this value the resultant signal will be a constant
signal with amplitude 𝑉 𝐺 1 , else 𝑉 . The comparator input signal 𝑌𝐴𝑀 (𝑡) and output
signal 𝑌′𝐴𝑀 (𝑡) can be observed in figure 3.19.
Figure 3.19 – Comparator input signal 𝑌𝐴𝑀 (𝑡) (red signal) and output signal 𝑌′𝐴𝑀 (𝑡) (blue signal).
39
3.5. Digital Decoder
The demodulator previously presented, generate a binary digital code (BDC) directly related with the
subcarrier frequency, by knowing the subcarrier frequency it is possible to obtain the original data. In
figure 3.20 the digital decoder circuit schematic is shown.
Figure 3.20 – Digital decoder circuit schematic.
First is used a counter with a clock frequency eight times higher than the higher subcarrier frequency.
The pulses generated are used as a counter reset. That way, a value that measures the respective
delay between each pulse is generated. This value is stored in a register and then it enters into a logic
circuit. If the value is between 0 and 4 the logic circuit output will be logic 0 otherwise will be logic 1.
The value generated by the logic circuit is stored in a chain of two flip-flop D not in phase with each
other, the first one with a clock frequency four times smaller than the original data rate and the second
one with a clock frequency equivalent to the original data rate. The flip-flop D chain guarantees that
the demodulated signal has the same frequency than the original one with a lower bit error rate.
In figure 3.21 it is possible to observe the original data 𝑑(𝑡), the BDC signal 𝑌′𝐴𝑀 (𝑡) and the
recovered data 𝑑′(𝑡). The major delay between the original data and the recovered data is due to the
flip-flop D chain.
Figure 3.21 – Original data 𝑑(𝑡), BDC signal 𝑌′𝐴𝑀 (𝑡) and recovered data 𝑑′(𝑡).
40
3.6. Conclusions
In this chapter the transmitter, the transmission channel and the receiver high-level model circuits
were studied. The circuits functioning was described and simulated to validate the theoretical concepts
involved in this type of modulation.
It was shown it is possible to create a carrier with flat spectrum and a very wide bandwidth with step
spectral roll-off by using a triangular subcarrier signal. Despite the low amplitude of the carrier, needed
to respect the EIRP limits, it was shown that is possible to recover the data after the main distortions
originated by the transmission channel. Also, it was proved that the proposed receiver does not need
a PLL circuit to recover the data, however, the band-pass filter needed to transform the FM signal into
AM signal could be the major concern in its implementation.
41
4. Circuit Implementation ____________________________________________________
42
4.1. Introduction
In the last chapter an explanation about the desired transmitter behavior and its functioning was made.
This chapter pretends to evolve that explanation into a description of each block composing the circuit,
with the respective functioning simulations.
To implement the transmitter, two relaxation oscillators were used, one to implement the subcarrier
generator and the other to implement the voltage controlled oscillator. In both blocks, different
characteristics of this type of oscillator were exploited. The main advantages of using this type of
oscillator are its simple operation mode and its simple architecture that could lead to reduced area and
power consumption.
4.2. Relaxation Oscillators
It is possible to separate the existing oscillators in two main types: quasi-linear oscillators and strongly
non-linear [17].
Quasi-linear oscillators include the LC oscillators, the most common in this category. LC oscillators
can use dielectric resonators, crystals, striplines and LC tanks as a resonator element and are known
by their good phase-noise performance.
Strongly non-linear (or relaxation) oscillators are usually implemented with RC-active circuits [18].
Relaxation oscillators are RC type and tend to have a higher phase-noise when comparing to LC
oscillators. However, by using only resistors and capacitors as passive devices, instead of inductors,
relaxation oscillators can achieve lower area and cost, being widely used in fully integrated circuits
where phase-noise requirements are not demanding [18], [19].
The relaxation oscillator is referred to as a first order oscillator, since its behaviour can be described in
terms of first order transients [20]. The simple operation mode is based on the alternated charging and
discharging of a capacitor between two threshold voltage levels that are pre-defined, and its oscillation
frequency is inversely proportional to its capacitance [21].
43
4.2.1. High-Level Model
To better understand the relaxation (or RC) oscillator operation mode, it is first described using a high-
level model, comprising an integrator and a Schmitt-trigger, as shown in figure 4.1.
Figure 4.1 – Relaxation oscillator block diagram.
The Schmitt-trigger controls the sign of the integration constant by imposing a threshold value to the
integrator, the sign is changed when that value is reached. The output waveforms of the two blocks
are represented in figure 4.2.
Figure 4.2 – Oscillator output waveforms VINT and VST.
As we can observe in the previous figure, when the Schmitt-trigger output (VST) is at its highest value,
the integrator output (VINT) increases. When VINT reaches the imposed maximum threshold value, the
Schmitt-trigger output will change to its lowest value and the integrator output starts to decrease until it
44
reaches the minimum threshold value. Again, the Schmitt-trigger output commutates to its highest,
and the process starts from the beginning.
4.2.2. Circuit Implementation
The oscillator implementation should be as simple as possible for operation at very high frequencies,
reducing this way the number of potential noise sources. A possible relaxation circuit implementation
is presented in figure 4.3.
Figure 4.3 – Relaxation oscillator implementation.
The integrator is simply implemented by a capacitor, where the capacitor current ( ) represents the
block input, and the capacitor voltage (𝑣 ) the respective block output. In figure 4.4 is represented the
integrator block and respective waveform.
Figure 4.4 – Integrator block and respective output waveform.
45
The Schmitt-trigger is implemented by two transistors NMOS, two resistances and two current
sources, being 𝑣 its input and its output. In figure 4.5 is represented the Schmitt-trigger and its
respective transfer function, assuming that the switching occurs abruptly when the sign of 𝑣 − 𝑣
changes.
Figure 4.5 – Schmitt-trigger block and respective transfer function.
Usually, the differential oscillator output is the voltage 𝑣 𝑣1 − 𝑣2 [20]. This value corresponds
to the threshold limits imposed by the Schmitt-trigger and it is described by the following equations
(4.1):
𝑣 {𝑉 − (𝑉 − ) 1 − (𝑉 − ) − 𝑉 − 2 −
, (4.1)
In this condition, the maximum amplitude of the output voltage is , as represented in figure 4.6.
Since the oscillator integration constant is ( 𝐶), for low oscillation frequencies, the oscillation
frequency can be calculated by equation (4.2).
𝑓0 1
2𝐶( )
1
𝐶, (4.2)
However, this equation can also be used to estimate the order of the components used for circuit
implementation, even for higher frequencies. A more accurate expression for the oscillation frequency
can be found in [22].
46
Figure 4.6 – Relaxation oscillator waveforms.
Despite its relaxation behaviour, corresponding to lower oscillator frequencies, this type of oscillator
can also operate with an almost sinusoidal behaviour, being this one of the characteristics that will be
explored for the VCO implementation. At very high frequencies the outputs are approximately
sinusoidal, since the harmonics are filtered out by the circuit parasitics.
4.3. Frequency-Shift Keying Implementation
It is usual to use a direct digital synthesizer (DDS) for this type of subcarrier generation [9], but due to
its complexity, it was chosen to use an oscillator. It is necessary to generate a triangular wave with two
different frequencies of operation and to do so, the characteristic that will be explored in the oscillator
is the integrator output waveform VINT rather than the usual Schmitt-trigger output waveform VOUT.
As seen before, in figure 4.4, the integrator is only a capacitor and its integration constant is the
current that flows through the capacitor over its capacitance ( 𝐶), meaning that changing the
current or the capacitance 𝐶, will linearly change the capacitor time of charging and discharging and
therefore the frequency of the output waveform VINT, as shown in figure 4.7.
The solution adopted was to change the oscillator capacitance value and to perform that, the oscillator
was implemented with two capacitors of about the same value, where one of them could be used or
not, by turning on or off a pair of transistors. This circuit is illustrated in figure 4.8.
47
Figure 4.7 – Integrator waveform 𝑣𝐶0 (corresponding to a capacitance C) and 𝑣𝐶1 (corresponding to
a capacitance 2C).
Figure 4.8 – Relaxation oscillator with variable capacitance.
This implementation leads to a differential output where the pretended waveform is 𝑣𝐶 𝑣 − 𝑣 .
When the signal DATA is equal to “0” the transistors M3 and M4 will be cut-off and 𝑣𝐶 will be equal
to 𝑣𝐶 , otherwise when DATA is equal to “1” the transistors M3 and M4 will be conducting and the
capacitance value will lead to 𝑣𝐶 equal to 𝑣𝐶1.
48
4.3.1. Control Circuit
To guarantee that the capacitance value only changes when the capacitor completely charges or
discharges, it is necessary to develop an additional control circuit. By performing this operation the
error introduced in the modulated signal is lower (confirmed by simulation results). However, it is
impossible to eliminate this error completely since it results from the impossibility to generate signals
with an infinite frequency precision, due to the discrete capacitance values allowed. The control circuit
schematic is shown in figure 4.9
Figure 4.9 – Control circuit schematic.
The control circuit is simply composed by a buffer and a flip-flop D edge-triggered. The buffer is used
to transform the signal 𝑣1 or 𝑣2 into a digital signal, that will be used as clock to control the flip-flop D.
This way, the flip-flop D only changes its value when 𝑣1 or 𝑣2 changes, synchronizing the signal
DATA with the integrator transitions. The signal 𝑑(𝑡) represents the data to be transmitted.
4.3.2. Simulation Results
In order to verify the behavior of the redesigned relaxation oscillator along with the use of a control
circuit (circuit represented in figure 4.10,), it is simulated using components from the UMC 0.13 µm
design kit, such as the RF transistors N_12_RF. Also, it was used two transistor implemented current
sources adjusted to the needed specifications. Instead of using capacitors, it was chosen to use RF
varactors, specifically designed to operate in RF frequencies and to allow a possible capacitance
adjustment. The elements parameters used in the simulations are indicated in table 4.1.
49
Figure 4.10 – Relaxation oscillator with variable capacitance and control circuit.
Width [m] Length [m] Gate Finger Multiplier Value
M1 2.5µ 120n 16 1 -
M2 2.5µ 120n 16 1 -
M3 2.5µ 120n 16 1 -
M4 2.5µ 120n 16 1 -
C1 10µ | 10µ 520n | 530n 8 | 8 5 | 5 2.45p + 2.50p [F]
C2 10µ | 10µ 520n | 530n 8 | 8 6 | 6 2.94p + 3.00p [F]
R 510n 18µ - 6 28.544 k [Ω]
Iav - - - - 15.81µ [A]
Table 4.1 – Parameters used in the simulation of the relaxation oscillator with variable capacitance.
The variable capacitances C1 and C2 were implemented with a pair of varactors assembled in anti-
parallel. In these conditions the frequencies obtained were of 4.361 MHz for 𝑣𝐶 and 2.070 MHz
for 𝑣𝐶1. The maximum voltage value for the differential output is 575 mV, and the minimum voltage is
225 mV.
50
The graphical result for the signal 𝑣𝐶 𝑣 − 𝑣 is depicted in figure 4.11, for a data signal 𝑑(𝑡) with
transitions between “0” and “1”. It is also possible to observe the graphical results for the control circuit
in figure 4.12 and figure 4.13.
Figure 4.11 – Simulation result for the signal 𝑣𝐶, for a data signal 𝑑(𝑡) with transitions between “0”
and “1”.
Figure 4.12 – Simulation result for the signal 𝑣1 and CLK, for a data signal 𝑑(𝑡) with transitions
between “0” and “1”.
51
Figure 4.13 – Simulation result for the signal DATA and comparison with the original data signal 𝑑(𝑡)
with transitions between “0” and “1”.
The results obtained in the circuit simulation are very satisfactory, proving that is possible to obtain a
triangular waveform and perform FSK with the modified relaxation oscillator and the help of a simple
control circuit. The modified relaxation oscillator along with the control circuit, from now on, will be
referred in this work as FSK oscillator.
4.4. Voltage Controlled Oscillator Implementation
The voltage controlled oscillator can be considered the key element in a frequency modulation
process (remember equation (2.5) and figure 2.8), and perhaps one of the most challenging blocks to
design in a high-performance system. It typically represents the bottleneck of the achievable noise
performance, making LC oscillators the primary choice in a VCO design [23]. However and as already
referred, this performance comes at great area expense, mainly due to the difficulty of realizing good
quality integrated spiral inductors, meaning a great cost when working with IC [23].
The lower area occupied by a relaxation oscillator, (when comparing to a LC oscillator), and its simple
mode of operation makes it a potential alternative in RF VCO design. For these reasons, the
relaxation oscillator is chosen to the VCO implementation instead of an LC oscillator.
The VCO can be simply implemented using an oscillator whose capacitance variation could be
controlled by a reference voltage in a linear way. Also, it is necessary to use a sinusoidal carrier for
52
transmission. Considering the oscillator output the signals 𝑣𝑜𝑢𝑡1 or 𝑣𝑜𝑢𝑡2, and knowing that for higher
oscillation frequencies these signals present an almost sinusoidal behavior, the major problem
concerns the wide variation needed to achieve at least 1 GHz bandwidth.
Commonly, varactors are used, being implemented as reverse-biased p-n junctions. In CMOS
technology, this can be accomplished using the available p+/n
− diffusions and p
-/n
- wells. Despite
having a modest maximum-to-minimum capacitance ratio 𝐶 𝑎𝑡𝑖𝑜 (4.3) that worsens as the supply
voltage scales, p-n junction varactors are adequate for applications with limited tuning needs [23],
which is not the case.
𝐶 𝑎𝑡𝑖𝑜 𝐶 𝑎 𝐶 𝑖𝑛⁄ , (4.3)
A possible solution, and perhaps the most usual, is to use an array of p-n junction varactors providing
a linear and continuous frequency tuning range within different bands. However, this solution implies a
control circuit with some complexity.
The solution adopted in this work, is to use the parasitic capacitances gate-source 𝐶 and gate-
drain 𝐶 of a MOS transistor as variable capacitances. The details of this solution are presented in
section 4.4.1. Combining some properties of the MOS transistors that affect its parasitic capacitance
and rearranging the basic relaxation oscillator topology, it is also possible to use the differential output
produced by the FSK oscillator, otherwise it would be necessary to use a differential-to-single
converter, introducing more complexity to the circuit and possible signal distortion. The proposed
topology for the VCO is presented in figure 4.14.
Figure 4.14 – Proposed topology for VCO with MOS varactors and differential tuning signals.
53
4.4.1. MOS Varactors
MOS varactors are variable, voltage-controlled capacitors based on the MOS structure. Their main
application is voltage-controlled oscillators [24]. The varactor capacitance 𝐶𝐺 (4.4) determines the
VCO frequency 𝑓0 (4.2).Their main advantage is an intrinsic capacitance ratio (𝐶 𝑎𝑡𝑖𝑜) that is much
higher than that of p-n junction varactors. For the small-signal model, 𝐶 𝑎𝑡𝑖𝑜 values of 2 to 5 can be
achieved in practice, even with control voltage swings as small as 1V [23].
𝐶𝐺 𝐶 𝐶 , (4.4)
In figure 4.15 is represented a cross section of an NMOS varactor and the small-signal model
generally assumed for varactors: a variable capacitance in series with a variable resistance (RV).
Figure 4.15 – Cross section of a conventional NMOS varactor (in depletion; left) and the generally
assumed model (right). The dashed line indicates the border of the depletion region [24].
The MOS varactor is not a four-terminal device as the transistor but a three-terminal device. The
source and drain regions are shorted to apply the voltage Vtune that tunes the variable capacitance.
The p- body is grounded and the voltage Vgate is applied to the gate node. The variable capacitance 𝐶𝐺
appears between the gate node and all the others. It is also possible to define 𝐶𝐺 as the series
connection of the gate oxide capacitance 𝐶𝑜 and the variable depletion region capacitance 𝐶 , as
shown in equation (4.5) [24].
1
𝐶
1
𝐶
1
𝐶 , (4.5)
Depending on the voltage applied at the gate terminal Vgate, it is possible to define different operation
regions for the MOS varactor, known as accumulation-, depletion- and inversion-mode. In figure 4.16
is possible to observe the variation of capacitance 𝐶𝐺 for small-signal model within these different
operation modes [24].
54
Figure 4.16 – Typical measured small-signal capacitance characteristic 𝐶𝐺 of a NMOS varactor
(bottom) and the relevant lumped elements (top) at zero tuning voltage [24].
Inversion- and accumulation-mode are the most common varactor configurations. Because electrons
are the majority carriers in the depletion and accumulation regions, the accumulation-mode device has
less parasitic resistance than the inversion-mode device, which uses holes as majority carriers [24].
The transition from depletion to inversion is determined by the voltage difference between gate and
source/drain and the threshold voltage. Therefore the transition voltage will be increased with
increasing tuning voltage, meaning that is possible to shift the inversion-mode characteristic granting
another degree of freedom to the design. In figure 4.17 is possible to observe the typical measured
small-signal capacitance characteristic of a conventional NMOS varactor at various tuning voltages
[24].
Figure 4.17 – Typical measured small-signal capacitance characteristic of a conventional NMOS varactor at various tuning voltages [24].
55
For the NMOS device, the source and drain are n+
doped. The substrate (or well) region between and
around source and drain is of opposite doping, i.e. p- type. Process determined the polysilicon gate is
of the same doping as source and drain, i.e. n+
type. A PMOS device is obtained when all regions
have opposite doping as in the NMOS [24]. Hence, the inversion-mode capacitance characteristic 𝐶𝐺
of a PMOS varactor is the symmetric of a NMOS varactor. In figure 4.18 are represented both NMOS
and PMOS inversion-mode capacitance characteristics 𝐶𝐺.
Figure 4.18 – PMOS and NMOS varactor inversion-mode capacitance characteristic 𝐶𝐺.
Due to these characteristics, since they are symmetric, using a NMOS varactor as variable
capacitance C1 and a PMOS varactor as variable capacitance C2, (remember figure 4.14), allow the
possibility to use the differential output produced by the FSK oscillator, (without using a differential-to-
single stage), generating a carrier with the desired frequency variation and avoiding this way distortion
in the subcarrier signal.
4.4.2. Simulation Results
To verify the behavior of the VCO using MOS varactors and controlled by a differential voltage input,
the circuit presented in figure 4.14 was simulated using components from the UMC 0.13 µm design kit.
Also, the two current sources used in the present simulations were designed with real components
and adjusted to the needed specifications. Once more, instead of using a capacitor to implement the
VCO capacitance C0, it was used a RF varactor for the same reasons presented in section 4.3.2,
dedicated to the FSK oscillator simulations. It was necessary to use a pair of varactors for both
capacitances (C1 and C2), not only to achieve the desirable capacitance variation values, but also to
create a common input control terminal, as shown in figure 4.19. The elements parameters used in the
simulations are indicated in table 4.2.
56
Figure 4.19 – Three terminal MOS varactor circuit schematic.
Width [m] Length [m] Gate Finger Multiplier Value
M1 2.6µ 120n 16 1 -
M2 2.6µ 120n 16 1 -
C0 2.5µ 150n 3 6 79.65f [F]
C1 M1C1 900n 270n 15 6 Cmax Cmin
M2C1 900n 270n 15 6 101.69f [F] 32.58f [F]
C2 M1C2 1.6µ 200n 15 6 Cmax Cmin
M2C2 1.6µ 200n 15 6 105.47f [F] 36.03f [F]
R 2µ 7.44µ - 3 1.36 k [Ω]
Iav - - - - 2.48m [A]
Table 4.2 – Parameters used in the VCO simulation.
The maximum-to-minimum capacitance ratio obtained for the variable capacitance C1, corresponding
to the varactor NMOS, was of =3.12. For the variable capacitance C2, corresponding to the
varactor PMOS, the value obtained was of =2.93. These values were obtained for a variation of
400 mV. It was used 200 mV as the minimum voltage (Vmin) and 600 mV as the maximum voltage
(Vmax), due to the minimum and maximum values expected for the FSK oscillator outputs.
By applying Vmin to one varactor and Vmax to the other varactor, it was possible to obtain the maximum
and minimum VCO oscillation frequencies within the referred interval. The maximum and minimum
frequencies obtain were 6.78 GHz and 8.56 GHz.
57
To verify the VCO frequency variation behaviour, it was applied a constant voltage value of 400 mV to
one varactor and to the other a linear variation from 200 mV to 400 mV during 500 ns. The results are
presented in figure 4.20.
Figure 4.20 – VCO frequency variation resulting from NMOS varactor excitation (1) and PMOS
varactor excitation (2).
As expected, it is not possible to obtain an accurate linear variation for the frequency, due to the many
constraints involved, such as the components dimensions allowed by technology, the simplification of
the models involved and the ambitious bandwidth considered. Also, it is possible to observe that the
variation obtained with the PMOS varactor excitation has a more smooth variation than that obtained
with the NMOS varactor excitation.
The VCO output 𝑣𝑜𝑢𝑡1 presented in figure 4.21, was obtained with a voltage value of 400 mV applied
to both varactors, and its frequency is of about 7.87 GHz.
Figure 4.21 – VCO output 𝑣𝑜𝑢𝑡1 with a frequency of 7.87 GHz.
58
4.5. Current Sources
The current sources used in the FSK oscillator and in the VCO share the same basic topology and
both have a terminal that allows the circuit power down. The difference between them is the different
current values obtained throw different transistors dimensions. The circuit schematic for both current
sources is depicted in figure 4.22.
Figure 4.22 – Current source circuit schematic.
The elements used in FSK current source and VCO current source are from the UMC 0.13 µm design
kit, and their parameters are presented in table 4.3 and table 4.4, respectively.
Width [m] Length [m] Gate Finger Multiplier Value
M1 1.1µ 120n 1 1 -
M2 1.1µ 120n 1 1 -
M3 1µ 120n 1 1 -
M4 1µ 360n 1 1 -
M5 1.6µ 120n 1 1 -
C - - - - 10p [F]
Iav - - - - 15.81µ [A]
Table 4.3 – Parameters used in the FSK current source.
59
Width [m] Length [m] Gate Finger Multiplier Value
M1 75µ 120n 1 1 -
M2 65µ 120n 1 1 -
M3 18µ 120n 1 1 -
M4 1µ 360n 1 1 -
M5 100µ 120n 1 1 -
C - - - - 10p [F]
Iav - - - - 2.48m [A]
Table 4.4 – Parameters used in the VCO current source.
4.6. Conclusions
In this chapter it was performed a study about the relaxation oscillator and the possibility of using this
type of oscillator to build the main cores of the proposed transmitter, the subcarrier generator
(frequency-shift keying oscillator) and the voltage controlled oscillator.
Regarding the FSK oscillator, the implementation of a second capacitance allows the possibility of
obtaining a signal with two different oscillation frequencies, controlled by the data to be transmitted
along with the help of a control circuit, of simple implementation.
Despite the differential output produced by the FSK oscillator, and as referred before, there is no need
of using a differential-to-single converter, since the VCO oscillator is redesigned to use a differential
input to control the two variable capacitances.
As expected, the VCO oscillator output for high frequencies, presents a sinusoidal behaviour and the
use of MOS varactors as variable capacitances, allows a frequency variation of 1.78 GHz.
The simulation results obtained for both circuits are considered very satisfactory, once they fulfill the
proposed specifications needed to generate a FM signal with UWB properties.
60
61
5. Transmitter Simulation
and Results ____________________________________________________
62
5.1. Introduction
In this chapter simulations of the FM-UWB transmitter are presented and analyzed. Both oscillators
presented earlier are connected to produce the desired FM-UWB signal. This chapter is divided in two
main sections: in section 5.2 the transmitter schematic is presented, along with a table containing the
final components parameters used in the transmitter implementation. Also, the circuit core simulation
results are presented, comprising already the output buffer, bonding wires, pads and ESD protections.
Finally, in section 5.3 the transmitter layout is presented.
5.2. Transmitter Schematic and Simulations
5.2.1. Circuit Schematic
The transmitter schematic is presented in figure 5.1. In order to simplify the drawn schematic the
current sources are only represented by its symbols, since their schematics are already shown in the
previous chapter, figure 4.22, and their elements parameters remained without any modification
(tables 4.3 and 4.4).
Figure 5.1 – Transmitter circuit schematic.
63
In table 5.1 the final elements parameters used in the transmitter are shown. All the elements used to
implement the transmitter are RF elements from the UMC 0.13 µm design kit. The transistor models
used to implement the NMOS and PMOS type were the N_12_RF and P_12_RF.
Width [m] Length [m] Gate Finger Multiplier Value
M1 – M4 2.5µ 120n 16 1 -
M5 – M6 2.6µ 120n 16 1 -
C0 2.7µ 150n 5 2 47.79f [F]
C1 10µ | 10µ 500n | 500n 8 | 8 4 | 5 1.89p + 2.36p [F]
C2 10µ | 10µ 500n | 500n 8 | 8 6 | 5 2.83p + 2.36p [F]
C3 M1C3 900n 270n 15 6 Cmax Cmin
M2C3 900n 270n 15 6 101.69f [F] 32.58f [F]
C4 M1C4 1.6µ 200n 15 6 Cmax Cmin
M2C4 1.6µ 200n 15 6 105.47f [F] 36.03f [F]
R1 – R2 510n 18µ - 6 28.54k [Ω]
R3 – R4 2µ 6.88µ - 3 1.26k [Ω]
I1av - - - - 16.17µ [A]
I2av - - - - 2.63m [A]
Table 5.1 – Parameters used in the transmitter implementation and simulation.
The variable capacitances C1 and C2 were implemented with a pair of varactors assembled in anti-
parallel. The variable capacitances C3 and C4 represent the MOS varactors implemented with two
pairs of MOS transistors. Transistors M1C3 and M2C3 correspond to capacitance C3 and transistors M1C4
and M2C4 to capacitance C4, as indicated in the previous table.
When comparing the element parameters from table 5.1, with those presented in tables 4.1 and 4.2
used to test the FSK oscillator and the VCO oscillator independently, it is possible to note that it was
necessary to perform small adjustments to maintain the desired characteristics.
64
5.2.2. Graphical Results
All the simulation results presented here were obtained considering a simulation interval of 8.5 µs. In
figure 5.2 is depicted the original data signal 𝑑(𝑡) and the signal DATA, resulting from the control
circuit, that is synchronized with the FSK oscillator transitions.
Figure 5.2 – Signal 𝑑(𝑡) with transitions between “0” and “1” and simulation result for signal DATA.
It is possible to observe that the signal DATA impulses have different duration. These different
duration impulses result from the constant synchronization between the output produced by the FSK
oscillator and the original data 𝑑(𝑡), this happens due to the impossibility of creating an oscillator with
an infinite precision frequency, as already referred.
The graphical comparison between these two signals is already established in figure 4.13, but it is
important to analyze it again, since the signal DATA is the primary signal that controls the entire
transmitter behaviour and it is necessary to establish comparisons between this signal and some other
results presented ahead.
The output 𝑉𝑡𝑢𝑛𝑒1 and 𝑉𝑡𝑢𝑛𝑒2 produced by the FSK oscillator, as well as the differential signal 𝑣𝐶
(resulting from the difference between them), are illustrated in figure 5.3 and figure 5.4.
65
Figure 5.3 – Simulation result for the differential output 𝑉𝑡𝑢𝑛𝑒1 and 𝑉𝑡𝑢𝑛𝑒2 produced by the FSK
oscillator.
Figure 5.4 – Simulation result for the signal 𝑣𝐶 (𝑉𝑡𝑢𝑛𝑒1 − 𝑉𝑡𝑢𝑛𝑒2).
The maximum voltage value for the differential output signals 𝑉𝑡𝑢𝑛𝑒1 and 𝑉𝑡𝑢𝑛𝑒2 is of 494m V, and the
minimum voltage is of 212 mV. The voltage difference between 𝑉𝑡𝑢𝑛𝑒1 and 𝑉𝑡𝑢𝑛𝑒2 is in the interval
[−240, 240] mV, which means that 𝑣𝐶 has an amplitude of about 480 mV. The two 𝑣𝐶 frequencies are
2.166 MHz and 4.442 MHz.
Figure 5.5 demonstrates the synchronization between the signal 𝑣𝐶 and the signal DATA.
66
Figure 5.5 – Simulation result showing the synchronism between the signal DATA and signal 𝑣𝐶.
In figure 5.6 is presented the signal 𝑉𝑜𝑢𝑡1 along with a graphical result of its instantaneous frequency
variation during the simulation.
Figure 5.6 – Simulation result for signal 𝑉𝑜𝑢𝑡1 (including its instantaneous frequency).
The signal 𝑉𝑜𝑢𝑡1 presents also some amplitude modulation. This phenomenon occurs due to the
frequency variation in the VCO. Since it is working at very high frequencies, the VCO output voltage
does not have enough time to reach its maximum and minimum values, (which also grants to the
67
signal an almost sinusoidal behavior). Thus, for higher frequencies the signal has a lower amplitude
and vice-versa. The signal presents a frequency variation between 7.37 GHz and 8.50 GHz and its
maximum and minimum voltage are 1.14 V and 545 mV respectively.
More graphical results concerning to this simulation can be consulted in Annex, figures A4 and A5.
5.2.3. Simulation Results using Pads, Bonding Wires, ESD
protection and Antenna
The signal 𝑉𝑜𝑢𝑡1 has an average amplitude of 425 mV, since the pretended amplitude value for
transmission is of about 50 mV (to respect the imposed EIRP level), a simple common drain stage is
used to drive the load.
This common drain stage is usually used as an isolator [25]. It was implemented with two transistors,
in which transistor M2 is used as a current source, as shown in figure 5.7.
Figure 5.7 – Common drain stage schematic.
Thus, it is guaranteed that the behavior of the VCO will not be affected since this buffer has a high
input impedance ( 𝑖) and low output impedance ( 𝑜). Ignoring the capacitive effects leads to 𝑜 𝑜
that can be given by (5.1) [25]:
𝑜 1 1 𝑜1
1 𝑜2, (5.1)
where 1 is the M1 transconductance, 𝑜1 is the M1 incremental resistance, an incremental
parameter due to body effect and 𝑜2 the dynamic resistance imposed by the simple current source.
68
Considering that 𝑜1 1 1 , 1
1 and 𝑜2 1 1 , equation (5.1) can be rearranged into equation
(5.2):
𝑜 1
, (5.2)
In order to connect the circuit core with the test board it is necessary to use pads and bonding wires,
thus, it is important to test the circuit with these elements, since they can affect the circuit performance
and its respective output signal. Moreover, it is also used electrostatic discharge (ESD) protection
elements to prevent permanent damage in the integrated circuit while it is being tested. ESD is easy to
occur by a simple human touch in the circuit terminals, so it is necessary to add this protection in the
nodes that are connected to the outside.
To implement the pads and the ESD protection elements it is used PAD_RF and DIOP_ESD_RF
elements from the UMC 0.13 µm design kit. The pad parameters are presented in table 5.2. The
schematic of the ESD protection is depicted in figure 5.8 and its dimensioning is shown in table 5.3.
Width [m] Length [m] Multiplier
74.2 µ 80.8 µ 1
Table 5.2 – Pads parameters.
Figure 5.8 – ESD protection schematic.
Width [m] Length [m] Multiplier
600n 120µ 1
Table 5.3 – ESD protection parameters.
69
The bonding wire model used is shown in figure 5.9, and the estimated parameters values are
presented in table 5.4.
Figure 5.9 – Bonding wire model schematic.
Value
R 1 Ω
L 2 nH
C 100 fF
Table 5.4 – Bonding wire parameters.
Another important element for the transmitter simulation is the antenna, once it is necessary to
evaluate the emitted signal. The antenna model used is depicted in figure 5.10, with the values
presented in table 5.5 for a nominal frequency of 8.3 GHz.
Figure 5.10 – Antenna model schematic.
Value
R 50 Ω
L 3.99 nH
C 92 fF
Table 5.5 – Antenna parameters.
70
The schematic showing the transmitter core, buffer, ESD protections, pads, bonding wires and
antenna can be consulted in Annex, figure A6.
Simulation results for the buffer output signal Vout1’ and the antenna input signal Vant_in, (this antenna
input signal also corresponds to the bonding output signal), are presented in figure 5.11.
Figure 5.11 – Simulation result for signal 𝑉𝑜𝑢𝑡1´ and Simulation result for signal 𝑉 𝑡_ .
Due to its filter behavior, it is possible to observe that the bonding wire has some distortion effect in
the signal amplitude leading to a higher attenuation at higher frequencies.
In figure 5.12 is shown the antenna output signal 𝑉𝑎𝑛𝑡_𝑜𝑢𝑡 along with two samples that allows a closer
look into the signal shape and amplitude. Also, in figure 5.13 is depicted the signal 𝑉 _
instantaneous frequency and its respective DFT.
Figure 5.12 – Simulation result for signal 𝑉𝑎𝑛𝑡_𝑜𝑢𝑡.
71
The signal 𝑉 _ presents the expected sinusoidal behaviour. The output buffer parameters were
adjusted to allow a signal 𝑉 _ maximum amplitude of [-36.23, 35.71] mV and a minimum amplitude
of [-17.07, 16.98] mV. The signal average amplitude is of about 53 mV (peak-to-peak), what is
concordant with the pretended value of 50 mV, although this undesired amplitude modulation will
slightly affect the transmitted PSD, as shown in figure 5.13.
Figure 5.13 – Simulation result for signal 𝑉𝑎𝑛𝑡_𝑜𝑢𝑡 instantaneous frequency and DFT.
The transmitted signal instantaneous frequency remains identical to the one concerning to the
signal 𝑉 𝑡1. It presents a frequency variation between 7.33 GHz and 8.50 GHz, corresponding to a
1.17 GHz bandwidth. The highest PSD sample value detected is of -40.937 dBm, corresponding to a
frequency of 7.372 GHz. Despite this value being out of the EIRP level allowed by the FCC, a closer
look in the DFT reveals that only three bins are above the FCC spectral mask. As shown in figure
5.13, the majority of the bins are way below the allowed value of -41.3 dBm, meaning that these three
bins are not significant in the final result.
A summary of the transmitter characteristics obtained with this simulation, including the expected
consumption of all circuits used in its implementation, are shown in table 5.6.
72
FSK Oscillator
Oscillator Frequency 2.166 MHz – 4.442 MHz
Current Consumption 20.084 µA + 35.320 µA + 19.580 µA + 31.633 µA = 106.617 µA
Voltage Controlled Oscillator
Oscillator Frequency From: 7.33 GHz – To: 8.50 GHz
Current Consumption 1.026 mA + 6.092 mA = 7.118 mA
Bandwidth 1.17 GHz
Output Buffer
Current Consumption 0.906 mA
Table 5.6 – Transmitter simulation results.
5.2.4. Simulation Results with High-level Models for Transmission
Channel, Receiver and Digital Decoder
In order to verify if the emitted signal has the same robustness as the one generated by the high-level
transmitter when facing the high-level transmission channel, and if the undesired signal properties,
such as the amplitude modulation and the different impulses duration, (originated by the data
synchronization), do not affect the high-level receiver and the recovered data, it was decided to run a
transmitter simulation along with the previous high-level circuit blocks.
In figure 5.14 is presented the channel output signal Vchannel and the preamplifier output signal Vpre_amp.
Figure 5.14 – Simulation result for signal Vchannel and signal Vpre_amp.
73
In figure 5.15 it is shown the band-pass filter output where the FM signal information is converted into
an AM signal (Vam_pulse) and the respective BDC code (Vbdc) generated by the demodulator.
Figure 5.15 – Simulation result for signal Vam_pulse and signal Vbdc.
Is possible to clearly identify the pulses generated by the band-pass filter according with the signal
frequency variation and as expected the digital decoder could recovered the original data DATA_Rec,
as depicted in figure 5.16.
Figure 5.16 – Simulation result for signal DATA_Rec and comparison with the signals 𝑑(𝑡) and DATA.
This 8.5 µs simulation is not enough to observe any bit error introduced by the signal data
synchronization and thereby the recovered data (DATA_Rec) is identical to the original data 𝑑(𝑡),
despite the delay introduced.
74
5.2.5. Corners
In order to analyze the robustness of an integrated circuit design against the semiconductor
manufacturing variation processes it is usual to test the circuit design for the case of extreme
parameter variations, which are known as process corners. Also, it is common to test if the circuit runs
at lower or higher temperatures and voltages than the specified [26]. To verify the transmitter design
robustness, several simulations were performed corresponding to different Process, Voltage and
Temperature (PVT) corners. The PVT corners used to test the transmitter are presented in table 5.7
and their results are shown in table.5.8.
Transistors Resistors Capacitors Temp [ºC] VDD [V]
Typical Typ Typ Typ Typ Typ
Corner 1 FastFast Max Max 0 1.26
Corner 2 FastFast Min Min 0 1.26
Corner 3 SlowSlow Max Max 100 1.14
Corner 4 SlowSlow Min Min 100 1.14
Corner 5 FastSlow Max Max 0 1.26
Corner 6 FastSlow Min Min 0 1.26
Corner 7 FastSlow Max Max 100 1.14
Corner 8 FastSlow Min Min 100 1.14
Corner 9 SlowFast Max Max 0 1.26
Corner 10 SlowFast Min Min 0 1.26
Corner 11 SlowFast Max Max 100 1.14
Corner 12 SlowFast Min Min 100 1.14
Table 5.7 – PVT corners simulated.
75
FSK Oscillator Voltage Controlled Oscillator
Current Consumption
[µA]
Oscillation Frequency
[MHz]
Current Consumption
[mA]
Oscillation Frequency
[GHz]
Bandwidth [GHz]
Typical 106.617 2.17 – 4.44 7.118 [7.33, 8.50] 1.17
Corner 1 79.694 1.96 – 3.91 6.228 [6.32, 7.43] 1.11
Corner 2 149.980 2.10 – 4.20 13.790 [7.78, 8.43] 0.65
Corner 3 70.221 1.60 – 3.10 3.940 [5.36, 6.06] 0.70
Corner 4 63.968 2.01 – 4.02 – – –
Corner 5 99.42 1.80 – 3.60 5.712 [6.34, 7.16] 0.82
Corner 6 110.920 2.46 – 4.90 12.698 [8.88, 9.96] 1.08
Corner 7 104.921 2.04 – 4.08 4.886 [6.01, 6.81] 0.80
Corner 8 95.112 2.64 – 5.29 – – –
Corner 9 126.592 1.53 – 3.07 4.496 [5.56, 6.61] 1.05
Corner 10 100.398 2.60 – 5.20 11.430 [8.25, 9.30] 1.05
Corner 11 123.598 1.70 – 3.40 3.232 [5.24, 6.21] 0.97
Corner 12 97.721 2.07 – 4.17 – – –
Table 5.8 – Corners simulation results.
From the analyses of the table 5.8 it is possible to observe that there is a wide dispersion in the results
obtained. Despite the use of current sources with variable reference currents, allowing some control in
the oscillator’s frequency, this design variable was not enough to ensure the circuit would work in all
conditions. The FSK oscillator is much more robust to PVT corners and both its current consumption
variation and its oscillation frequency variation are inside acceptable values that will not affect
significantly the transmitter characteristics. The VCO is much more sensitive to corners, mainly due to
the oscillation frequency dependence on the MOS varactors implementation. The lower capacitance
C0, despite being variable, is not dominant for the oscillator frequency, due to its small value. Thus,
facing these design limitations the oscillation conditions were not guaranteed to start the VCO for
corners 4, 8 and 12. These corners share minimum value models for the R and C simultaneously, with
100ºC and supply voltage at a minimum of 1.14V. These conditions can be relaxed depending on the
application, for instance if it is biomedical the temperature can often be limited to a much lower value
and in case we have a smart-power block the supply voltage is also much more controlled than ±5%.
76
In case it is necessary to ensure these corners, a way to overcome these variations is to add a
variable resistor, not implemented, as it is another fundamental variable to control the oscillation
frequency.
Despite corners 4, 8 and 12 for the remaining ones, the minimum bandwidth remains above the
500 MHz, (minimum bandwidth to be considered a UWB signal), and all the bandwidths remain inside
the FCC allocated band for UWB transmission. However, for the lower bandwidths, it is necessary the
use of a signal attenuator before the antenna drive, to respect the imposed EIRP level.
Monte Carlo analyses are also a common test for analog circuits, testing the circuit robustness against
process variations and mismatches. However, at the time of this writing, this type of analysis is
unavailable for 0.130 µm UMC technology with RF models.
5.3. Layout Design
The final step of the FM-UWB transmitter implementation is its layout design. Some important rules
and precautions should be taken in order to improve the circuit reliability:
All connections should be as short and as straight as possible, in order to minimize their
resistances, parasitic inductances and magnetic fields;
Maximization of the connections between metal layers to decrease resistance and guard against
possible manufacturing errors;
The connections to the substrate are distributed along the circuit to reduce parasitic effects and
improve the biasing of the substrate;
All connections should be designed to support higher current than the expected.
For these specific designed circuits, the layout should be made as symmetrical as possible in order to
mitigate eventual asymmetric delays in the signals propagation.
In figure 5.17 is depicted the FM-UWB transmitter without pads, ESD protections and output buffers.
The layout presented has an area of 351 mm x 202 mm ≈ 0.071 mm2 and it is possible to observe the
different circuit blocks that compose the transmitter.
The layout presented in figure 5.18 is the FM-UWB transmitter already including the pads, the ESD
protections and the respective output buffers designed for the VCO output signals. This layout has an
area of 360 mm x 366 mm ≈ 0.132 mm2.
77
Figure 5.17 – FM-UWB transmitter layout without pads, ESD protections and output buffers.
Figure 5.18 – FM-UWB transmitter layout with pads, ESD protections and output buffers.
78
5.4. Conclusions
In this chapter it is described the parameters and changes made to the FSK oscillator and to the VCO
in order to implement the transmitter with real transistor level RF models.
First it was performed a standalone simulation to the transmitter core, analyzed its main
characteristics, such as frequencies, amplitudes, driving signals, wave shapes and bandwidth.
Following, it was performed a simulation considering the output buffer, pads, ESD protections,
bonding wires and also the antenna model. These circuits’ parameters were also presented along with
the circuit current consumption, exhibiting very satisfactory results for the transmitted signal, including
its instantaneous frequency variation, its PSD and its respective bandwidth.
It also was performed a simulation using the transmission channel and receiver high-level circuits,
presented in chapter 3. This simulation also shows very satisfactory results, despite small differences
when compared with the transmitter high-level circuit simulation.
The PVT corners revealed some weaknesses concerning the VCO for a particular group of
occurrences. It should be further evaluated if this combination of factors is indeed required or should
be corrected in future project stages.
Finally, the circuit layout is presented, with an area of ≈ 0,071 mm2 without pads and ≈ 0.132 mm
2 with
pads, fulfilling one of the work goals of designing low-area circuits.
79
6. Conclusion and Future
Work ____________________________________________________
80
6.1. Conclusion
The objective of this work was to implement a radio frequency CMOS transmitter operating around
8 GHz, based on Frequency Modulation in Ultra-Wideband and using a 0.13 µm CMOS technology
It was performed a study about Ultra-Wideband technology basic concepts and how to adapt
traditional narrowband frequency modulation into the modern Ultra-Wideband regulated band, taking
advantage of the properties that this technology has to offer. To fulfill this task a high level model of a
link budget was designed to evaluate the system feasibility. The transmitter, the receiver and a model
of the channel were implemented to allow a first study of the signal characteristics and how to adapt
those according to the established project specifications.
Based on the previous results, a circuit topology was selected to fit the requirements. Once the basic
purpose of the circuit, since the beginning, was to target low cost applications, it was decided to use
inductorless circuits reducing the circuit area and leading to a price decrease. The solution was to use
a relaxation oscillator topology, (based only on resistors and capacitors as passive elements), that
after a brief study revealed that had the potential to be adapted in order to obtain the desired results.
To implement the FSK oscillator, a second capacitance and a pair of switches was added to the RC
oscillator, allowing the possibility of obtain a triangular signal with two different frequencies, capable of
being controlled by the transmitted data with the help of a simple control circuit.
Taking advantage of the differential output produced by the FSK oscillator and using an NMOS and a
PMOS varactor as variable capacitances, it was possible to redesign an RC oscillator and convert it
into a VCO capable of achieving 1.17 GHz of bandwidth.
The standalone simulation of the transmitter core and the simulation including all the circuits and
components needed to broadcast the signal showed very satisfactory results, proving the capabilities
of the designed circuit. Despite the need of some improvements to fulfill all PVT corners, it was shown
that the circuit could comprise a large range of operation.
Finally and after the circuit layout design it was possible to conclude that there is a huge area saving
when compared with circuits using inductors.
Therefore and based on all the results presented in this thesis it is possible to conclude that the
implementation of a radio frequency transmitter in FM-UWB has been successfully achieved.
81
6.2. Future Work
To minimize the current consumption in the transmitter, especially in the VCO, since this circuit is the
critical element concerning the transmitter total power consumption, a different current source
topology should be studied and developed in order to fit the VCO power requirements.
Also, regarding the transmitter power consumption, the possible use of lower power RC oscillators
(RC oscillators that only uses one current source proposed in [18]) should be studied, since during the
time of this work it was not possible to test the circuit behaviour using this modified topology, and a
possible reduction to half the power consumption can be obtained.
To further reduce the die area, a circuit implementation only using MOSFET transistors should be
considered. Replacing the capacitors and resistors in the circuit for MOSFET transistors can reduce
the total area of implementation. Also, when subjected to process variations and mismatches the
transistors typically present lower variations when compared to capacitors and resistors leading this
way to an increase in circuit robustness.
82
83
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87
Annex
88
Figure A1 - Verilog-A code for the attenuator block.
Figure A2 - Verilog-A code for white noise Gaussian source.
Figure A3 - Verilog-A code for band-pass biquadratic filter.
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Figure A4 - Vout1' instantaneous frequency Vs DATA signal.
Regarding Figure A4, is possible to observe that the Vout1' instantaneous frequency transmissions from
2 MHz to 4 MHz, and vice-versa, are well synchronized with the transmitted DATA signal variations.
Figure A5 - Vout1' Power Spectral Density.
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Figure A6 - Circuit schematic with the transmitter core, buffers, ESD protections, pads, bonding wires and antennas.
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Figure A7 - FM-UWB transmitter layout.
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