Reduction of the Envelope Correlation Coefﬁcient with improved total efficiency for mobile LTE

7/22/2019 Reduction of the Envelope Correlation Coefficient with improved total efficiency for mobile LTE

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3280 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 6, JUNE 2013

Reduction of the Envelope Correlation CoefficientWith Improved Total Efficiency for Mobile LTE

MIMO Antenna Arrays: Mutual Scattering ModeShuai Zhang, Andrs Alayn Glazunov, Senior Member, IEEE, Zhinong Ying, Senior Member, IEEE, and

Sailing He, Fellow, IEEE

AbstractA mutual scattering mode is introduced in this paper.Utilizing this mode, the correlation of a lossy long-term evolution(LTE) multiple-input and multiple-output (MIMO) antenna arraycan be reduced efficiently, even down to zero, by increasing the Qfactors of the MIMO antenna elements. In practice, the Q factorscan be straight forwardly tuned throughdifferent input impedancematching. The zero correlation occurs at a Q factor higher thanthat resulting from the conjugate input impedance matching. On

one hand, if the inter-element distance is larger than a certain dis-tance (what we denominate as the Critical Distance), the total effi-ciency can also be improved in additionto reducing the correlation.On the other hand, when the inter-element distance is less than thecritical distance, a reference MIMO antenna with high correlationand high total efficiency is obtained. This antenna can well be pro-posed for over-the-air (OTA) measurement applications. The in-troduced scattering mode is investigated for dual monopoles on alarge lossy ground plane and for various mobile terminal MIMOantenna designs. A wideband MIMO antenna, with multiple res-onances, covering the band 746870 MHz is proposed with theenvelope correlation coefficient and total efficiency less than 0.5and higher than 50% ( 3 dB), respectively. Measurements andsimulations agree well for all the fabricated prototypes. The enve-lopecorrelations and the multiplexing efficiencies of the prototypes

are also investigated in propagation channels with Gaussian dis-tributed angle of arrivals.

Index Termscorrelation, mobile antennas, multiple-input andmultiple-output (MIMO) system, Q factor.

I. INTRODUCTION

I N current and future wireless telecommunications systems,such as the long-term evolution (LTE) and LTE-Advanced,the multiple-input and multiple-output (MIMO) systems are an

integral part of mobile terminals. In the LTE standards, several

new channels are allocated to the lower bands of 700960 MHz.

Manuscript received August 25, 2012; revised November 30, 2012; acceptedJanuary 16, 2013. Date of publicationFebruary22, 2013;date of current versionMay 29, 2013. This work was supported in part by a scholarship within the EUErasmus Mundus External Cooperation Window TANDEM and in part by theSwedish VR grant of the Swedish VR grant (# 621-2011-4620).

S. Zhang and S. He are with the Centre for Optical and ElectromagneticResearch, Zhejiang University, Hangzhou 310058, China, and also with theDepartment of Electromagnetic Engineering, Royal Institute of Technology(KTH), SE-100 44 Stockholm, Sweden (e-mail: [email protected]).

A. A. Glazunov is with the Department of Electromagnetic Engineering,Royal Institute of Technology (KTH), SE-100 44 Stockholm, Sweden.

Z. Ying is with Research and Technology, Corporate Technology Office,SonyMobile Communications AB, SE-221 88 Lund, Sweden.

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2013.2248071

The effectiveness of (MIMO) systems in increasing channel ca-

pacity without the need of more spectrum or power has been

well known for many years [1][3]. In order to guarantee a good

multiplexing MIMO performance, the elements in the MIMO

antenna array should have a low correlation and a high total ef-

ficiency. In general, these requirements are quite challenging to

achieve, due to the limited space in mobile phones and other

small terminals. However, in the higher bands (normally above1.7 GHz), low correlation and good efficiency in small MIMO

terminals can be realized efficiently by reducing the mutual cou-

pling, i.e., the parameter, between the antenna elements,

[4][11]. That is not the case in the lower bands.

This poses new engineering challenges on the practical real-

ization of mobile terminals with good MIMO performance in

these bands. Unlike the higher bands, the wavelengths at the

lower frequencies are much longer and this will result in four

main problems: 1) each MIMO antenna element has to be re-

designed to obtain a compact structure of the device; 2) the

decorrelating structures have to be small enough and still work

well; 3) the MIMO elements and the decorrelating structures aremore closely positioned, causing high correlation and low effi-

ciencies; 4) the chassis mode will be efficiently excited, which

makes the radiation pattern of each MIMO element quite sim-

ilar leading to a very high correlation [23]. Moreover, an en-

velope correlation coefficient (ECC) less than 0.5 and a total

efficiency higher than 40% are rules of thumb for designing

cellular LTE MIMO antennas in the lower bands according to

industrial research reports including field trials and mock ups

[12]. Recently, many studies have been reported attempting dif-

ferent solutions to the problems stated above. For example, the

neutralization line method presented in [5] proposes canceling

the coupling field by providing anotherfield by the line. Since

this method has a less stringent requirement for the size of the

decoupling element, it has become one of the most promising

ways for dealing with coupling at the LTE lower bands. How-

ever, this neutralization line mode is more difficult to excite at

the lower frequencies (700960 MHz) than at the higher fre-

quencies and can only be used in some specific cases. The single

band LTE MIMO antenna in [13] is one of the examples, where

two branches have to be added to excite the neutralization line

mode. Another promising way is the use of decoupling networks

as given in [11], [14], [15], and [22]. It does not require any

physical decoupling structure between two MIMO antenna el-

ements, but only a decoupling network at the end of each port.

However, the result presented in [11] is based on the lossless

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ZHANG et al.: REDUCTION OF THE ENVELOPE CORRELATION COEFFICIENT WITH IMPROVED TOTAL EFFICIENCY 3281

MIMO antenna system assumption. In practice, the losses will

not only change the total efficiency but also the correlation co-

efficient [17]. It has been also reported in [11] that the conjugate

input impedance matching (uncoupled matching network) facil-

itates low antenna correlation for any antenna separation. How-

ever, large efficiency degradation may occur due to the strong

mutual coupling. The coupled matching network in [14], [22]

and the multiport conjugate (MC) matching in [11] can improve

the correlation and efficiency, yet at the expense of bandwidth

reduction.

Additionally, nowadays, most studies dealing with MIMO

antenna performance at the lower LTE bands utilize the S-Pa-

rameters method (or more specifically the mutual coupling eval-

uated with the help of the parameter) to compute the enve-

lope correlation coefficient, [16]and [25]. However, the S-pa-

rameter method for correlation computation presented in [16]

is limited to the lossless antennas case and the 3-D uniform an-

gular distribution of the angle-of-arrival (AoA) of the waves im-

pinging at the antennas. The differences between correlations

obtained with the methods described in [16] and [17] are al-ready quite considerable for radiation efficiencies less than 1

dB (i.e., the common values for the mobile MIMO antennas op-

erating in LTE lower bands, [18]). Therefore, this paper will use

the 3-D E-field radiation patterns as defined in [17] to calculate

all the envelope correlation coefficients instead of S parameters.

In this paper, we introduce a simple method to reduce the

envelope correlation coefficient with improved total efficiency

and large bandwidth for LTE MIMO mobile terminals. The

losses have been taken into account in all the analyses. Gener-

ally, the correlation of a MIMO antenna system is determined

by the antenna element types and inter-element distance.

Here we will illustrate that the antenna Q factor is anotherimportant parameter highly affecting the correlation. Indeed,

by increasing the Q factor, each antenna element will become

a scatterer to the other antenna elements. Consequently, the

radiation patterns will be separated automatically achieving a

low (even zero) correlation without adding any decorrelating

structure into the MIMO system. This result is valid for both

the high and the low radiation efficiency cases. Due to the

losses, the Q factors for zero correlation are usually higher than

those from the best (conjugate) input impedance matching in

[11]. However, in order to optimize MIMO performance and

bandwidth at the same time, the zero correlation may not be

required and the Q factors can be lower than those from the

conjugate input impedance matching. Then, the improved Q

factors can be straightforwardly realized through the better

input impedance matching ( parameters). In our ap-

proach, the and parameters are treated as factors that

affect the total efficiency, but not the correlation coefficient.

Moreover, we define a new parameter: the Critical Distance

at which the relative contribution of in improving the

total efficiency will be equal to that of in reducing the

total efficiency. We show that when the inter-element distance

exceeds the Critical Distance, a better impedance matching can

both reduce the correlation and improve the total efficiency. In

this case, we just need to make the input impedance matching

as wideband and good as possible to improve the bandwidthand multiplexing MIMO performance regardless the mutual

Fig. 1. Dual monopoles on a large ground plane.

TABLE ICAPACITANCE AND LENGTH FOR DIFFERENT MATCHING LEVELS WITH THE

MONOPOLE INTER-ELEMENT DISTANCE OF 50 mm

is shown in Fig. 1

coupling . In the following analysis we will show how

the Critical Distance can be realized in the mobile handset and

propose a wideband mobile LTE MIMO antenna.

II. DUAL MONOPOLES ON A LARGE GROUND

PLANE WITH HIGH LOSSES

In order to explain the mechanism behind the introduced

method in general terms, we considerfirst the caseof two folded

monopoles on a large ground plane as shown in Fig. 1. Thechosen central operating frequency is 740 MHz and the ground

plane is set to 600 mm 600 mm to eliminate the chassis mode

effects. Two different inter-element distances are considered:

mm and mm. The electrical conductivity of

the copper used for the ground plane and monopole has been

set to 100 S/m (high loss condition). The central operating

frequency and input impedance matching levels of monopoles

can be tuned through the length (see Fig. 1) and a capacitor

(with resistive loss of 0.1 ) in series with the discrete port.

For all the matching levels and inter-element distances, the

volume (the smallest half radian sphere) of each monopole

is kept the same. The Q factors in this paper are calculatedaccording to [24, Eq. 96]. The resistance and reactance of the

input impedance required in [24, Eq. 96] is obtained from [11,

Eq. 10], where the load on the non-operating port is set to 50 .

A. Dual Monopoles With Inter-Element Distance mm

and High-Loss Copper (100 S/m)

Let us set the distance between the two monopoles to 50

mm, i.e., a distance less than 0.125 wavelength of the central

frequency. The capacitance and length in Fig. 1 are chosen

according to Table I corresponding to five different matching

levels.

The Q factors for the considered matching levels are shown

in Fig. 2; and as we can clearly see, the Q factor increases aswe go from Matching 1 to Matching 5. The corresponding S


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Fig. 2. Q factors for different impedance matching levels of dual monopoleswith an inter-element distance of 50 mm.

Fig. 3. S Parameters for different impedance matching levels of dualmonopoles with an inter-element distance of 50 mm.

Fig. 4. Envelope correlation coefficients (ECC) with 3-D uniform angular dis-tribution for dual monopoles with an inter-element distance of 50 mm

parameters are shown in Fig. 3. Here, as we go from Matching

1 to Matching 4, the becomes better matched, yet with nar-

rower bandwidth than the former due to the increasing Q fac-

tors. Matching 4 and 5 have the best matching levels and the

highest Q factors around the resonant frequency in all cases.

With this behavior in mind, a quite interesting phenomenon can

be observed in Fig. 4. Indeed, the envelope correlation coeffi-

cient decreases as the Q factors become larger and for Matching

5 it becomes practically zero. This shows that the Q factor is the

parameter that is really affecting the correlation coefficient in

Fig. 5. Radiation efficiency (RE) and total efficiency (TE) for dual monopoleswith an inter-element distance of 50 mm.

Fig. 6. Radiation patterns (realized gain): (a) Antenna 1 (Port 1) in Matching5; (b) Antenna 2 (Port 2) in Matching 5; (c) Antenna 1 in Matching 1; and (d)Antenna 2 in Matching 1.

our case, but not the input impedance matching level as men-

tioned in [11].

The radiation efficiencies and the total efficiencies for dif-

ferent matching levels are presented in Fig. 5. It is worthwhile

mentioning that, throughout the paper, all the computed or mea-

sured values of the total efficiency (TE) include the radiation

effi

ciency, the mismatch and the mutual coupling. Several ob-servations can be made from Fig. 5. First, all the computed ra-

diation efficiencies are close to 1.5 dB as expected for small

terminal MIMO antennas operating at the LTE 700 MHz band.

Second, the total efficiencies of Matching 2 to 5 (higher than

3 dB) are better than that of Matching 1.

The radiation patterns corresponding to Matching 5 (with the

highest Q factor) and Matching 1 (with the lowest Q factor) are

shown in Fig. 6. The radiation patterns for Matching 5 are much

more directional (due to the mutual scattering) than those for

Matching 1, which can explain the low correlation in the highest

Q factor case (see Figs. 2 and 4).

To evaluate the effects of resistive losses on the envelope cor-

relation coefficient, we now introduce a resistor at the end of

each monopole in series with the capacitor and discrete port. The


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Fig. 7. Effects of resistive losses on envelope correlation coefficients.

TABLE IICAPACITANCE AND LENGTH FORDIFFERENT MATCHING LEVELS WITH A

MONOPOLE INTER-ELEMENT DISTANCE OF 20 mm

, and represent the capacitor and the length , respectively.

Fig. 8. Q factors for different impedance matching levels of dual monopoleswith an inter-element distance of 20 mm.

capacitance and the length used here are the same as those in

Matching 3 (see Table I). As can be seen from Fig. 7, the corre-

lation becomes higher as the resistance (i.e., the loss) increases.

This is because high losses lead to low Q factors, which, in turn,

can cause the weak scattering effects.

B. Dual Monopoles With Inter-Element Distance mm

and High Loss Copper (100 S/m)

We now reduce the inter-element distance to mm

with unchanged ground plane geometry and materials. Table II

shows the length and capacitance that are chosen to keep the

central frequency at 740 MHz and achieve different matching

levels. In Fig. 8, the Q factors are shown to increase as we go

from Matching 1 to Matching 3. The corresponding S parame-

ters and envelope correlation coefficients for different matching

levels are presented in Fig. 9 and Fig. 10, respectively. Similarly

to the former case (i.e., mm), the correlation decreases

with the Q factor, but not as significantly as in the former case

due to the much smaller inter-element distance. Fig. 11 shows

how the total efficiencies become smaller as the matching (and

Fig. 9. S Parameters for different impedance matching levels of dualmonopoles with an inter-element distance of 20 mm.

Fig. 10. Envelope correlation coefficients with 3-D uniform angular distribu-tion for dual monopoles with an inter-element distance of 20 mm.

Fig. 11. Comparison of radiation efficiency (RE) and total efficiency (TE) fordual monopoles with an inter-element distance of 20 mm.

the Q factor too) become better. We can therefore conclude that

in this case the inter-element distance mm is less than

the Critical Distance. Hence, in order to improve the total ef-

ficiency in addition to achieving low correlation, the proposed

correlation reduction method should be used for inter-element

antenna distances larger than the Critical Distance.

Based on the results above we conclude that he inter-element

distance mm in the dual monopoles on a large ground

plane case is larger than what we call the Critical Distance. We

will show in Section III that this Critical Distance is not deter-

mined by the physical distance but by the current distribution.


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Fig. 12. Geometries of collocated dual PIFAs for MIMO applications.(unit: mm).

Moreover, when the inter-element distance is less than the Crit-

ical Distance a higher Q factor will produce a lower correlation

but a lower total efficiency as well. In other words, if we reduce

the Q factor in this case, the correlation and the efficiency willincrease simultaneously. We can apply this property to produce

a reference MIMO antenna for MIMO over-the-air (OTA) mea-

surement, which has a high correlation and high efficiency. For

example, two straight monopoles with an inter-element distance

of 15 mm and lengths of 80 mm can be chosen to be mounted on

a copper ground plane with a conductivity of S/m and

a size of 600 mm 600 mm. A 10-nH inductor is in series with

the discrete port. In this case, the envelope correlation coeffi-

cient can be up to 0.925 with a total efficiency of over 3 dB

(50%) at 740 MHz, which is quite effective and useful in the

OTA measurement.

III. DUAL-PIFA MIMO ANTENNAS ON MOBILE CHASSIS

In the previous section, we showed that the correlation of dual

monopoles on a large ground plane decreases by increasing the

Q factor of each monopole. This result is independent of the dis-

tance between the monopoles. In this section, we will apply this

method to mobile handsets and investigate the Critical Distance

on the mobile chassis. Both collocated (same end of the chassis,

see Fig. 12) and separately-located (opposed ends of the chassis,

see Fig. 25) dual PIFAs configurations are investigated next.

A. Collocated Dual PIFAs for MIMO Applications

The configuration of collocated dual PIFAs is shown in

Fig. 12. Two PIFAs are located at the same end of the chassis

with a ground plane of 105 mm 60 mm and a 0.8-mm-thick

FR4 PCB board with the loss tangent of 0.025. The distance

between the two PIFAs is 10 mm. Each PIFA is fed by the

capacitive coupling method (C-fed method). Through changing

the coupling gap between the metal strip and PIFA, different

matching levels of PIFAs can be achieved. On the grounding

point of each PIFA, an inductor with a resistive loss equal to

0.2 is added to tune the central frequency to 740 MHz. In

order to make the PIFAs radiate efficiently, vertical metal walls

with heights of 6 mm are soldered on the arms of the PIFAs.

Table III shows the values of the coupling gap and inductance

for each of the four impedance matching levels considered.

TABLE IIIFEEDING GAP AND INDUCTANCE FOR DIFFERENT MATCHING

LEVELS OF COLLOCATED DUAL PIFAS

and represent the coupling gap (see Fig. 12) and the inductance,

respectively.

Fig. 13. Q Factors for different impedance matching levels of dual PIFAs.

Fig. 14. S parameters for different impedance matching levels of dual PIFAs.

The Q factors and S parameters for different matching levels

are given in Figs. 13 and 14, respectively. As can be seen, the

Q factor increases as we go from Matching 1 to Matching 5;Matching 4 is the best among all matching. As opposed to the

monopole cases, the mutual coupling does not become

stronger as the Q factor increases. Fig. 15 shows that the en-

velope correlation coefficient becomes smaller for the larger Q

factor, as we expected. The radiation patterns of Matching 1 and

Matching 5 at 740 MHz are illustrated in Fig. 16. With a higher

Q factor the MIMO antenna elements scatter the radiation pat-

terns of each other more efficiently. This is because a high Q

factor can make each element very sensitive to the nearfield en-

vironment and enlarge the differences in the radiation patterns of

MIMO antenna elements. The current distributions of Matching

5 and Matching 1 are shown in Fig. 17(a) and (b), respectively,

when Port 1 is operating (see Fig. 10) with Port 2 connected to a

50 cable. The currents on the chassis of Matching 5 are more


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Fig. 15. Envelope correlation coefficients with 3-D uniform angular distribu-tion for dual PIFAs.

Fig. 16. Radiationpatterns(realizedgain)of dual PIFAs:(a) Antenna 1 (Port 1)in Matching 5; (b) Antenna 2 (Port 2) in Matching 5; (c) Antenna 1 in Matching1; and (d) Antenna 2 in Matching 1.

Fig. 17. Currents distributions for: (a) Matching 5 and (b) Matching 1, whenAntenna 1 (Port1) operates with Port 2 terminated.

asymmetrically distributed as compared to those of Matching

1 (see the current distribution within the red circle), which can

lead to more distinct patterns of MIMO antenna elements. In ad-

dition, the currents on the ground plane look very weak; how-

ever, when the antennas operate at frequencies below 1 GHz the

ground plane contributes most of the radiated powers due to its

large current area [23].

Fig. 18. Radiation efficiency (RE) and total efficiency (TE) for differentimpedance matching levels of dual PIFAs in mobile MIMO applications.

Next, we investigate the Critical Distance in the collocated

dual-PIFA case. Fig. 18 shows the radiation efficiencies and the

total efficiencies for different Matching levels. We can observethat as the Matching is improved from 7 dB (Matching 1) to

20 dB (Matching 3), the total efficiency has been enhanced

by more than 0.5 dB in addition to the reduction of correla-

tion. Matching levels from 7 dB to 20 dB are within the

common range for mobile terminal antennas that have multiple

resonances in the lower bands. The efficiencies of Matching 4

and 5 are less than those of Matching 2 and 3 due to the larger

inductance, but still better than Matching 1. Therefore, it can

be concluded that in the collocated dual PIFAs case, the MIMO

inter-element distance of 10 mm is still larger than the Critical

Distance, even though this distance is much smaller than that in

the dual monopoles case with mm. This is because, for a

PIFA, the current at the opening end is much stronger compared

with that at the shorting end. Hence, when the opening ends of

two PIFAs are facing different directions, their currents can also

be separated. As we mentioned above, the Critical Distance is

determined by the current distribution (not by the physical dis-

tance), so the collocated dual PIFAs with 10 mm can still realize

a distance larger than the Critical Distance.

In order to study the improvement of MIMO system perfor-

mance as a function of the Q factor, the multiplexing efficiency

(ME) or is considered next according to [20]. The MIMO

multiplexing efficiency is evaluated for a propagation channel

with Gaussian distribution of the AoA given by (9) in [20]. The

mean incidence direction is denoted by and (as opposed tothe isotropic channel, the likelihood of impinging waves is not

the same inalldirectionsbut has a maximum atAoA and ).

It is assumed that the angular spread is the same in both eleva-

tion and azimuth and approximately equal 30 . We have further

restricted our analysis to channels with balanced polarizations,

i.e., with cross-polarization ratio . For reference, we

have taken two ideal cross-polarized antennas, which give zero

correlation. Following the above conditions, the multiplexing

efficiency is a function of the mean incidence direction:

(1)


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Fig. 19. Multiplexing efficiency (ME) for different impedance matching levelsof dual PIFAs in mobile MIMO applications.

Fig. 20. Configuration of the proposed wideband MIMO antenna array.

where and are the mean effective gains

of each antenna port [21], and is the complex enve-

lope correlation of the received signal. With the assumption of

high SNR and isotropic environment (i.e., equal likelihood of

impinging waves from any direction), and the for-

mula (1) can be simplified to that in [19]:

(2)

where and are the total efficiencies of the MIMO antenna

elements. Fig. 19 then shows the MEs evaluated with (2) as

a function of frequency for different matching levels. As we

can see, increasing the Q factor can improve the ME by almost

2.5 dB at 740 MHz.

As an example of the wideband application of the method

described above, a wideband LTE MIMO antenna is shown in

Fig. 20. As we can see, the MIMO array volume is kept the

same as the one in Fig. 12. However, one additional branch for

each PIFA has been introduced to generate an extra resonance

at the lower frequencies. The resistive losses of capacitors and

inductors are 0.1 and 0.2 , respectively.

Fig. 21 shows the S Parameters, ECC and Efficiency. As

can be seen, the proposed MIMO antenna covers the band

746870 MHz (i.e., 15.35% fractional bandwidth) with an ECC

lower than 0.5 and a total efficiency higher than 3 dB. In ad-

dition, in Fig. 21(c), one may note that with the same matching

level of around 15 dB, the radiation efficiency at 775 MHz is

around 1.2 dB, which is a little higher than that at 740 MHz

(around 1.7 dB) shown in Fig. 8. There are two main reasons

to this: 1) the large distance between feeding a shorting pin

in the wideband design improves the radiation efficiency by

0.2 dB; 2) the use of the lumped capacitor elements to replace

the metal feeding strip in the Fig. 12 improves the radiation

efficiency by 0.30.4 dB. This is because the electric energy

in the high Q capacitor (with a resistive loss of 0.1 ) suffers

from fewer losses than that in the FR 4 substrate (with a loss

tangent of 0.025).

B. Dual PIFAs on the Two Ends of Mobile Chassis for MIMO

Applications

Here we now investigate dual PIFAs on the two opposite ends

of the mobile chassis. The geometries of dual PIFAs are shown

in Fig. 25. The materials are the same as in the collocated caseconsidered above. Also here, three kinds of matching levels are

selected with detailed parameters given in Table IV.

The Q factors, the S parameters and the envelope correlation

coefficients are shown in Fig. 22, Fig. 23 and Fig. 24, respec-

tively. The mutual coupling is now much stronger than in

the collocated case, but as the matching (or Q factor) improves,

the correlation is reduced. Fig. 26 shows that the efficiency be-

comes lower as the matching becomes better. This is because,

although the physical inter-element distance of the MIMO ele-

ments is very large as compared to the collocated case, it is still

smaller than the Critical Distance due to the very strong chassis

mode.

IV. MEASUREMENT RESULTS

In order to verify the proposed method, four collocated dual

PIFAs with different matching levels were fabricated as shown

in Fig. 27.During the fabrication we found that the actual capac-

itances in the C-fed method realized from the metal strip with

gap (see Fig. 12) were around 0.5 pF smaller than the simu-

lated results. It is difficult for Matching 1 and 2 to increase the

capacitance values through further reducing the gap . There-

fore, for Matching 1 and 2 we utilize a high Q lumped capacitor

(with a resistive loss of 0.1 ) directly in series with the feeding

line to replace the coupling metal strip. The detailed capacitancevalues/coupling gaps and inductance are provided in Table V.

During all the measurements, a balun was employed at the

operating port to mitigate the leakage currents on the cable with

the other port terminated with a 50 load. Due to the presence

of the feeding cables and balun, the input impedance is hard to

measure accurately for the purpose of computing the Q factor.

This is because the input impedance measured at the end of the

feeding cable and balun has the same amplitude but different

phase compared with the actual input impedance of each MIMO

element. Furthermore, since the cables have already been sol-

dered to the ground plane and the baluns are not ideal, it is dif-

ficult to measure the exact phase delay. However, according to

[24, Eq.87], the Q factor can also be estimated from the matched

VSWR fractional bandwidth. The measured S parameters for


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Fig. 21. (a) S Parameters, (b) ECC with 3-D uniform angular distribution, and (c) Efficiency of the proposed wideband MIMO antenna.

TABLE IVCOUPLING GAP AND INDUCTOR VALUE FOR DIFFERENT MATCHING LEVELS

OF DUAL PIFAS ON THE TWO ENDS OF CHASSIS

and represent the coupling gap and the inductor value, respectively

(see Fig. 25).

Fig. 22. Q factors for dual PIFAs on the two ends of the mobile chassis.

Fig. 23. S Parameters for dual PIFAs on the two ends of the mobile chassis.

the four fabricated prototypes are shown in Fig. 28. The band-

width measured at the 6 dB level in parameters (or 3 in

VSWR) decreases from Matching 1 to Matching 4, so an in-

creasing Q factor is expected.

Fig. 24. Envelope correlation coefficient with 3-D uniform angular distributionfor dual PIFAs on the two ends of the mobile chassis.

Fig. 25. Geometries of dual PIFAs on the two ends of mobile chassis.

Fig. 29 shows the envelope correlation coefficients computed

based on measured 3D E-field patterns. The radiation patterns

were measured in a Satimo Chamber shown in the left lower

corner in Fig. 27. As can be seen from Fig. 29, the correlation

decreases by increasing the Q factor (indirectly inferred from

Fig. 28), which is in good agreement with simulations presented

above. The radiation gain patterns of Matching 1 and Matching

4 on the -plane (see Fig. 21) are shown in Fig. 30. Clearly, the

radiation patterns of the portsin Matching 4 have been separated

much more compared with those in Matching 1, which explains

the lower correlation for Matching 4 shown in Fig. 29.

The total efficiency of the prototypes has also been mea-

sured and is presented in Fig. 31. As can be seen, the total ef-

ficiencies of Matching 2, 3, and 4 are better than Matching 1


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Fig. 26. Radiation efficiency (RE) and total efficiency (TE) for differentimpedancematching levelsof dual PIFAs on the two ends of themobile chassis.

Fig. 27. Fabricated collocated dual PIFAs and measurement setups.

TABLE VCOUPLING GAPS (OR CAPACITANCE VALUES) AND INDUCTANCE VALUES FOR

DIFFERENT MATCHING LEVELS OF THE COLLOCATED DUAL PIFAS

, and represent the capacitance value/coupling gap and the inductance

value, respectively (see Fig. 12).

at 740 MHz. Moreover, the measured efficiency of all the pro-

totypes is around 0.5 dB lower than the simulations, due to

fabrication imperfections. However, compared with the simu-

lations in Fig. 18, the decrease of the measured total efficiency

of Matching 3 and 4 are a little larger than those of Matching 1

and 2. This is because the coupling metal strips in the prototypes

of Matching 1 and 2 are replaced by the lumped capacitors. As

we mentioned above, this high Q lumped capacitors will intro-

duce fewer losses than the metal strips, while in the simulations

the coupling metal strips are applied for Matching 1 and 2.

The ME was computed according to (2), i.e., under the

isotropic channel assumption, based on the measured total

Fig. 28. Measured S Parameters of fabricated dual PIFAs with differentmatching levels.

Fig. 29. Measured envelope correlation coefficients of fabricated dual PIFAswith different matching levels.

Fig. 30. Radiation gain patterns on the plane for (a) Matching 1 and(b) Matching 2.

efficiency and correlation coefficients. Results are summarized

for different matching levels in Fig. 32. As we can see, mea-

surements also show that the ME has been improved by almost

2.5 dB at 740 MHz. In order to further evaluate the expected

in-channel performance of the prototypes, we have also com-

puted their ME according to (1), i.e., under the non-isotropic

channel assumption. Fig. 33 shows numerical results for dif-

ferent values of and covering the whole sphere showing

a strong dependence of the MIMO multiplexing efficiency on

the spatial distribution of the field impinging at the antennas.


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Fig. 31. Measured efficiencies of fabricated dual PIFAs.

Fig. 32. Multiplexing efficiencies of fabricated dual PIFAs through the mea-sured total efficiencies and envelope correlation coefficients.

Fig. 33. Multiplexing efficiency in the propagation channel with Gaussiandistribution of AoA.A1, A2, A3, and A4 stand for Matching 1, Matching 2,Matching 3, and Matching 4, respectively.

As expected, the performance of Matching 1 (A1 in Fig. 33)

is worse as compared to the other prototypes. Moreover, the

in-channel performance of Matching 4 (A4 in Fig. 33) is ex-

pected to be rather similar to Matching 3 (A3 in Fig. 33) and,

in most cases both are better than Matching 2 (A2 in Fig. 33).

To further illustrate this fact we show results for two specific

cases , , and , in Table VI.

Fig. 34 shows the envelope correlation coefficients in the prop-

agation channel with Gaussian distribution of AoA. The ECC

of Matching 3 (A3) and Matching 4 (A4) are always low for

different and .

TABLE VIMULTIPLEXING EFFICIENCY OF THE PROTOTYPES IN TWO SPECIFIC CASES

Fig. 34. Envelope correlation coeffi

cient in the propagation channel withGaussian distribution of AoA. A1, A2, A3 and A4 stand for Matching 1,Matching 2, Matching 3 and Matching 4, respectively.

From Figs. 33 and 34, we find that the MIMO antenna per-

formance in non uniform AoA can also be improved with the

matching optimized in uniform AoA.

V. CONCLUSION

A generally available mutual scattering mode has been in-

troduced in this paper. This mode can be excited through in-

creasing the Q factors of MIMO antenna elements. In practice,

for a given array, the only two requirements for low correlation

and improved efficiency are a high Q factor (matching) and anelement distance larger than the Critical Distance. This method

has the following advantages:

1) Easy to realize: input impedance matching techniques (in-

cluding impedance matching network and optimizing an-

tenna structure) have been successfully utilized in industry

for many years.

2) No specific requirement for geometry of each MIMO an-

tenna element: i.e., they do not need to be the same and

may have arbitrary structures.

3) Not only valid for the single band, but also for wide band

and multiple bands: In the lower bands the antennas can

have multiple resonances. A wide band and low correlationMIMO antenna can be proposed with improved efficiency.

4) Easy to use together with other known decoupling

methods: our proposed method only requires a high Q

factor or good matching to achieve decorrelation.

5) Possible to use for MIMO antennas with more than two

elements: however, due to the larger coupling between el-

ements the Critical Distance will become larger compared

to the dual-element MIMO antennas. Therefore, if the de-

sign purposes are reducing correlation as well as improving

efficiency, the inter-element distances should be larger than

that in the dual-element case.

6) The conclusions from this paper are also valid for the

on-ground MIMO antennas, i.e., on-ground MIMO

PIFAs.


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The present method has been investigated for both the col-

located and two-end located cases for mobile terminal appli-

cations. A wideband mobile LTE MIMO antenna is proposed.

The prototypes have been fabricated and measurements have

been carried out. All the measured results have agreed well with

the simulations and indicated an improvement of MIMO perfor-

mance through our introduced mode. Furthermore, the purpose

of this paper is to address a mechanism for the correlation re-

duction with the improved total efficiency. Besides the mutual

scattering mode, the correlations will also be affected by the

user [26]. The detailed studies are presented in [27] about how

the user affects the correlation and total efficiency. In addition

to the correlations, one should note that other factors such as

branch power ratio, power absorbed and overall mean effective

gain will also have effects on the perceived channel capacity and

system diversity [26].

ACKNOWLEDGMENT

The authors would like to thank Sony Mobile Communica-tions AB and Ericsson Research for providing the measurement

equipment used for this paper.

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Shuai Zhang was born in Liaoning, China, in 1983.He received the B.E. degree from the Universityof Electronic Science and Technology of China(UESTC), Chengdu, China, in 2007 and the Ph.D.degree in electromagnetic engineering from theRoyal Institute of Technology (KTH), Stockholm,Sweden, in February of 2013.

Currently, he is a Research Fellow in the Depart-ment of Electromagnetic Theory, KTH. In 2010,he obtained an Erasmus Mundus scholarship forhis Ph.D. studies in KTH. From September 2010

to June 2011, he was a Guest Researcher at the Department of Electrical andInformation Technology, Lund University, Sweden. From June 2011 to March2012, he was a Visiting Researcher in the Corporate Technology Office, SonyEricsson Mobile Communication AB, Sweden. His research interests include

ultrawideband (UWB) antennas, MIMO antenna systems, body-centric com-munications, mm-wave antennas, RFID antennas, and multiple antennas-userinteractions.


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Andrs Alayn Glazunov (SM11) was born inHavana, Cuba, in 1969. He received the M.Sc. (En-gineer-Researcher) degree in physical engineeringfrom Saint Petersburg State Polytechnical Univer-sity, Saint Petersburg, Russia, and the Ph.D. degreein electrical engineering from Lund University,Lund, Sweden, from 1988 to 1994 and 2006 to 2009,respectively.

From 1996 to 2001, he was a member of theResearch Staff at Ericsson Research, Ericsson AB,Kista, Sweden, where he conducted research in the

areas of advanced receiver performance evaluation for UMTS, applied elec-tromagnetic wave propagation and stochastic channel modeling for wirelesscommunications systems. During this period, he also contributed to the Euro-pean COST Action 259 project in the directional channel modelling workinggroup. In 2001, he joined Telia Research, Sweden, as a Senior ResearchEngineer. Later, starting in 2003, he held a position as a Senior Specialist inAntenna Systems and Propagation at the newly formed Telia Sonera Sweden,where he pursued research in smart antennas and MIMO, network optimizationand over-the-air (OTA) performance evaluation of handsets and their impacton wireless network performance. From 2001 to 2005, he was the Swedishdelegate to the European COST Action 273 and was active in the handsetantenna working group. He has been one of the pioneers in establishing OTAmeasurement techniques. He has contributed to the EVEREST and NEWCOMEuropean research projects as well as to the 3GPP and the ITU standardization

bodies. From 2009 to 2010, he held a Marie Curie Senior Research Fellowshipat the Centre for Wireless Network Design (CWiND), University of Bedford-shire, UK. Currently, he is a post-doc in the Department of ElectromagneticEngineering, KTH-The Royal Institute of Technology, Stockholm, Sweden. Heis the author of various scientific and technical publications. He is the coauthorand coeditor ofLTE-Advanced and Next Generation Wireless Networks (Wiley2012). His current research interests include, but are not limited to, statisticalsignal processing, electromagnetic theory, fundamental limitations on an-tenna-channel interactions, RF propagation channel measurements, modelingand simulations for network optimization, and OTA testing of wireless devices.

Zhinong Ying (SM05) is an expert of antennatechnology in Network Research Lab. Technologyoffice, Sony Mobile Communication AB, Lund,Sweden. He joined Ericsson AB in 1995. He becameSenior Specialist in 1997 and Expert in 2003 inhis engineer career at Ericsson. His main researchinterests are small antennas, broad and multi-bandantenna, multi-channel antenna (MIMO) system,nearfield and human body effects and measurementtechniques. He has authored and coauthored over 80papers in various of journal, conference and industry

publications. He holds more than 70 patents and pending in the antenna andmobile terminal areas. He contributed a book chapter to the well known Mobile

Antenna Handbook 3rd edition (Artech House, 2008). He had invented anddesigned various types of multi-band antennas and compact MIMO antennasfor the mobile industry. One of his contributions in 1990s is the developmentof nonuniform helical antenna. The innovative designs are widely used inmobile terminal industry. His patented designs have reached a commercialpenetration of more than several hundreds million products in worldwide.He received the Best Invention Award at Ericsson Mobile in 1996 and KeyPerformer Award at Sony Ericsson in 2002. He was nominated for PresidentAward at Sony Ericsson in 2004 for his innovative contributions. He has beena Guest Professor in Zhejiang University, China, since 2002.

Dr. Ying served as TPC Co-Chairmen in International Symposium on An-tenna Technology (iWAT), 2007, and served as session organizer of several in-

ternational conferences including IEEE APS, and a reviewer for several aca-demic journals. He was a member of scientific board of ACE program (AntennaCentre of Excellent in European 6th frame) from 2004 to 2007.

Sailing He (M92SM98F13) received theLicentiate of Technology and the Ph.D. degree inelectromagnetic theory from the Royal Institute ofTechnology (KTH), Stockholm, Sweden, in 1991and 1992, respectively.

Since then he has worked at the same division oftheRoyalInstitute of Technologyas an AssistantPro-fessor, an Associate Professor, and a Full Professor.He is also with Zhejiang University (ZJU, China) asa distinguished professor of a special program orga-

nized by the central government of China, as well asa joint research center between KTH and ZJU. His current research interestsinclude electromagnetic metamaterials, optoelectronics, microwave photonics,and biomedicalapplications. He hasfirst-authored one monograph (Oxford Uni-versity Press) and authored/coauthored about 400 papers in refereed interna-tional journals. He has given many invited/plenary talks in international confer-ences, and has served in the leadership for many international conferences.

Prof. He is a Fellow of Optical Society of America (OSA) and The Interna-tional Society for Optical Engineering (SPIE).

Documents

Reduction of the Envelope Correlation Coefﬁcient with improved total efficiency for mobile LTE