Upload
haha2012
View
28
Download
12
Embed Size (px)
DESCRIPTION
Mixer design
Citation preview
(e) FET Mixers have conversion gain (not loss) Pozar (RF Ch 7)
R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976 Single-Ended FET Mixer
There are several FET parameters that offer nonlinearities used for mixing The strongest is the transconductance gm when the FET is operated in a
common source configuration with a negative gate bias (Vgs )
When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and operates as a linear device
When the gate bias Vgs is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage causing a large change in transconductance
leading to a nonlinear response
2008 copyH-R Chuang EE NCKU
7-2
Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high amp low transconductance states =gt provide mixing as the switching model
(see the Diode Large-Signal Model for Mixer)
RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET
A bypass capacitor at the drain provides a return path for the LO signal and a LPF provides the final IF output signal
Based on the standard unilateral equivalent circuit for a FET
⎩⎨⎧
ω=ω=
tVtvtVtv
LOLOLo
RFRFRFcos)(cos)(
Zg = Rg + jXg Thevenin source impedance for the RF input port ZL = RL + jXL Thevenin source impedance at the IF output port
LO port has a real generator impedance of ZO
=gt since we are not concerned with maximum power transfer for the LO signal
The same as for the large-signal analysis of the diode mixer the LO pumped FET transconductance is espressed as a Fourier series of harmonics of LO signal
suminfin
=
ω+=1
0 cos2)(n
LOn nggtg
not having an explicit formula for the transconductance must rely on measurements for values of ng in the switching model the desired down-conversion is due to (n = 1) only need 1g coefficient amp the typical measured value in the range of 10 mS
2008 copyH-R Chuang EE NCKU
7-3
Conversion gain of the FET mixer can be found as ()
2
22
224
4
RF
IFD
L
Lg
gRF
LLIF
D
availRF
availIFc V
V
Z
RR
RV
ZRV
PPG ===
minus
minus (see ch3 p3-26 conjugate matching formula)
IFDV IF drain voltage
Zg amp ZL chosen for maximum power transfer at the RF and IF ports
The RF signal across the gate-to-source capacitance is given as
)(1)]1()[( gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
VCjZRCj
VV+ω+
=ω++ω
= ()
tVtv RFRF
cRFc ω= cos)( amp
[ ] [ ]+ωω+ω=
ωω+=
ω+=
sum
suminfin
=
infin
=
)cos()cos(2)cos(
)cos()](cos2[)()(
)cos(2)(
10
10
10
ttVgtVg
tVtnggtvtg
tnggtg
LORFRF
cRFRF
c
RFRF
cn
LOnRFcm
nLOn
From
The down-converted IF signal can be extracted from the second term by using
the usual trigonometric identity
)cos(]|)()[( 1 tVgtvtg IFRF
cRFcm IF ω=ω=ω ()
Then the IF component of the drain voltage (in phasor form) is (by using () )
( )
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus=
⎟⎟⎠
⎞⎜⎜⎝
⎛+
minus=minus=
Ld
Ld
gigsRF
RF
Ld
LdRFcLd
RFc
IFD
ZRZR
ZRCjVg
ZRZRVgZRVgV
)(1
)(
1
11
2008 copyH-R Chuang EE NCKU
7-4
The conversion gain GC (before conjugate matching) is then
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus==
Ld
Ld
gigsRFRFIF
DRF
IFD
L
Lgc ZR
ZRZRCj
VgVVV
Z
RRG
)(14
1
2
2 amp from
we have
])[(])()[(2
)(4|
22212
21
2
)(11
2
LLd
L
Cggi
g
gsRF
d
RF
ZRZR
ZRCjV
L
Lg
matchednotc
XRRR
XRR
RCRg
V
g
Z
RRG
gsRF
LdLd
gigsRFRF
++minus++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
⎟⎠⎞⎜
⎝⎛minus
=
ω
++ω+
By conjugately matching the RF amp IF ports
( 01 ==ω== LdLgsRFgig XRRCXRR )
igsRF
d
d
d
i
i
gsRF
dc RC
RgR
RR
RCRgG 2
21
2222
21
42
])0()2[(])0()2[(2
ω=
++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
Practical mixer circuit generally use matching circuits to transform the FET impedance to 50 ohm for the RF LO amp IF ports
2008 copyH-R Chuang EE NCKU
7-5
Diode Large-Signal Model for Mixer (Pozar RF P233)
)1()( minus= αVs eIVI
tVtVVVtv oorrLoRF ω+ω=+= coscos)( For a diode small-signal approximation
( ) ( )( ) ( )
( )
( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡ω+ω+ωminusω+ω+ω++=
ω+ωω+ω=
ω+ω=primerArr
+prime++=+
prime
prime
prime
tVVtVVtVtVVV
tVttVVtV
tVVGv
GvvGItvVI
ororororooroG
ooororrG
oorrG
d
ddoo
rrd
rd
d
cos2cos22cos2cos
coscoscos2cos
coscos2
2)(
22224
22222
22
2
2
signal IF desired
- Usually LO power (typical 5~10 dBm) will violate the small signal approximation of the diode
- Large signal model is needed for fully nonlinear analysis
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-2
Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high amp low transconductance states =gt provide mixing as the switching model
(see the Diode Large-Signal Model for Mixer)
RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET
A bypass capacitor at the drain provides a return path for the LO signal and a LPF provides the final IF output signal
Based on the standard unilateral equivalent circuit for a FET
⎩⎨⎧
ω=ω=
tVtvtVtv
LOLOLo
RFRFRFcos)(cos)(
Zg = Rg + jXg Thevenin source impedance for the RF input port ZL = RL + jXL Thevenin source impedance at the IF output port
LO port has a real generator impedance of ZO
=gt since we are not concerned with maximum power transfer for the LO signal
The same as for the large-signal analysis of the diode mixer the LO pumped FET transconductance is espressed as a Fourier series of harmonics of LO signal
suminfin
=
ω+=1
0 cos2)(n
LOn nggtg
not having an explicit formula for the transconductance must rely on measurements for values of ng in the switching model the desired down-conversion is due to (n = 1) only need 1g coefficient amp the typical measured value in the range of 10 mS
2008 copyH-R Chuang EE NCKU
7-3
Conversion gain of the FET mixer can be found as ()
2
22
224
4
RF
IFD
L
Lg
gRF
LLIF
D
availRF
availIFc V
V
Z
RR
RV
ZRV
PPG ===
minus
minus (see ch3 p3-26 conjugate matching formula)
IFDV IF drain voltage
Zg amp ZL chosen for maximum power transfer at the RF and IF ports
The RF signal across the gate-to-source capacitance is given as
)(1)]1()[( gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
VCjZRCj
VV+ω+
=ω++ω
= ()
tVtv RFRF
cRFc ω= cos)( amp
[ ] [ ]+ωω+ω=
ωω+=
ω+=
sum
suminfin
=
infin
=
)cos()cos(2)cos(
)cos()](cos2[)()(
)cos(2)(
10
10
10
ttVgtVg
tVtnggtvtg
tnggtg
LORFRF
cRFRF
c
RFRF
cn
LOnRFcm
nLOn
From
The down-converted IF signal can be extracted from the second term by using
the usual trigonometric identity
)cos(]|)()[( 1 tVgtvtg IFRF
cRFcm IF ω=ω=ω ()
Then the IF component of the drain voltage (in phasor form) is (by using () )
( )
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus=
⎟⎟⎠
⎞⎜⎜⎝
⎛+
minus=minus=
Ld
Ld
gigsRF
RF
Ld
LdRFcLd
RFc
IFD
ZRZR
ZRCjVg
ZRZRVgZRVgV
)(1
)(
1
11
2008 copyH-R Chuang EE NCKU
7-4
The conversion gain GC (before conjugate matching) is then
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus==
Ld
Ld
gigsRFRFIF
DRF
IFD
L
Lgc ZR
ZRZRCj
VgVVV
Z
RRG
)(14
1
2
2 amp from
we have
])[(])()[(2
)(4|
22212
21
2
)(11
2
LLd
L
Cggi
g
gsRF
d
RF
ZRZR
ZRCjV
L
Lg
matchednotc
XRRR
XRR
RCRg
V
g
Z
RRG
gsRF
LdLd
gigsRFRF
++minus++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
⎟⎠⎞⎜
⎝⎛minus
=
ω
++ω+
By conjugately matching the RF amp IF ports
( 01 ==ω== LdLgsRFgig XRRCXRR )
igsRF
d
d
d
i
i
gsRF
dc RC
RgR
RR
RCRgG 2
21
2222
21
42
])0()2[(])0()2[(2
ω=
++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
Practical mixer circuit generally use matching circuits to transform the FET impedance to 50 ohm for the RF LO amp IF ports
2008 copyH-R Chuang EE NCKU
7-5
Diode Large-Signal Model for Mixer (Pozar RF P233)
)1()( minus= αVs eIVI
tVtVVVtv oorrLoRF ω+ω=+= coscos)( For a diode small-signal approximation
( ) ( )( ) ( )
( )
( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡ω+ω+ωminusω+ω+ω++=
ω+ωω+ω=
ω+ω=primerArr
+prime++=+
prime
prime
prime
tVVtVVtVtVVV
tVttVVtV
tVVGv
GvvGItvVI
ororororooroG
ooororrG
oorrG
d
ddoo
rrd
rd
d
cos2cos22cos2cos
coscoscos2cos
coscos2
2)(
22224
22222
22
2
2
signal IF desired
- Usually LO power (typical 5~10 dBm) will violate the small signal approximation of the diode
- Large signal model is needed for fully nonlinear analysis
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-3
Conversion gain of the FET mixer can be found as ()
2
22
224
4
RF
IFD
L
Lg
gRF
LLIF
D
availRF
availIFc V
V
Z
RR
RV
ZRV
PPG ===
minus
minus (see ch3 p3-26 conjugate matching formula)
IFDV IF drain voltage
Zg amp ZL chosen for maximum power transfer at the RF and IF ports
The RF signal across the gate-to-source capacitance is given as
)(1)]1()[( gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
VCjZRCj
VV+ω+
=ω++ω
= ()
tVtv RFRF
cRFc ω= cos)( amp
[ ] [ ]+ωω+ω=
ωω+=
ω+=
sum
suminfin
=
infin
=
)cos()cos(2)cos(
)cos()](cos2[)()(
)cos(2)(
10
10
10
ttVgtVg
tVtnggtvtg
tnggtg
LORFRF
cRFRF
c
RFRF
cn
LOnRFcm
nLOn
From
The down-converted IF signal can be extracted from the second term by using
the usual trigonometric identity
)cos(]|)()[( 1 tVgtvtg IFRF
cRFcm IF ω=ω=ω ()
Then the IF component of the drain voltage (in phasor form) is (by using () )
( )
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus=
⎟⎟⎠
⎞⎜⎜⎝
⎛+
minus=minus=
Ld
Ld
gigsRF
RF
Ld
LdRFcLd
RFc
IFD
ZRZR
ZRCjVg
ZRZRVgZRVgV
)(1
)(
1
11
2008 copyH-R Chuang EE NCKU
7-4
The conversion gain GC (before conjugate matching) is then
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus==
Ld
Ld
gigsRFRFIF
DRF
IFD
L
Lgc ZR
ZRZRCj
VgVVV
Z
RRG
)(14
1
2
2 amp from
we have
])[(])()[(2
)(4|
22212
21
2
)(11
2
LLd
L
Cggi
g
gsRF
d
RF
ZRZR
ZRCjV
L
Lg
matchednotc
XRRR
XRR
RCRg
V
g
Z
RRG
gsRF
LdLd
gigsRFRF
++minus++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
⎟⎠⎞⎜
⎝⎛minus
=
ω
++ω+
By conjugately matching the RF amp IF ports
( 01 ==ω== LdLgsRFgig XRRCXRR )
igsRF
d
d
d
i
i
gsRF
dc RC
RgR
RR
RCRgG 2
21
2222
21
42
])0()2[(])0()2[(2
ω=
++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
Practical mixer circuit generally use matching circuits to transform the FET impedance to 50 ohm for the RF LO amp IF ports
2008 copyH-R Chuang EE NCKU
7-5
Diode Large-Signal Model for Mixer (Pozar RF P233)
)1()( minus= αVs eIVI
tVtVVVtv oorrLoRF ω+ω=+= coscos)( For a diode small-signal approximation
( ) ( )( ) ( )
( )
( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡ω+ω+ωminusω+ω+ω++=
ω+ωω+ω=
ω+ω=primerArr
+prime++=+
prime
prime
prime
tVVtVVtVtVVV
tVttVVtV
tVVGv
GvvGItvVI
ororororooroG
ooororrG
oorrG
d
ddoo
rrd
rd
d
cos2cos22cos2cos
coscoscos2cos
coscos2
2)(
22224
22222
22
2
2
signal IF desired
- Usually LO power (typical 5~10 dBm) will violate the small signal approximation of the diode
- Large signal model is needed for fully nonlinear analysis
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-4
The conversion gain GC (before conjugate matching) is then
⎟⎟⎠
⎞⎜⎜⎝
⎛+⎟
⎟⎠
⎞⎜⎜⎝
⎛
+ω+minus==
Ld
Ld
gigsRFRFIF
DRF
IFD
L
Lgc ZR
ZRZRCj
VgVVV
Z
RRG
)(14
1
2
2 amp from
we have
])[(])()[(2
)(4|
22212
21
2
)(11
2
LLd
L
Cggi
g
gsRF
d
RF
ZRZR
ZRCjV
L
Lg
matchednotc
XRRR
XRR
RCRg
V
g
Z
RRG
gsRF
LdLd
gigsRFRF
++minus++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
⎟⎠⎞⎜
⎝⎛minus
=
ω
++ω+
By conjugately matching the RF amp IF ports
( 01 ==ω== LdLgsRFgig XRRCXRR )
igsRF
d
d
d
i
i
gsRF
dc RC
RgR
RR
RCRgG 2
21
2222
21
42
])0()2[(])0()2[(2
ω=
++⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
Practical mixer circuit generally use matching circuits to transform the FET impedance to 50 ohm for the RF LO amp IF ports
2008 copyH-R Chuang EE NCKU
7-5
Diode Large-Signal Model for Mixer (Pozar RF P233)
)1()( minus= αVs eIVI
tVtVVVtv oorrLoRF ω+ω=+= coscos)( For a diode small-signal approximation
( ) ( )( ) ( )
( )
( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡ω+ω+ωminusω+ω+ω++=
ω+ωω+ω=
ω+ω=primerArr
+prime++=+
prime
prime
prime
tVVtVVtVtVVV
tVttVVtV
tVVGv
GvvGItvVI
ororororooroG
ooororrG
oorrG
d
ddoo
rrd
rd
d
cos2cos22cos2cos
coscoscos2cos
coscos2
2)(
22224
22222
22
2
2
signal IF desired
- Usually LO power (typical 5~10 dBm) will violate the small signal approximation of the diode
- Large signal model is needed for fully nonlinear analysis
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-5
Diode Large-Signal Model for Mixer (Pozar RF P233)
)1()( minus= αVs eIVI
tVtVVVtv oorrLoRF ω+ω=+= coscos)( For a diode small-signal approximation
( ) ( )( ) ( )
( )
( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡ω+ω+ωminusω+ω+ω++=
ω+ωω+ω=
ω+ω=primerArr
+prime++=+
prime
prime
prime
tVVtVVtVtVVV
tVttVVtV
tVVGv
GvvGItvVI
ororororooroG
ooororrG
oorrG
d
ddoo
rrd
rd
d
cos2cos22cos2cos
coscoscos2cos
coscos2
2)(
22224
22222
22
2
2
signal IF desired
- Usually LO power (typical 5~10 dBm) will violate the small signal approximation of the diode
- Large signal model is needed for fully nonlinear analysis
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-6
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-7
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-8
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-9
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-10
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-11
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-12
EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer
tgtggtg LOLO ω+ω+= 2cos2cos2)( 210
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
minusminus
minus
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡=
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
gIM
IFIF
gRFRF
IM
IF
RF
RIRI
RIV
ggggggggg
III
012
101
210
gg
RFRIFSC RgRg
VgIIIF 20
10 1 ++=minus= =
)1(2 002021
10
ggg
RFIIFOC
RggRggRgVgVV
IF +minusminus== =
gg
g
OC
SCIF RgRg
Rgg
VIG
20
21
0 12
++minus==
IF
SCavailIF G
IP
4
2=minus
g
RFavailRF R
VP
4
2=minus
[ ]g
ggggg
availIF
availRFc
Rg
RgRgRggRgRgPPL 2
1
2120020 2)1()1( minus++++
==minus
minus
bxbaxaxLc
))((2 minus++=
)( baaxopt minus=
[ ][ ] [ ]abab
bbaaa
baabbaababaaa
Lc 11112
)(2)(
)()(22
min minusminusminus+
=minus+
=minus
minus+minusminus+=minus
LO
VsLOnsn V
eIVIIgLO
πααcongαα=
α
2)(
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-13
Mixer Conversion Noise 1dB 3rd Order Type Gain Figure Compression Intercept Diode -5 dB 5-7dB -6 to ndash1 dBm 5 dBm FET 6 dB 7-8dB 5 to 6 dBm 20 dBm
1(
2
)200
21 cong+
=ggg
gab
)1(2
minus= ce LnTT
tnnn
tg LOn
ωπ
π+= sum
infin
=cos
2sin2
21)(
1
[ ]⎥⎥⎦
⎤
⎢⎢⎣
⎡ωminusω+ω+ω
ππ
+ω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡ωω
ππ
+ω==
sum
suminfin
=
infin
=
tntnn
tV
ttnnn
ttvtgti
LORFRFRFn
RFRF
RFLOn
RFRF
)cos()cos(2
sin2cos21
coscos2
sin2cos21)()()(
1
1
suminfin
=ω+=
10 cos2)(
nLOn nggtg
2
22
2
2
4
4RF
IFD
L
Lg
g
RF
L
LIF
D
availRF
availIFc V
V
Z
RR
RV
Z
RV
PPG ===
minus
minus
( ) )(1 gigsRF
RF
gsRFgigsRF
RFRFc ZRCj
V
CjZRCj
VV+ω+
=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
ωminus+ω
=
+ωω+ω= ttVgtVgtvtg LORFRF
cRFRF
cRFcm coscos2cos)()( 10
tVgtvtg IFRF
cRFcm RF
ω=ω cos|)()( 1
⎟⎟⎠
⎞⎜⎜⎝
⎛++ω+
minus=⎟⎟
⎠
⎞⎜⎜⎝
⎛+
minus=Ld
Ld
gigsRF
RF
Ld
LdRFc
IFD ZR
ZRZRCj
VgZR
ZRVgV)(1
11
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-14
Using the small-signal approximation of (720) gives the total diode current as The first term in (724) is the DC bias current which will be blocked from the IF by the DC blocking
capacitors The second term is a replication of the RF and LO signals which will be filtered out by the low-pass IF filter This leaves the third term can be rewritten using trigonometric identities as
This result is seen to contain several new signal components only one of which produces the desired
IF difference product The two DC terms again will be blocked by the blocking capacitors and the 2ampgtRF 2ampgtLo^ and amp)RF + (ULO terms will be blocked by the low-pass filter This leaves the IF output current as
where (UIF = O)RF a LO is the IF frequency The spectrum of the down- converting single-ended
mixer is thus identical to that of the idealized mixer shown in Figure 71b Large-Signal Model While the small-signal analysis of a mixer demonstrates the key process of frequency conversion
it is not accurate enough to provide a realistic result for conversion loss This is primarily because the power supplied to the mixer LO port is usually large enough to violate the small-signal approximation Here we consider a fully nonlinear analysis of a resistive diode mixer [3]-[4] with the goal of deriving an expression for the conversion loss defined in (710) The term resistive in this context means that reactances associated with the diode junction and package are ignored to simplify the analysis Our results should be useful in understanding the nonlinear operation and losses of the diode mixer but for actual design purposes modem computer-aided design (CAD) software is preferred [5]Such software can model the diode nonlinearity as well as the effects of diode reactances and impedance matching networks
We again assume a diode I-V characteristic as given by (718) with a relatively low-level RF input voltage given by (722) and a much larger LO pump signal given by (723) A DC bias current may also be present but will not directly enter into our analysis As we have seen from the small-signal mixer analysis these two AC input signals generate a multitude of harmonics and other frequency products
amp)RF RF input signal (low power) ampIIF = ampIRF LO IF output signal (low power) (DIM = (ULO t^iF image signal (low power) (ULO LO input signal (high power)
( )( )[ ]222
2
21
1
2|LLd
L
gsRFggi
g
gsRF
d
matchednotc
XRRR
CXRR
RCRgG
++
⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡
⎟⎟⎠
⎞⎜⎜⎝
⎛
ωminus++
⎟⎟⎠
⎞⎜⎜⎝
⎛
ω=
RiCRgG
gsRF
dc 22
21
4ω=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-15
gtLO harmonics of LO (high power) racfLO MIF harmonic sidebands of LO (low power)
is leaves the IF output current as In a typical mixer harmonics of the LO and the harmonic sidebands are terminated reactively
and therefore do not lead to much power loss This leaves three signal frequencies of most importance ampIRF agtw and n)iw To evaluate conversion loss we will find the available power of the RF input signal the power of the IF output signal and the power lost in the image signal The image signal is important because it is relatively close in frequency to the RF signal and thus sees essentially the same load We will see that approximately half the input power gets converted to the image frequency Note that the image term at frequency ampgtIM = WLO - RF = ^LO - lt^RF was not explicitly shown in the small-signal expansion of (724) since this product is generated by the u3 term of (720)
Under the assumption that the RF input voltage is small we can write the AC diode current as a Taylor series expansion about the LO voltage as
This Taylor series is similar in form to (720) as used for the small-signal analysis but with the
important difference that the expansion point here is about the LO voltage where as(720) was expanded about the DC bias point The first term in (726) is due only to the LO input and does not enter into the calculation of conversion loss The second term is a function of the RF and LO input voltages and will provide a good approximation for the three products at frequencies ww w^ and ampgtIM with a large LO pump signal The coefficient of the second term has dimensions of conductance so we can use (718) to write the differential conductance as
Then for small input voltages v(t) we can write the resulting diode current as We see from (727) that g(t) is a real number (consistent with our description as a resistive mixer)
and is a periodic function of the LO frequency Thus g(t) can be expressed as a Fourier cosine series in terms of harmonics of w^o-
with Fourier coefficients given by where In(x) is the modified Bessel function of order n defined in Appendix B Now let the AC
diode current consist of three components at the frequencies amp)RF (DIF and where pp IF and [M are the amplitudes of the RF IF and image signals to be determined If the
RF voltage of (722) is applied to the diode through a source resistance Rg and the IF and image ports are terminated in load resistances ip and Rg respectively then the voltage
FIGURE 74 Equivalent circuit for the large-signal model of the resistive diode mixeracross the
diode can be written as (732) The equivalent circuit consists of a three-port network with one port for each of the frequency
components at amp)RF agtip and agtm[ as shown in Figure 74 We assume the terminations for the RF and image ports are identical because aiRp is very close to ugtwi while the termination for the IF port may be different
Using the first three terms of the Fourier series of (729) for the diode differential conductance gives
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-16
(733) Multiplying the voltage of (732) by the conductance in (733) and matching like frequency
terms with the current of (731) gives a system of three equations for the unknown port currents (734) where VRF is the source voltage and the gn s are defined in (730) Note that multiplication of
(732) by (733) creates several frequencies in addition to oipp (UIF and (UIM but we assume these frequencies to be reactively terminated so that they do not lead to significant power dissipation
The easiest way to find the available power from the IF port is to first find the Norton equivalent source for the IF port As shown in Figure 75 this consists of a current source equal to sc the short-circuit current at the IF port and Gip the conductance seen looking into the IF port This conductance can be found as GIF = sc Voc where Voc is the open-circuit voltage of the IF port The short-circuit IF port current can be found by setting R^p == 0 in (734) and solving for ip After some straightforward algebra we obtain
FIGURE75 Norton equivalent circuit for the IF port of the large-signal model of the resistive diode mixer
The open-circuit IF port voltage is found by setting ip = 0 and solving (734) for Vy Then the Norton conductance of the IF port is The available output power at the IF port is and the available input power from the RF source is So from (710) the conversion loss is (not in dB) Note that the conversion loss does not depend on the IF port termination ip because of the use of
available powers It does depend on Rg the RF and image port terminations so it is possible to minimize the conversion loss by properly selecting Rg If we let x = I a = go + g2and b = 2g^go then (739) can be rewritten as
(740) Differentiating with respect to x and setting the result to zero gives the optimum value of x as for which the minimum value of conversion loss is (741) We can evaluate this result by approximating values for go gi and g^ For an LO input power of
10 mW VLO is about 0707 V rms and a = 128 mV so aV^n the argument of the modified Bessel functions for gn given in (730) is approximately 25 Thus the modified Bessel functions can be approximated asymptotically using the large-argument formula given in Appendix B and the gns simplified as
and the minimum conversion loss of (742) reduces to L( = 2 or 3 dB This means that half the
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-17
RF input power is converted to IF power and half is converted to power at the image frequency In principle this result could be improved by terminating the image port with a reactive load but it is usually difficult in practice to separate the image termination from the RF termination Also this result is highly idealized in that it assumes no power loss at higher harmonic frequencies and it ignores diode reactances
This same model can be used to derive the SSB noise temperature for the resistive mixer as (745) where n is the diode ideality factor and T is the physical temperature of the diode [3] Switching Model The large-signal model suggests that the diode mixer can be viewed as a switch As the LO
voltage cycles between positive and negative values of cosecant the diode becomes conducting or nonconducting respectively Thus the diode conductance (the ratio of diode current to diode voltage) switches between large values and zero at the same rate as the LO voltage Figure 76 shows a typical diode conductance waveform where T = ITCJW^O is the period of the LO waveform
The conductance waveform of Figure 76 can be calculated directly from the diode V-I characteristic of (718) or from the Fourier series representation of (729) But since a conductance greater than a few Siemens is essentially a short circuit we can approximate the diode conductance as the square wave shown in Figure 77 This square wave has a Fourier transform given by
(746) which is similar in form to the Fourier series of (729) An equivalent circuit of the diode mixer then consists of the RF input voltage applied across a
load resistor in series with an ideal switch as shown in Figure 78 The time-varying FIGURE 76 Conductance waveform of a mixer diode pumped with a large-signal FIGU運77 LO voltage waveform and idealized square-wave diode conductance waveform for the switching model of a diode mixer switch conductance is given by (746) The diode current can be found by multiplying the RF input
voltage of (722) by the conductance of (746) Filtering all but the lowest-frequency component for the n = 1 term of the summation gives the
desired IF output as The switching model is useful for mixers of any type including the FET mixer discussed in the
following section Note that the switching model of a mixer can be considered as a linear but time-varying circuit
FIGURE 78 Equivalent circuit for the switching model of the diode mixer 73 - FET MIXERS Mixers can also be implemented by using the nonlinear properties of transistors FETs in
particular offer low noise characteristics and easy integration with other circuitry such as switches and low-noise amplifiers Transistor mixers can provide conversion gain but their noise figure is generally not as good as can be obtained with diode mixers PET mixers also offer
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-18
higher dynamic range The following table compares the characteristics of typical diode and FET mixers
Because a FET mixer has conversion gain but usually worse noise figure the proper comparison with a diode mixer should include the cascade effect of adjacent stages
In this section we will analyze the single-ended FET mixer and derive an expression for its conversion gain We will also discuss a few other popular FET mixer configurations
Single-Ended FET Mixer There are several FET parameters that offer nonlinearities that can be used for mixing but the
strongest is the transconductance ^ when the FET is operated in a common source configuration with a negative gate bias Figure 79 shows the variation of transconductance with gate bias for a typical FET When used as an amplifier the gate bias voltage is near zero or positive so the transconductance is near its maximum value and the transistor operates as a linear device When the gate bias is near the pinch-off region where the transconductance approaches zero a small variation of gate voltage can cause a large change in transconductance leading to a nonlinear response Thus the LO voltage can be applied to the gate of the FET to pump the transconductance to switch the FET between high and low transconductance states and provide mixing in much the same manner as the switching model discussed in the previous section
The circuit for a single-ended FET mixer is shown in Figure 710 A diplexing coupler is used to combine the RF and LO signals at the gate of the FET An impedance matching net-work is also usually required between the inputs and the FET which typically presents a verylow input impedance RF chokes are used to bias the gate at a negative voltage near pinch-off and to provide a positive bias for the drain of the FET A bypass capacitor at the drain pro-vides a return path for the LO signal and a low-pass filter provides the final IF output signal
FIGURE 79 Variation of FET transconductance versus gate-to-source voltage
FIGURE 710 Circuit for a single-ended FET mixer
Our analysis of the mixer of Figure 710 follows the original work described in reference [6] The simplified equivalent circuit is shown in Figure 711 and is based on the standard unilateral equivalent circuit for a FET The RF and LO input voltages are given in (722) and (723) Let Zg = Rg + jXg be the Thevenin source impedance for the RF input port and ZL = RL + JX^ be the Thevenin source impedance at the IF output port These impedances are complex to allow us to conjugately match the input and output ports for maximum power transfer The LO port has a real generator impedance of Zo since we are not concerned with maximum power transfer for the LO signal
As we did for the large-signal analysis of the diode mixer we express the LO pumped FET transconductance as a Fourier series in terms of harmonics of the LO signal
Because we do not have an explicit formula for the transconductance we cannot calculate directly
the Fourier coefficients of (748) but must rely on measurements for these values As in the case of the switching model the desired down-conversion result is due to the n = 1 term of the Fourier series so we only need the g coefficient Measurements typically give a value in the range of 10 mS for g
The conversion gain of the FET mixer can be found as where V^f is the IF drain voltage and the impedances Zg and Z^ are chosen for maximum power
transfer at the RF and IF ports The RF frequency component of the phasor voltage FIGURE 711 Equivalent circuit for the FET mixer for Figure 710
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-19
across the gate-to-source capacitance is given in terms of the voltage divider between Z RSindCgs
Multiplying the transconductance of (748) by ^(r) = V^COSCURF gives terms oftr form The down-converted IF frequency component can be extracted from the second ten of (751) using the usual trigonometric identity Then the IF component of the drain voltage is in phasor form where (750) has been used Using this result in (749) gives the conversion gain (before conjugate
matching) as We must now conjugately match the RF and IF ports Thus we let Rg = RiX^ = loiRpCg RL = Rd
and XL = 0 which reduces the above result to The quantities gt R^ R and Cg are all parameters of the FET Practical mixer circuit generally use
matching circuits to transform the FET impedance to 50 ^ for the RF LC and IF ports EXAMPLE 72 MIXER CONVERSION GAIN A single-ended FET mixer is to be designed A single-ended FET mixer is to be designed for a wireless local area network receiver operating at
24 GHz The parameters of the FET are Rd = 300 R = 100 Cgs =03 pF and g = 10 mS Calculate the maximum possible conversion gain
Solution This is a straightforward application of the formula for conversion gain given in (754) Note that this value does not include losses due to the necessary impedance matching networks FIGURE 712 A dual-gate FET mixer Other FET Mixers There are several variations of mixer circuits that can be implemented using FET Figure 712 shows a single-ended mixer using a dual-gate FET where the RF and LO inputs
are applied to separate gates of the PET This provides a high degree of RF-LO isolation generally an inferior noise figure relative to the transconductance mixer of Figure 710
Another configuration is shown in Figure 713 using two FETs in a differential amplifier configuration The balun (balanced-to-unbalanced) networks on the LO and IP ports provide a transition between a two-wire line that is balanced with respect to ground and a single line that is unbalanced relative to ground Baluns may be implemented with center-tapped transformers or with 180 hybrid junctions
The differential mixer operates as an altenating switch with the LO turning the top two FETs on and off on alternate cycles of the LO These PETs are biased slightly above pinch-off so each PET will be conducting for slightly more than half of each LO cycle Thus one of the upper FETs is always conducting and the lower FET will remain in saturation The RF and LO ports should each be impedance matched The IF output circuit must provide a return path to ground for the LO signal
An extension of the differential FET mixer is the Gilbert cel1 mixer show in Figure 714 This mixer us differential FET mixer stages to form a double balanced mixer This circuit
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-20
achieves high RF-LO isolation and a high dynamic range It also cancels all even-order intermodulation products This circuit is very popular for wireless integrated circuits
FIGURE 713 A differential FET mixer
FIGURE 714 A Gilbert cell mixer 74 OTHER MIXER CIRCUITS The single-ended diode and FET mixers discussed above provide frequency conversion but often
have poor RF input matching and RF-LO isolation This reduces the performance of wireless systems but fortunately it is possible to improve these characteristics by combining two or more single-ended mixers with hybrid junctions
Balanced Mixers RF input matching and RF-LO isolation can be improved through the use of a balanced mixer
which consists of two single-ended mixers combined with a hybrid junction Figure 715 shows the basic configuration with either a 90hybrid (Figure 715a) or a 180hybrid (Figure 715b) As we will see a balanced mixer using a 90hybrid junction will ideally lead to a perfect input match at the RF port over a wide frequency range while the use of a 180hybrid will ideally lead to perfect RF-LO isolation over wide frequency range In addition both mixers will reject all even-order intermodulation products
Microwave quadrature or ring hybrids [1] can be used to implement balanced mixers but at lower frequencies a center-tapped transformer can be used As shown in Figure 716 the secondary of the transformer provides outputs with a 180phase shift to the two mixer diodes The LO signal is applied to the center tap of the secondary
The double-balanced mixer of Figure 717 uses two hybrid junctions or transformers and provides good isolation between all three ports as well as rejection of all even harmonics of the RP and LO signals This leads to very good conversion loss but less than ideal input matching at the RF port The double-balanced mixer also provides a higher third-order intercept point than either a single-ended mixer or a balanced mixer
FIGURE 715 Balanced mixer circuits (a) Using a 90hybrid (b) Using a 180 hybrid
FIGURE 716 Balanced mixer using a hybrid transformer FIGURE 717 Double-balanced mixer circuit The following table summarizes the characteristics of several types of mixers Small-Signal Analysis of the Balanced Mixer We can analyze the performance of a balanced mixer using the small-signal approachthat was
used in Section 72 Here we will concentrate on the balanced mixer with a 90 hybrid shown in Figure 715a and leave the 180hybrid case as a problem
As usual let the RF and LO voltages be defined as The scattering matrix for the 90hybrid junction is [1] where the ports are numbered as shown in Figure 715a Then the total RF and LO voltages applied
to the two diodes can be written as Using only the quadratic term from the small-signal diode approximation of (720) gives the diode
currents as
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-21
where the negative sign on (2 accounts for the reversed diode polarity and K is a constantfor the quadratic term of the diode response Adding these two currents at the input to the low-pass filter gives
where the usual trigonometric identities have been used and ampIIF = WRF (ULO is the IF
frequency Note that the DC components of the diode currents cancel upon combining After low-pass filtering the IF output is
as desired We can also calculate the input match at the RF port and the coupling between the RF and LO
ports If we assume the diodes are matched and each exhibits a voltage reflection coefficient F at the RF frequency then the phasor expression for the reflected RF voltages at the diodes will be
and (761a)
These reflected voltages appear at ports 2 and 3 of the hybrid respectively and combine to form the following outputs at the RF and LO ports
Thus we see that the phase characteristics of the 90hybrid lead to perfect cancellation of reflections
at the RF port The isolation between the RF and LO ports however is dependent on the matching of the diodes which may be difficult to maintain over a reasonable frequency range
Image Reject Mixer We have already discussed the fact that two distinct RF input signals at frequencies ampIRF ==
ampgtLO ^IF will down-convert to the same IF frequency when mixed with iMLo These two frequencies are the upper and lower sidebands of a double-sideband signal The desired response can be arbitrarily selected as either the LSB (ampJLO ^w) or the USB ((ULO + 展F) assuming a positive IF frequency The image reject mixer shown in Figure 718 can be used to isolate these two responses into separate output signals The same circuit can also be used for up-conversion in which case it is usually called a single-sideband modulator In this case the IF input signal is delivered to either the LSB or the USB port of the IF hybrid and the associated single sideband signal is produced at the RF port of the mixer
We can analyze the image reject mixer using the small-signal approximation Let the RF input signal be expressed as
FIGURE 718 Circuit for an image reject mixer
where Vu and VL represent the amplitudes of the upper and lower sidebands respectively Using the S-matrix given in (757) for the 90hybrid gives the RF voltages at the diodes as
After mixing with the LO signal given in (756) and low-pass filtering the IF inputs to the IF hybrid
are where K is the mixer constant for the squared term of the diode response The phasor representation of the IF signals of (765) is Combining these voltages in the IF hybrid gives the following outputs which we see are the separate sidebands of the downconverted input signal of (763) These outputs can be expressed in time-domain form as
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-22
which clearly shows the presence of a 90phase shift between the two sidebands Aso note that the
image rejection mixer does not incur any additional losses beyond the usual conversion losses of the single rOection mixer
A practical difficulty with image rejection mixers is in fabricating a good hybrid at the relatively low IF frequency Losses and hence noise figure are also usu3ly greater than for a simpler mixer
REFERENCES [1] D M Pozar Microwave Engineering 2nd edition Wiley New York 1998 [2] S Y Yngvesson Microwave Semiconductor Devices Kluwer Academic Publishers 1991 [3] K Chang Handbook of Microwave and Optical Components vol 2 Chapter 2 Mixers and
Detectors by E L Kollberg Wiley InterScience New York 1990 [4] C T Torrey and C A Whitmer Crystal Rectifiers MIT Radiation Laboratory Series vol 14
McGraw-Hill New York 1948 [5] S A Maas Microwave Mixers 2nd edition Artech House Dedham MA 1993 [6] R A Pucel D Masse and R Bera Performance of GaAs MESFET Mixers at X Band IEEE
TramMicrowave Theory and Techniques vol MTT-24 pp 351-360 June 1976
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-23
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-24
33 57GHz CMOS雙端平衡式混波器設計與製作
為了減少共模雜訊以及clock feed-through透過基板寄生耦合電容對電路的干擾且增
加LO-IFLO-RF隔離度抑制RF和LO訊號的偶次項諧波故選擇double-balanced mixer為主要架構(圖310)Double-balanced mixer 架構較single-balanced mixer架構複雜在於混波
器的輸入端需接上一個單端轉雙端的balun增加電路面積與功率消耗由於使用在外差
式系統中LO頻率(5465~5525GHz)遠高於IF(280MHz)的情形下LO洩漏至IF輸出端之
訊號可以輕易被中頻濾波器所衰減更提高了LO-IF隔離度
VLO-
RLRL
VDD
VLO-VLO+
M3 M4 M5M6
M7 M8
VIF+VIF-
VRF-M2VRF+ M1
圖310 double-balanced mixer電路架構圖
在設計double-balanced mixer時首先決定轉導電晶體大小及偏壓M1M2為轉導電晶體
負責將RF電壓訊號轉為電流訊號因此其轉導大小為首要考量轉導要大偏壓電流勢必要
增大需要較大的功率消耗然而較大的直流偏壓卻可獲得較佳的雜訊表現所以首先考量
轉導電晶體M1M2在偏壓電流各為1mA的條件下參考[8]找出最佳雜訊表現之電晶體寬
度決定轉導電晶體M1M2大小為各為20μm018μm 本地振盪信號VLO+VLO-為大訊號輸入M3 M4M5M6閘極端形成開關電晶體負責
切換RF電流訊號達到混波目的開關電晶體大小決定了開關特性通常電晶體寬度越大開
關特性越佳但是其source端較大的雜散電容容易使RF電流訊號衰減因此需審慎的選擇
M3~M6的大小經過模擬微調後選擇各為40μm018μm已知訊號產生器輸出阻抗為50歐
姆為了達到最大的功率傳輸利用on chip電阻與電容再加上bondwire等效電感將開關電
晶體閘極輸出阻抗匹配至50歐姆由於本地振盪信號為雙端差模輸入故利用一rat race ring將原本單端本地振盪訊號分為雙端差動訊號
混波器負載部分利用RL電阻接成diode connected PMOS型式負載大小由電阻RL與
PMOS汲極與遠源級間的阻抗所構成M7M8電晶體寬度大小取決於負載壓降與負載阻抗
在相同偏壓電流情況下為了達到最小壓降可增大M7M8電晶體寬度此時電晶體較容易
操作在低飽和區易造成線性度因非線性負載而下降增加M7M8電晶體寬度的結果也增加
了電晶體本身之寄身效應導致整體負載阻抗因寄生效應而變小造成轉換增益下降取決
於線性度與轉換增益之考量下選擇M7M8電晶體寬度為160μm018μm考量量測時所
使用的量測儀器皆是50歐姆輸入阻抗之負載為了避免嚴重之負載效應在混波器核心負載
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-25
輸出端與50歐姆負載端間置入一緩衝放大器此緩衝放大器由一共源極電晶體所組成因此
對混波訊號也有放大之功能 Double-balanced mixer 之電路與電路佈局圖如圖 311 所示輸出為雙端輸出因
為 mixer 後級接單端中頻濾波器因此加上 21 的 balun 作為雙端轉單端電路[10]IF端 balun 使用 TOKO 616PT-1039insertion loss 為 3dBBalun 單端輸出為 480MHz 中頻訊號利用 off-chip 的晶片電感 56nH並聯電容 7pF 匹配至 50 歐姆
M1ampM2 gate width = 20um018um
pad equivalent circuit
bondwire equivalent circuit
2nH
0065pF 625O
065O
M3ampM4ampM5ampM6 gate width = 40um018um
M7ampM8 gate width = 160um018umM9ampM10 gate width = 15um018um
IF_OUT
11V
07V
LO+LO+LO -
MLI
N
IF -
MLIN
IF +
18V
18V
08V
M7
45K45K
M8
M6M5M4M3
M1 M2
M10 M9
IF +
IF -
56nH
7pF
圖311 57GHz CMOS double-balanced mixer電路及晶片佈局圖
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-26
34 模擬與量測結果 Double-balanced mixer設計電路的RF輸入範圍為5725~5825GHz中頻輸出為480MHz頻寬
20MHzmixer核心部分直流偏壓為18V18mA緩衝放大器部分為各18V14mA在IF balun 插入損耗為3dB代入轉換損代入模擬結果轉換增益約1276dB input P1dB約-15dBmIIP3約-69dBmLO-RF的隔離度皆在42dB左右LO-IF的隔離度約100dB
量測上利用FR-4製作測試基板來量測晶片mixer核心部分直流偏壓量測為18V2mA緩
衝放大器部分為各18V1mARF輸入訊號端與本地振盪埠(LO)的輸入訊號由訊號產生器
經由rat race ring產生差動訊號饋入混波器量測特性結果為轉換增益約1106dBinput P1dB約-164dBmIIP3約-75dBm LO-RF的隔離度皆在19dB左右LO-IF的隔離度約
50dB模擬與量測結果比較如圖312表31所示晶片照片圖測試板佈局圖及測試板
照片如圖313所示 35 結果與討論
量測結果除了LO-RF隔離度外在輸入輸出匹配轉換增益P1dBOIP3與模擬大
致吻合而在LO-IF隔離度方面設計之時因為balun沒有model可以模擬因此利用一個
理想balun加上3dB的insertion loss代入模擬軟體由於量測使用之IF balun的高頻響應對LO有額外的衰減因此得到比模擬更佳的LO-IF隔離度LO-RF隔離度方面模擬量測差距較
大理想上single-balanced的架構LO-RF隔離度應該是相當大原因可能是因為製程偏移
使得M2及M3電晶體些微不對稱或是substrate coupling造成LO洩漏至RF端改進之處為
佈局方面可以再將M2及M3電晶體對稱性提高及將substrate coupling問題列入佈局考量
55 555 56 565 57 575 58 585 59 595 6
RF Frequency (GHz)
-35
-30
-25
-20
-15
-10
-5
0
S 11(
dB)
57GHz double-balanced mixermeasurementsimulation
300 350 400 450 500 550 600 650 700
IF Frequency (MHz)
-35
-30
-25
-20
-15
-10
-5
0
S 22(
dB)
57GHz double-balanced mixermeasurementsimulation
(a) RF input return loss (b) IF output return loss
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
15
17
19
21
23
25
LO-R
F is
olat
ion
(dB
)
57GHz double-balanced mixerLO-RF isolation
5245 5265 5285 5305 5325 5345 5365
LO Frequency (MHz)
48
49
50
51
52
LO-IF
isol
atio
n(dB
)
57GHz double-balanced mixerLO-IF isolation
(c) LO-RF isolation (d) LO-IF isolation
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-27
-50 -45 -40 -35 -30 -25 -20 -15 -10RF Input Power (dBm)
6
8
10
12
14
16
Con
vers
ion
Gai
n (d
B)
57GHz double balanced mixermeasurementsimulation
470 475 480 485 490
IF Channel Frequency (MHz)
10
11
12
13
14
15
Noi
se F
igur
e(dB
)
57GHz double-balanced mixermeasurementsimulation
(e)轉換增益及input P1dB (f)雜訊指數
479 481477 483
-200
-150
-100
-50
-250
0
freq MHz
dBm
(OU
T)
m1
m7
(g) Two tone test OIP3模擬 (h) Two tone test OIP3量測
圖312 57GHz CMOS double-balanced mixer模擬量測結果
(a) (b) (c)
圖313 57GHz CMOS double-balanced mixer (a)晶片照片圖 (b)測試板照片圖 (c)測試板佈局圖
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-28
表31 57GHz CMOS double-balanced mixer模擬量測特性表
57GHz CMOS Double-Balanced Mixer (TSMC 018μm) RF Frequency Range 5725~5825GHz
IF Frequency 480MHz LO Frequency 5265~5325GHz
Vdd 18V Simulation Measurement
LO Power -3dBm -3dBm Core Each Buffer Current 1814mA 21mA
Conversion Gain 1276dB 1106dB RF Input Return Loss gt21dB gt18dB Output Return Loss 15dB280MHz 20dB280MHz
LO-RF Isolation(LO=-3dBm) 42dB 19dB LO-IF Isolation(LO=-3dBm) gt100dB gt50dB RF-IF Isolation(RF=-30dBm) gt200dB -
Noise Figure 124dB 128dB IIP3 -69dBmRF=-28dBm -75dBmRF=-28dB
Input P1dB -15dBm -164dBm Die size 0627 x 0649 mm2
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-29
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
DesiredChannel
Band SelectFilter Response
BPF1
1LOω
Channel SelectFilter Response
BPF3
2LOω
Channel SelectFilter Response
BPF4
IF Amp
1BPF
Band SelectFilter
Band SelectFilter Response
BPF2
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-30
A
Duplexer Image-RejectFilter
IFFilter
Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6
dB12
dB5dB15
4
4
4
=
==
NF
AA
p
v
dB12dB15
2
2==
NFAv
dB21 =LdB63 =L
A BLNA
C D E F G
IF Amplifier
dB55 =LdB106 =NFLO
Figure 635 Calculation of noise figure in a cascade of stages
Duplexer LNA Image-RejecFilter
Mixer IF Filter
IF Amplifier
Stage Gain (dB) Voltage Power
-2 -2
15 15
-6 -6
15 5
-5
Cumulative Voltage Gain (dB)
Stage NF (dB) 2 2 6 12 5 10
Cumulative NF (dB)
Stage IP3 +100 dBm 12 dBm +100 dBm +5 dBm 1000 Vrms 700mVrms
Cumulative IP3 -106 dBm -126 dBm +11 dBm +5 dBm 221Vrms 700mVrms
Figure 636 Level diagranm corresponding to the cascade of Fig 635
B C D E F
-2 13 7 22 17
879 679 201 141 15 10
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-31
LO
B
NF=10dB
IFA
DDV
1R Ω500dB5dB15
==
p
vAA
SSB NF =10 dB
Figure 632 Cascade of a mixer and an IF amplifier
YXsR
Loω
+inV
Spectrum at X
Spectrum at YLoω
IFω
ω
ω
ThermalNoise
SignalBand Image
Band
Figure 617 Folding of RF and image noise into the IF band
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-32
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
(Tx ) ( Rx ) For a transceiver if transnitting and receiving use different frequencies (for example
celluLOr phone system) a duplexer is used to Separate T Rx xamp duplexer is a kind of BPF Hence a duplexer will also act as a BPF ( before RF stage ) for image rejection
( )( )
V t V t
V t V tRF r r
Lo o o
=
=
⎧⎨⎪
⎩⎪
cos
cos
ω
ω
V V VRF Lo= +α α1 2 darr darr depending on combiner Where to add DC bias lumped circuit form impedance matching is implemmted in the inductor-coie winding
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-33
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
RF IF
LO
RF IF
LO
KTB Frequency
Am
plitu
de a
tm
ixer
RF
port
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
(e) Dual-Gate MESFET Mixer MESFET V V I mA G dBV V I V j k
DS DS
G S G S
= = == minus = minus minus
3 65 2 5 80 78 0 78 980 11 2
output impedance = Ω Ω
RF amp LO signal are combined in dual-gate FET structure MODEL RR-12 LO 1 to 2 GHz RF 1 to 2 GHz IF 0 to 064 GHz Characteristic Min Typ Max Test ConditionConcersion
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-34
Loss dB 5 6 Phase Deviation From 90o
Deg
plusmn4
Amplitude Unbalance dB
plusmn05
All measure- ments made
lsolation dB LO-RF LO-IF RF-IF
18 20 20
25 25 25
in 50 Ohm system RF=1-2GHz LO=1-2GHz
VSWR RF LO IF
15 15 15
20 20 20
IF=003GHz PLO = +7 dBm PRF =minus10 dBm
1dB Compression dBm
+5
3-rd order HP dBm
+15
ltConversion Loss of Mixergt In mixer design several frequencies (RFLOIF) and Their harmonics are involved Impedance matching design at three ports (RFLOIF) is complicated by the above
situation Undesired harmonic signal can be dissipated in resistive termination rarr increase mixer loss or blocked with reactive terminetion rarr frequency dependent An important figure of merit of the mixer Conversion loss Lc = 10 log avaialable RF input power
IF output power
typical Lc= 5 ~ 8 dB for passire mixersa active mixer can have a gain High local oscillator signal (pump) power can rgeduce the mixer conversion loss
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-35
minimum Lcusually sccurs for 0 ~ 10 dBm LO powers But (0 ~ 10 dBm) LO power will violate the small signal approximation of the mixer
diode precise design requires numerical ssolution of the nonlinear equation theat describes
the diode charactoristics
( )( ) ( ) ( )
let
ω ω ω ω ω ω ω
ω ω ω ω
r o i r o i i
RF r r r o r o r o i
IF
V V t V V t V V t
= + minus =
= rarr minus prime = prime
rarr
cos sin sin2 2
( )
( ) ( ) ( )let
prime = minus minus = minus
= prime prime rarr prime prime minus = minus prime
rarrω ω ω ω ω ω
ω ω ω ω
r o i r o i
RF r r r o r o r o iV V t V V t V V tcos sin sin2 2
f f ff f f
fr LO IF
im LO IFIF
= minus
= +⎧⎨⎩
⎫⎬⎭
all produce a same IF frequency and cause image interference
Fig 12 Measured characteristics of a 24 GHz bandpass filter (FDK 2450B)
( ) ( )
( )[ ] ( )[ ]V t t
V t t
rA V
o iV
o i
rB V
o io V
o io
V t VV t V
U L
U L
rA
o iA
rB
o iB
= + + minus
= + minus + minus minus
⎧
⎨⎪
⎩⎪
rarrrarr
2 2
2 290 90
cos cos
cos cos
coscos
ω ω ω ω
ω ω ω ω
ωω
IF mixing
( )[ ]
( ) ( )
V V t V t V t V t
kV t
V kV t kV t
V kV t kV t
kU i L i U i
oL i
L i
iA
U i L i
iB
U io
L io
1 2180
2
90 90
= minus + minus minus
= minus
= minus
= minus minus +
⎧⎨⎪
⎩⎪
sin sin sin sin
sin
sin sin
sin sin
ω ω ω ω
ω
ω ω
ω ω
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-36
Mixer Performance Characteristics
bull Mixer Conversion Loss [ ]Lc =10 log available RF input powerIF output power dB
=gt stringly depend on LO power level
bull Mixer Noise Figure
混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及 贅餘(spurious)頻帶之雜訊混入中頻
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
Loωω
ω
ThermalNoise
SignalBand Image
Band
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同 bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-37
KTB
f + fLO IF
f - fLO IF
f LO
fRF
Frequency
Imagenoise
DesiredSignal
Am
plitu
de a
tm
ixer
RF
port
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-38
APPENDIX 3 SOME APPLICATIONS OF MICROSTRIP CIRCUITS Fooks Microwave Engineering using Microstrip Circuits (Ch 12) This part introduces selected practical microstrip circuits or subsystems It brings together
some of the combination of microwave passive components amp active devices to produce functioning self-contained building blocks that in turn may be a part of a complete microwave systems
FIGURE 1116 Frequency conversion in a receiver and transmitter (a) Down-conversion in a heterodyne (b) Up-conversion in a transmitter f f fIF RF Lo= minus and a much higher frequency signal f fRF Lo+ (filtered out)
2-stage mixers rarr for vetter image frequency rejetion
Portable communication Receiver BLOck Diagram ltMixer in Rransmittergt ( ) ( ) ( ) ( )
( )( ) ( ) ( ) ( )
( )
v t V t V V t
V t V V t
v t V t V V t
V t V V t
r ro
o n oo
r r o n o
r ro
o n oo
r r o n o
1
2
90 180
180 90
= minus + + minus
= + +
= minus + + minus
= minus + +
⎫
⎬⎪⎪
⎭⎪⎪
cos cos
sin cos
cos cos
cos sin
ω ω
ω ω
ω ω
ω ω
conside only quadratic term of the diode which givers thedesired mixer product
混頻器的雜訊指數根據IEEE的定義單旁波帶(SSBsingle side band)雜訊指數與雙埠放大器之定義相同如(24)式但在實際測量上較為困難若以濾波器阻隔假像頻率(image
grequency)頻帶之雜訊則濾波器將造成輸入阻抗改變且增加溫度雜訊因此在測量上多採用雙旁波帶(DSBdouble side band)雜訊指數較方便一般的定義雙旁波帶雜訊指數為單旁波帶雜訊指數的兩倍圖25說明混頻器雜訊之造成混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時也將假像(image)頻帶及贅餘(spurious)頻帶之雜訊混入中頻因此混頻器的雜訊指數將比放大器來得大圖26為測量混頻器雜訊指數之架構所得為雙旁波帶雜訊指數
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-39
fIF2
=gt channel bandwidth
IF smallrarr amp requiring higher Q of the IF filter
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-40
Pt
Gt Gr
Pr
R
fRF
t
RF DC
f
t
f
t
f
t
f
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
fRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector
(c) Mixer (frequency conversion)
RF IF
LO
f
f
ffRF
fLO
f fRF LOminus f fRF LO+
For a single-ended mixer the noise terms will be ( ) ( )[ ] ( )tVVtVtVV nonno 212
21 =++
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-41
f
A
f
B
f
C
f
D
f
E
f
F
f
G
f
H
DesiredChannel Image
imωrω rω
rω 1IFω
imω
ImageDesiredChannel
imω
Image
DesiredChannel
interfers
1IFω 2IFω
2IFω 2IFω
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-42
Mixer Nonlinear Simulation by Libra
( )
( ) ( )
( ) ddso
VsVdVV
jjdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
RRGIIeIdVdI
eIVI
oooo
oo
prime=α=+α=
α===rArr
==+α=α=rArr
minus=
α
minusα
α
2
222
1
)1()(
resistance junction
Since
( )
( ) ( )
( )
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
prime=α=+α=
α===
==+α=α=
minus= α
minusα
α
resistance junction
Since
ddso
VsVdVV
j
jdsoV
sV
Vs
GGII
eIdVdGdVdVdIddVId
R
RGIIeIdVdI
eIVI oooo
oo
2
222
1
)1()(
Mixer
Basebandfilter
fMAntenna
Localoscillator
+ fMfLo
Poweramplifier
LOf Up-conversion(for transmitting)
DetectorIFamplifier
LNA
+ fMfIF fIFfMfLo +
fMfLo-fo =
Mixer
Localoscillator
IF filter
Down-conversion (receiving)
bull The band-stop response of the BPF will determine the image-rejection ratio
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-43
bull
1BPF 2BPF 3BPF 4BPF
2LOω1LOω
A B C D E F G H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifierLNA
1stMixer
2ndMixer
ImageRejectFilter
t0cosω
LNAChannelSelectFilter
LO
MixerRF IF
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
o90
( )[ ] ttVVV onoLo ω+= cos
tVV rrRF ω= cos
1v
2v
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-44
3-dBpowerdivider
0Z
rvRF input
Arv
Brv
RF
RF
Mixer B
Mixer A
LO LOinput
oω
hybrido90
hybrido90
1v
2v
LSB
USB
IFout
⎩⎨⎧
ω+ω+ω+ω
=tVtV
vioL
ioUr )(cos
)cos(
LO
IF
IF
( ) ( ) ( )( ) ( ) ( )
( )
( )⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=+minus=
+=ΓminusΓ=+=
ΓΓΓ
ΓΓ
ΓΓΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
Lo
LorLorLoRF
LoRFRFRFRF
VV
VjVVV
VV
Vj
VVjjVjVVVj
VVVjVVVV
21
22
1121
21
21
21
21
21
2
222
( ) ( ) ( )
( ) ( )[ ]( ) ( )[ ]
( )
( )⎪⎪⎪⎪⎪⎪
⎩
⎪⎪⎪⎪⎪⎪
⎨
⎧
Γminus=+=
Γminus=
+ΓminusminusΓminusΓ=
+ΓminusminusΓminusΓ=
ΓminusΓ=minusΓ+Γ=+=
ΓΓΓ
ΓΓΓ
port LO at appears signal but reflection no
port RF at appears signal but reflection no
rLo
rLOLOLO
LOr
LO
LOrLOr
LOrLOr
RFRFRF
VV
VjVVV
VV
Vj
VVjjVjV
VVjjVjV
VjVjVVVVV
21
21
21
21
21
21
2121
21 22
monolithic quad DMOS FET for mixer application
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-45
FET mixer has a gain (Diode mixer has no gain) TABLE 111 Frequencies and Relative Amplitudes of the Square-Law Output of a
Detected AM signal Frequency Relative Amplitude 0 1 22+ m
ωω
mm2 2
22m
m rarrdesired demodulated output
2ωo 1 22+ m 2ω ωo mplusmn m
( )2 ω ωo mplusmn m2 4
EXAMPLE 14 A diode in as axial-lead package has the following equivalent circuit parameters Cp =010 pF Lp=20nH Cj=015pF Rs=10Ω and I s=01μA Calculate and plot the
impedance of this diode from 4 to 14 GHz for a bias current Io=0 and Io=60μA Ignore the change in Cj with bias and assume α =125 mV
solution From (1127) the junction resistance for the two bias states is for ( )I Ro j I I
mVAo s
= = = = times+
0 22 5 101 2501
5 α μ Ω
for ( ) ( )I A Ro j I ImV
Ao s= = = =
+ +60 4171 25
60 01μα μ
Ω
Then the input impedance can be calculated from the equivalent circuit of Figure 1112 the result is plotted versus frequency on a 50Ω Smith chart in Figure 1113
Diode impedance is frequency dependent FIGURE 1113 Impedance of the diode of Example 114 for to bias states from 4 to
14 GHz (1127) ( )R Gj d I Io s
= =minus+
1 1α
( )V t 2
( )( )
( )14
222
2
12
2
V Gd m t to mV Gd m
V
mo
o
prime = primecos cosω ω
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-46
bull FM broadcasting IF = 107MHz bull Cellular phone IF = 45MHz IF = 455MHz
RF
trr ωcosMatchingnetwork
Combiner
LO
tv oo ωcos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Diode2 LP
filters
+ If output
Diode1 1i
2i
RF input
LO input3 dB hybrid
)18090( ooor
Fig
1019
1BPF 2BPF 3BPF 4BPF
2LOω1LOωA B C D E F G
H
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter
IFAmplifier
1BPF 2BPF 3BPF 4BPF
A B C D E F GH
BandSelectFilter
ImagerejectFilter
ChannelSelectFilter
ChannelSelectFilter IF
Amplifier
f
DesiredChannel
Image
imωrω
interfersinterfers
1LOω 2LOω
2IFω
DesiredChannel
Fig 6 Measured characteristics of a 24-GHz single-ended resistive FET mixer
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-47
Mixer Noise Figure 混頻器除了射頻(RF)頻帶之雜訊混至中頻(IF)帶外同時有image頻帶及
贅餘(spurious)頻帶之雜訊混入中頻iexcl
fIF
fLO
fRF
Mixer noise
2 fLO
- fIF
2 fLO
+ fIF
bull 單旁波帶(SSB single side band)雜訊指數與雙埠放大器之定義相同
bull 測量上多採用雙旁波帶(DSBdouble side band) 雜訊指數較方便iexcl
=gt 單旁波帶雜訊指數為單旁波帶雜訊指數的兩倍
Mixer
LOSignal Generator
Broad bandNoise Source
Pre Amp
Noise Figure Meter
Mixer DSB noise figure measurement
混頻器為雙旁波帶雜訊指數測量
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-48
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-49
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ
tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
LP filter
+ If outputDiode 1
1i
2i
RF input
LO input
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cos
RFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓtVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-50
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-51
RF
tvV rrRF ω= cos
Matchingnetwork
Combiner
tvV ooLO ω= cos
DCreturn
or
orωplusmnω
ωωLP filter
DCbias
tv ori )cos( ωminusω
Single-ended mixer circuit
LO
IF
tVtVVVv oorrLoRF ω+ω=+= coscos For a diode small-signal approximation
( ) ( )
( ) ( )
( )( )[ ] tVVtVV
tVttVVtV
tVVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
=ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
coscoscos2cos
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-52
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-53
(a) Single-ended Mixer
Pozar (RF Ch 7)
Mixer in a Transmitter
bull In a transmitter a mixer is used to mix with IF signal to up-convert the signal frequency for efficient radio-wave transmission from antenna
MixerBaseband filterfM
Antenna
Local oscillator
+ fMfLo
Poweramplifier
LOfUp-conversion(for transmitting)
t
( )( )⎪⎩
⎪⎨⎧
minus
+
LSB SidebandLower
USB SidebandUpper
IFLo
IFLo
ff
ff Double sideband (DSB)= USB + LSB
=gt For a single sideband transmission (SSB)
( )USB f fLo IF+ or ( )LSB f fLo IFminus
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-54
We use a sideband filter or an image rejection mixer to remove a sideband signal ( also celled single sideband modulator )
direct-conversion transmitter
tcωcos
BasebandQ
BasebandI
tcωsin
MatchingNetwork
Duplexer
PA
drawback leakage of PA output to LO
LO
I
Q
BPF
LOω ω
PA
56 565 57 575 58 585 59Frequency (GHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S11
(dB
)
measurementsimultaioon
400 420 440 460 480 500 520 540 560
Frequency (MHz)
-40
-35
-30
-25
-20
-15
-10
-5
0
S22
(dB
)
measurementsimultaioon
(a) (b)
-40 -35 -30 -25 -20 -15 -10 -5 0input power (dBm)
0
2
4
6
8
10
12
14
Con
vers
ion
Gai
n (d
B)
measuemsntsimulation
(c) (d)
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-55
(中正大學電機MS Thesis)
f - fLO IFf + fLO IF
3f - fLO IF
f LO
2fLO3fLO
2f - fLO IF2f + fLO IF 3f + fLO IF
Spurious chart due to LO harmonics
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-56
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-57
image rejection filter The BPF band-stop response
determines the image-rejection ratio
RF IF
LOf f
fRF f fRF LOminus f fRF LO+
(c) Mixer (frequency conversion)
fLOf
DetectorIFamplifierLNA
fIF
Mixer
Local oscillator
IF filter
MLORF fff plusmn= Mf
)( IFLO
off
fminus=
MIF
MoLO
oRF
fffff
ff
plusmn=plusmnminus=
minust
t
Down-conversion (for receiving)
ImageRejectFilter
LNA
tA LOωcos0
IFω2ωrω imω
Image RejectFilter Response
DesiredBand
image
ωrω
DesiredBand
image
A 24 GHz bandpass filter =gtpassband = 100 MHz =gt insertion loss lt 1 dB
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-58
bull
fRF
t
RF DC
f
t
f
t
f
t
ffRF fm
ModulatedRF
Modulation
(a) Diode rectifier
(b) Diode detector Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
⎪⎪⎪⎪
⎩
⎪⎪⎪⎪
⎨
⎧
ωminusωminusω+ωminus=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
tt
ttV
tt
ttV
IFoV
IFoV
oIFo
VoIFo
VBr
IFoV
IFoV
oIFo
VoIFo
VAr
LU
LU
LU
LU
)cos()cos(
]180)cos[(]180)cos[(
)sin()sin(
]90)cos[(]90)cos[(
22
22
22
22
bull After mixing with an LO signal of cos ωot the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ωminusminusω=
ωminusω=
)90sin()90sin(
sinsino
IFLo
IFUB
i
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o hybrid at the IF output gives
component LSB tkV
tVtVtVtV
tkVtkV
tkVtkVV
IFL
IFLo
IFUIFLIFUk
ooIFL
ooIFU
IFLIFU
ωminus=
ωminusminusω+ωminusω=
⎥⎥⎦
⎤
⎢⎢⎣
⎡
minus+ωminusminusminusω
+ωminusω=
sin2
]sin)180sin(sinsin[
)]9090sin()9090sin([
)sinsin(
2
21
1
Filter 1 Filter 2 Detector
Injectionfilter
~ ~
st1 mixer mixer2nd
RFamplifier
1 IFstages
st 2 IFstages
nd
2 localoscillator
nd1 localoscillator
st
1 IFamplifier
st
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-59
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr ⎪⎩
⎪⎨
⎧
minusωminusω+minusω+ω=
ωminusω+ω+ω=
]90)cos[(]90)cos[(
)cos()cos(
22
22o
IFoVo
IFoVB
r
IFoV
IFoVA
r
ttV
ttV
LU
LU
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are
IF outputs rarr ⎪⎩
⎪⎨⎧
+ω+minusω=
ω+ω=
)]90cos()90cos([
]coscos[
221
221
oIFL
oIFU
Bi
IFLIFUA
i
tkVtkVV
tkVtkVV
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
component LSB
]
tV
tVtVtVtV
tkVtkV
tkVtkV
V
IFLk
IFLIFLo
IFUIFUk
ooIFL
ooIFU
IFLIFU
ω=
ω+ω+minusω+ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minus+ω+minusminusω
+
ω+ω
=
=
cos
]coscos)180cos(cos[
)]9090cos()9090cos([
coscos[
2
04
221
221
21
1
component USB tV
tVtV
tVtV
V
IFUk
oIFL
oIFU
oIFL
oIFU
k
ω=
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
minusω+minusω
+
+ω+minusω
=
sin
)]90cos()90cos([
)]90cos()90cos([
2
221
221
22
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-60
bull Then the input to the two mixers through a 90o hybrid is
RF inputs rarr
[ ][ ]
[ ][ ]⎪
⎪⎪
⎩
⎪⎪⎪
⎨
⎧
ωminusω+ω+ω=
minusωminusω+minusω+ω=
ωminusω+ω+ω=
minusωminusω+minusω+ω=
minus tVtV
ttVttVv
tVtV
ttVttVv
IFoLIFoU
oIFoL
oIFoU
BRF
IFoLIFoU
oIFoL
oIFoU
ARF
)cos()cos(
)180cos()180cos(
)sin()sin(
)90cos()90cos(
212
12
12
1
bull After mixing with an LO signal of cos ωot amp lowpass filtered the IF outputs of the mixers are (K = the mixer constant for the squared term of the diode)
IF inputs (to IF hybrid)rarr ⎪⎩
⎪⎨⎧
ω+=
ωminus=
minus tVVVv
tVVVv
IFLULOKB
IF
IFLULOKA
IF
cos][
sin][
22
22
Phasor representation ( )
( )⎪⎪⎩
⎪⎪⎨
⎧
+minus
=
minusminus
=
LULOB
IF
LULOA
IF
VVVKV
VVVjKV
22
22
bull Combining these two signals in the 90o IF hybrid (transformer) at the IF output gives
222
222
2
1
ULOB
IFA
IF
LLOB
IFA
IF
VjKVVjVV
VKVVVjV
=minusminus=
=minusminus=
sin2
)(
cos2
)(
2
1
tVKVtv
tVKVtv
IFULO
IFLLO
ωminus
=
ω=
3-dBpowerdivider
0Z
rvRF input
ARFv
BRFv
RF
RFMixer B
Mixer A
LO LOinput
)cos( tVv
oLO
LOω
=
hybrido90
IF hybrid(transformer)
o90
1v
2v
LSB
USB
IFout
⎪⎩
⎪⎨
⎧
ωminusω+
ω+ω=+
tV
tVvv
IFoL
IFoU
RFIM)(cos
)cos()
LO
IF
IFLPF
LPF
AIFv
BIFv
IFω2rω imω
Desired Bandimage
rω
Desired Band
imω
image
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=
2008 copyH-R Chuang EE NCKU
7-61
( ) ( )
( ) ( )
( )[ ] tVVtVV
tVttVVtV
tVtVGv
tVtVVVvGvvGIVI
IForG
ororG
ooororrG
oorrG
d
oorrLoRFddo
dd
rd
d
ω=ωminusω=rArr
ω+ωω+ω=
ω+ω=primerArr
ω+ω=+=+prime++=
primeprime
prime
prime
coscos2
)coscoscos2cos(
coscos2
)coscos2
24
22222
22
2
2
output IF
(note
poorisolation
LP filter
+ IF outputDiode 1
1i
2i
RF
LO
3 dB hybrid)90( o
Diode 2
LP filtertVV rrRF ω= cos
tVV ooLo ω= cosRFVΓ
LOVΓ
)2()2(1 Lor VjVv minus+=
1vΓ
2vΓ tVKVi iorIF ωminus= sin2
)2()2(2 Lor VVjv +minus=