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UNIVERS ITY OF OULU P.O.B . 7500 F I -90014 UNIVERS ITY OF OULU F INLAND
A C T A U N I V E R S I T A T I S O U L U E N S I S
S E R I E S E D I T O R S
SCIENTIAE RERUM NATURALIUM
HUMANIORA
TECHNICA
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SCIENTIAE RERUM SOCIALIUM
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OECONOMICA
EDITOR IN CHIEF
PUBLICATIONS EDITOR
Professor Mikko Siponen
University Lecturer Elise Kärkkäinen
Professor Hannu Heusala
Professor Olli Vuolteenaho
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ISBN 978-951-42-9086-2 (Paperback)ISBN 978-951-42-9087-9 (PDF)ISSN 0355-3213 (Print)ISSN 1796-2226 (Online)
U N I V E R S I TAT I S O U L U E N S I SACTAC
TECHNICA
U N I V E R S I TAT I S O U L U E N S I SACTAC
TECHNICA
OULU 2009
C 320
Mikko Komulainen
BANDWIDTH ENHANCED ANTENNAS FOR MOBILE TERMINALS AND MULTILAYER CERAMIC PACKAGES
FACULTY OF TECHNOLOGY,DEPARTMENT OF ELECTRICAL AND INFORMATION ENGINEERING,UNIVERSITY OF OULU;INFOTECH OULU,UNIVERSITY OF OULU
C 320
ACTA
Mikko K
omulainen
C320.fm Page 1 Tuesday, March 24, 2009 12:26 PM
A C T A U N I V E R S I T A T I S O U L U E N S I SC Te c h n i c a 3 2 0
MIKKO KOMULAINEN
BANDWIDTH ENHANCED ANTENNAS FOR MOBILE TERMINALS AND MULTILAYER CERAMIC PACKAGES
Academic dissertation to be presented with the assent ofthe Faculty of Technology of the University of Oulu forpublic defence in Raahensali (Auditorium L10), Linnanmaa,on 23 June 2009, at 12 noon
OULUN YLIOPISTO, OULU 2009
Copyright © 2009Acta Univ. Oul. C 320, 2009
Supervised byProfessor Heli Jantunen
Reviewed byProfessor Matthias HeinDoctor Pekka Salonen
ISBN 978-951-42-9086-2 (Paperback)ISBN 978-951-42-9087-9 (PDF)http://herkules.oulu.fi/isbn9789514290879/ISSN 0355-3213 (Printed)ISSN 1796-2226 (Online)http://herkules.oulu.fi/issn03553213/
Cover designRaimo Ahonen
OULU UNIVERSITY PRESSOULU 2009
Komulainen, Mikko, Bandwidth enhanced antennas for mobile terminals andmultilayer ceramic packagesFaculty of Technology, Department of Electrical and Information Engineering, University ofOulu, P.O.Box 4500, FI-90014 University of Oulu, Finland; Infotech Oulu, University of Oulu,P.O.Box 4500, FI-90014 University of Oulu, Finland Acta Univ. Oul. C 320, 2009Oulu, Finland
Abstract
In this thesis, bandwidth (BW) enhanced antennas for mobile terminals and multilayer ceramicpackages are presented. The thesis is divided into two parts. In the first part, electrically frequency-tunable mobile terminal antennas have been studied. The first three antennas presented were of adual-band planar inverted-F type (PIFA) and were tuned to operate in frequency bands appropriateto the GSM850 (824–894 MHz), GSM900 (880–960 MHz), GSM1800 (1710–1880 MHz),GSM1900 (1850–1990 MHz) and UMTS (1920–2170 MHz) cellular telecommunicationstandards with RF PIN diode switches. The first antenna utilized a frequency-tuning methoddeveloped in this thesis. The method was based on an integration of the tuning circuitry into theantenna. The tuning of the second antenna was based on a switchable parasitic antenna element.By combining the two frequency-tuning approaches, a third PIFA could be switched to operate ineight frequency bands.
The planar monopole antennas researched were varactor-tunable for digital television signalreception (470–702 MHz) and RF PIN diode switchable dual-band antenna for operation at fourcellular bands. The key advantage of the former antenna was a compact size (0.7 cm3), while forthe latter one, a tuning circuit was implemented without using separate DC wiring for controllingthe switch component.
The second part of the thesis is devoted to multilayer ceramic package integrated microwaveantennas. In the beginning, the use of a laser micro-machined embedded air cavity was proposedto enable antenna size to impedance bandwidth (BW) trade-off for a microwave microstrip in amultilayer monolithic ceramic media. It was shown that the BW of a 10 GHz antenna fabricatedon a low temperature co-fired ceramic (LTCC) substrate could be doubled with this technique.Next, the implementation of a compact surface mountable LTCC antenna package operating near10 GHz was described. The package was composed of a BW optimized stacked patch microstripantenna and a wide-band vertical ball grid array (BGA)-via interconnection. Along with theelectrical performance optimization, an accurate circuit model describing the antenna structurewas presented. Finally, the use of low-sintering temperature non-linear dielectric BariumStrontium Titanate (BST) thick films was demonstrated in a folded slot antenna operating at 3GHz and frequency-tuned with an integrated BST varactor.
Keywords: frequency-tunable antennas, LTCC, microstrip antennas, mobile terminals,package integration, planar inverted-F antennas, planar monopole antennas, slotantennas, small antennas
5
Acknowledgements
This work was done at the Microelectronics and Materials Physics Laboratories
of the University of Oulu and EMPART research group of Infotech Oulu during
2005–2008. The work was carried out in the framework of the TAMTAM-project
(future active multiband antennas). The project was funded by the Finnish
Funding Agency for Technology and Innovation (TEKES), Nokia Mobile Phones
Business Group, Pulse Finland Oy, and Elektrobit Microwave Oy (2005–2006).
I wish to thank my supervisor Professor Heli Jantunen who gave me this
great opportunity and the support to perform this research. In addition, Dr. Tero
Kangasvieri, Dr. Charles Free, Professor Erkki Salonen, Markus Berg, Timo Tick,
Vamsi Palukuru and Jyri Jäntti along with all those who participated in the
TAMTAM-project and the staff of the Microelectronics and Materials Physics
Laboratories are acknowledged.
In the year 2008 the work was financially supported by Infotech Oulu
Graduate School. The Nokia Foundation, the Seppo Säynäjäkangas Foundation
and the University of Oulu Support Foundation are acknowledged for supporting
the work twice. The Finnish Foundation of Electrical Engineers, the Tauno
Tönning Foundation, the Riitta and Jorma J. Takanen Foundation, the Jenny and
Antti Wihuri Foundation, and the Ulla Tuominen Foundation are acknowledged
for supporting the work.
6
7
List of abbreviations and symbols
εr Relative permittivity
λ0 Free space wavelength
λg Guided wavelength
σ Conductivity of tissue
р Density of tissue
ηrad Radiation efficiency
ηref Reflection efficiency
ηtot Total efficiency
ωr Angular resonant frequency
Ω Ohm
~ Approximately
A Ampere
BGA Ball Grid Array
BW Bandwidth
BST Barium Strontium Titanate
C Capacitance
°C Celsius degree
cm Centimetre
dB Decibel
dBi Decibel relative to isotropic radiator
DC Direct Current
DVB-H Digital Video Broadcast -Handheld
FSA Folded Slot Antenna
GaAs Gallium Arsenide
GCPW Grounded Co-Planar Waveguide
GPS Global Positioning System
GSM Global System for Mobile communications
E Electric field
EM Electro-Magnetic
EMC Electro-Magnetic Compatibility
fr Resonant frequency
FM Frequency Modulation
HFSS High Frequency Structure Simulator
IC Integrated Circuit
ICNIRP International Commission on Non-Ionizing Radiation Protection
8
IEC International Electrotechnical Commission
IEEE Institute of Electrical and Electronics Engineers
k Wave number
L Inductance
LTCC Low Temperature Co-fired Ceramic
MEMS Micro Electro Mechanical System
MIMO Multiple Input Multiple Output
mm Millimetre
NFC Near Field Communication
P Power
Pdiss Average power dissipation per period.
PCB Printed Circuit Board
PIN Positive Intrinsic Negative
PIFA Planar Inverted-F Antenna
Q Quality factor
QC Conductive quality factor
QD Dielectric quality factor
QE External quality factor
QL Loaded quality factor
QR Radiation quality factor
QR,min Theoretical limit for the smallest radiation quality factor
Q0 Un-loaded quality factor
R Resistance
R0 Input resistance at resonance
r Radius
RF Radio Frequency
RF ID Radio Frequency IDentification
RX Receive
S Bandwidth criteria
S11 Scattering parameter
SAR Specific Absorption Rate
SiP System in Package
SMD Surface Mount Device
SPDT Single Pole Double Throw
T Characteristic coupling co-efficient
Tc Curie temperature
Tan δ Dielectric loss tangent
9
TCT Thermal Cycling Test
TM Transverse Magnetic field
TX Transmit
UMTS Universal Mobile Telecommunications System
UWB Ultra Wide Band
V Volt
Varactor Variable capacitor
VSWR Voltage Standing Wave Ratio
W Total energy stored into a resonator
WLAN Wireless Local Area Network
Z0 Characteristic impedance
X-band Microwave frequency range 8-12 GHz
2G Second Generation
3G Third Generation
10
11
List of original papers
The thesis is based on the following eight papers, which are cited in the text by
their Roman numerals.
I Komulainen M, Berg M, Jantunen H, Salonen E & Free C (2008) A Frequency Tuning Method for a Planar Inverted-F Antenna. IEEE Transactions on Antennas & Propagation 56(4): 944–950.
II Komulainen M, Palukuru VK & Jantunen H (2007) Frequency-Tunable Dual-Band Planar Inverted-F Antenna Based on a Switchable Parasitic Antenna Element. Frequenz –Journal of RF engineering and telecommunications 61(9–10): 207–212.
III Komulainen M & Jantunen H (2008) Switching a dual-band planar inverted-F antenna to operate in eight frequency bands. Proceedings of Loughborough Antennas and Propagation Conference: 85–88.
IV Komulainen M, Berg M, Jantunen H & Salonen E (2007) Compact varactor-tuned meander line monopole antenna for DVB-H signal reception. IET Electronics Letters 43(24): 1324–1326.
V Komulainen M, Berg M, Palukuru VK, Jantunen H & Salonen E (2007) Frequency-Reconfigurable Dual-Band Monopole Antenna for Mobile Handsets. IEEE Antennas & Propagation International Symposium Digest: 3289–3292.
VI Komulainen M, Mähönen J, Tick T, Berg M, Jantunen H, Henry M, Free C & Salonen E (2007) Embedded air cavity backed microstrip antenna on an LTCC substrate. Journal of European Ceramic Society 27(8–9): 2881–2885.
VII Komulainen M, Kangasvieri T, Jäntti J & Jantunen H (2008) Compact Surface-Mountable LTCC-BGA Antenna Package for X-band Applications. International Journal of Microwave and Optical Technology 3(4): 451–459.
VIII Palukuru V K, Komulainen M, Tick T, Peräntie J & Jantunen H (2008) Low Sintering-Temperature Ferroelectric-Thick films: RF Properties and an Application in a Frequency-Tunable Folded Slot Antenna. IEEE Antennas and Wireless Propagation Letters 7: 461–464.
Ways to enhance the bandwidth of small antennas for mobile terminals and
multilayer ceramic packages are presented. Papers I–V relate to improving the
efficient use of mobile terminal antennas with electrically controlled frequency-
tuning. Papers VI–VIII deal with BW enhanced planar microwave antennas on
multilayer ceramic packages.
Papers I–III are on the topic of electrically controlled frequency-tuning of a
planar inverted-F antenna (PIFA). The antennas investigated are of the dual-band
type and tuned with RF switches. The antennas operate at frequency bands
appropriate to the GSM850 (824–894 MHz), GSM900 (880–960 MHz),
GSM1800 (1710–1880 MHz), GSM1900 (1850–1990 MHz) and UMTS (1920–
12
2170 MHz) telecommunication standards (the cellular bands). In paper I, a
frequency tuning method for PIFAs is presented. The method is based on the
integration of a tuning circuit to the antenna element. The key advantages are that
no separate DC wiring is used in the tuning circuit and the tunable antenna used
for demonstrating the method showed high total antenna efficiency in all of its
four frequency bands. Paper II describes another approach for implementing a
similar antenna. The tuning is based on a switchable parasitic antenna element.
Finally, in paper III, the two frequency-tuning methods are combined into the
same antenna structure, which can be electrically switched to operate in eight
frequency bands.
Papers IV and V describe an implementation of frequency-tunable planar
monopole antennas. A varactor tuned antenna investigated in paper IV is for
digital television signal reception in a small portable device. The RF switch
tunable antenna in paper V operates at four cellular bands. The main advantage of
the former antenna is its compact size, while the tuning circuit of the latter
antenna does not use a separate DC wiring.
In paper VI, a laser micro-machined embedded air cavity is proposed to
enable an antenna size to bandwidth and efficiency trade off in a dielectric media
typical for a Low Temperature Co-fired Ceramic (LTCC) microwave System in
Packages (SiPs). Paper VII describes the implementation of a surface mountable
LTCC ball grid array (BGA) antenna package operating at X-band. The package
consists of a BW optimized LTCC integrated stacked patch microstrip antenna
and wide-band package transitions. Along with the electrical design optimization,
a circuit model for the antenna package was presented and the thermal fatigue
reliability of board level solder joints of the package assemblies was
experimentally verified. Finally, paper VIII describes the dielectric properties of
low sintering temperature Barium Strontium Titanate (BST) thick films at RF
frequencies and demonstrates the use of them as an integrated varactor in a
frequency-tunable folded slot antenna. The low sintering temperature of the
material enables compatibility with well conducting silver electrodes and LTCC
material systems.
In papers I–VI, the idea, design and all the experimental work were mostly
contributed by the author with the assistance of the co-authors. The ceramic
antennas in papers VI–VII were fabricated by the co-authors. All the
measurements were done by the author in part with the kind help of the co-
authors, excluding the radiation pattern measurements in paper VI, which were
done by the co-authors. The idea in paper VII was mostly contributed by the co-
13
author, however the design and modelling of the antenna, as well as the
measurements, are the work of the author. In paper VIII, the author’s contribution
was participation in the idea and the experimental work. All the manuscripts were
written by the author with the kind help of the co-authors.
14
15
Contents
Abstract Acknowledgements 5 List of abbreviations and symbols 7 List of original papers 11 Contents 15 1 Introduction 17
1.1 Fundamental limitations of small antennas............................................. 17 1.2 Package integrated antennas ................................................................... 19 1.3 Mobile terminal antennas........................................................................ 21 1.4 Objectives and outline of the thesis ........................................................ 23
2 Frequency-tunable antennas for mobile terminals 25 2.1 Frequency- tunable dual-band planar inverted-F antennas ..................... 26
2.1.1 A frequency tuning method for a planar inverted-F
antenna ......................................................................................... 27 2.1.2 Use of a switchable parasitic antenna element ............................. 32 2.1.3 Switching a dual-band planar inverted-F antenna to
operate in eight frequency bands.................................................. 37 2.2 Frequency-tunable planar monopole antennas ........................................ 41
2.2.1 Compact varactor-tuned meander line monopole antenna
for DVB-H signal reception ......................................................... 42 2.2.2 Frequency-tunable planar monopole antenna for a quad-
band operation .............................................................................. 46 2.3 Discussions ............................................................................................. 49
3 Ceramic package integrated microwave antennas 53 3.1 Embedded air cavity backed microstrip antenna on an LTCC
substrate .................................................................................................. 54 3.2 Surface-mountable LTCC-BGA antenna package................................... 57 3.3 Integrated BST thick film varactor-tuned folded slot antenna ................ 60 3.4 Discussions ............................................................................................. 64
4 Conclusions 67 References 71 Original papers 81
16
17
1 Introduction
The rapid evolution of information technology and wireless communications has
enabled the development of applications that one was not even able to dream of a
few decades ago. Personal communication has become an integral part of our
everyday lives. Almost everything has gone wireless and mobile today. This is
mainly due to the tireless and everlasting efforts of the electronics and
telecommunications industry to increase the functionality and performance of the
devices while squeezing their size and cost.
Among the most critical components from the view-point of system
integration and miniaturization in wireless devices are antennas. Antennas
represent a category of components that fundamentally differs from others. An
antenna should radiate efficiently in an intended manner to free space, while other
components should be more or less isolated from their surroundings. Along with
the increased functionality of devices, a growing number of wireless
communication standards are used in a device. Devices need to accommodate an
ever-increasing number of antennas, or there would be a need for a significant
bandwidth enhancement for the existing ones. Meanwhile, the reduction in device
size has caused an increasingly higher space constraint in the implementation
environments for antennas. This creates a major challenge for antenna engineers,
since the downsizing of an electrically small antenna has fundamental limitations
and has to be professionally done.
1.1 Fundamental limitations of small antennas
Virtually all small antennas used in wireless communications devices are of the
resonant type. Their size is bound to a fraction of the wavelength at the operating
frequency. The smallest usable fraction of a wave length is one quarter. However,
many advanced antenna designs are slightly smaller. The fundamental theory for
resonant type antennas was derived decades ago (Wheeler 1947, Chu 1948,
Wheeler 1959, Harrington 1960, Wheeler 1975, Hansen 1981). The input
impedance of an antenna that is small compared to the wavelength changes
rapidly as a function of frequency. The antenna can be impedance matched to its
feed only in a narrow frequency band. This frequency band is called the
impedance bandwidth. It is defined as the frequency range where the return loss
(or Voltage Standing Wave Ratio, VSWR) stays below a certain predefined level
e.g. 3 dB (VSWR = 6), 6 dB (VSWR = 3) or 10 dB (VSWR = 1.92). A usable BW
18
can also be limited by other parameters, such as, gain, radiation pattern or
polarization. In most of the cases, these factors are better maintained over a wider
frequency band than the impedance match.
The amount of input power accepted by an antenna is described by the
reflection efficiency (ηref). The radiation efficiency (ηrad) describes how much of
the input power accepted by the antenna is converted to a far field radiated power.
This describes the internal losses of an antenna structure. The total efficiency (ηtot
= ηrefηrad) accounts for both impedance mismatch and internal losses. The
impedance bandwidth and radiation efficiency can be also expressed with quality
factors. The quality factor of a resonator Q = ωrW/Pdiss describes the rate at which
energy decays in the resonator. Here ωr is angular resonant frequency, W is the
total energy stored in the resonator and Pdiss is average power dissipated per
period. The total power loss of an antenna can be divided into separate terms,
each of which has its own quality factor (Eq. 1):
0
1 1 1 1 1 1 1 1.
L E E R C D SWQ Q Q Q Q Q Q Q= + = + + + + (1)
The loaded quality factor (QL) is the total quality factor of the antenna. It consists
of an unloaded quality factor (Q0) and an external quality factor (QE). Q0 accounts
for the internal losses of the antenna and QE describes the losses of the antenna’s
external connections, such as the feed. Internal losses can be further divided into
radiation (QR), conductive (QC), dielectric (QD) and surface wave losses (QSW).
An antenna’s radiation efficiency can be expressed as a ratio of un-loaded quality
factor and radiation quality factor (Eq. 2):
rad
rad Q
Q0=η . (2)
Antennas impedance bandwidth (BW) can be expressed with depend on these
quality factors in the manner shown in Eq. 3:
S
TSTS
QBW
))(1(1
0
−−= . (3)
Here, S denotes the VSWR bandwidth criteria and T is the characteristic coupling
coefficient to a resonant circuit formed by the antenna. T = Z0/R0 for series
resonant circuit and T = R0/Z0 for parallel resonant circuit, where Z0 and R0 are an
19
input resistance at resonance and Z0 a characteristic impedance of a transmission
line feeding the antenna.
It has been shown that relationships originating from fundamental physics
link the size of an antenna, being small compared to operation wavelength, to its
bandwidth and radiation efficiency (Chu 1948, McLean 1996). The theoretical
limit for the smallest radiation quality factor (QR,min), which also sets the limit for
the maximum obtainable impedance bandwidth, is given as
,1
)(
13min, krkr
QR += (4)
where k is the wave number (k=2п/λ) and r is the radius of the smallest sphere
enclosing the antenna. The equation is valid for linearly polarized single mode
antennas. The smaller the antenna volume on the scale of the wavelength is, the
smaller is the maximum attainable impedance bandwidth. In addition, Eq. 3
shows when the radiation quality factor increases, the unloaded quality factor
should increase as well to maintain the radiation efficiency. Consequently, to
obtain the same radiation efficiency, the sum of the inverse values of the QC and
QD should be smaller (less lossy) for a narrow band antenna compared to a
similarly sized wide-band antenna. Downsizing an electrically small antenna is,
thus, a size to efficiency and impedance bandwidth trade-off.
1.2 Package integrated antennas
Electronics miniaturization requires the downsizing of individual components as
well as an increase in their packaging density. The best result is often obtained by
increasing the integration level. A real single-chip radio, including a high
performance on-chip antenna, would be the ultimate packaging solution for
miniaturized wireless transceivers. The integration of an antenna into an
integrated circuit (IC) is, however, considered to be one of the most significant
un-solved problems for realizing single-chip radios. On-chip antennas have
researched over few decades (Buechler et al. 1986, Camilleri & Bayraktaroglu
1988, Rebeiz et al. 1988). On-chip antennas operating at frequencies below 10
GHz are mainly transducers suitable for use in very short-range inductive/near
field communications (Lin et al. 2004, Jau-Jr Lin et al. 2007). This is due to the
fact that the size of resonator type antennas operating at such frequencies is,
basically, too large compared with the typical size of an integrated circuit chip
20
(Kenneth et al. 2005, Jau-Jr Lin et al. 2007). Thus, at frequencies f > 10 GHz
antenna sizes are in same magnitude with the common chip areas. Accounting for
the unit cost per millimetre of chip area it may become affordable to integrate an
antenna on a chip. However, then inherent moderate relative permittivity of
semiconductor substrates, high dielectric losses and thin and poorly conducting
conductor materials are the major challenges for realizing efficient and wideband
antennas (Babakhani et al. 2006, Jau-Jr Lin et al. 2007, Ojefors et al. 2007,
Shamim et al. 2008).
To some extent, the problems of on-chip antennas can be overcome by using
System in Package (SiP) or System on Package (SoP) packaging strategies (e.g.,
Tummala & Laskar 2004, Tummala et al. 2004, Tummala 2006). In this approach,
a complete system (e.g. transceiver) is built and interconnected on a multilayer
organic or ceramic package (Fig. 1). Various passive functions are integrated and
embedded, while ICs are flip chipped and wire bonded on the surfaces. Highly
conducting metallization and low-loss dielectric materials are available. Wireless
communications SiPs are mainly focused on applications at microwave and
millimeter wave frequencies (~ 5–100 GHz) (Heyen et al. 2003, Brzezina et al.
2006, Pfeiffer et al. 2006, Sang-Hyuk Wi et al. 2006). In many of these
applications, a package integrated antenna has been used.
Fig. 1. A principal view of wireless communication SiP.
A planar antenna is usually printed on top of a package such that it utilizes the
topmost dielectric and metallization layers. SiP integrated antennas are virtually
not reported for applications at frequencies below 5 GHz. A probable reason for
this is that a planar antenna should be smaller than the overall package. Moreover,
for operating frequencies below 5 GHz, the antenna substrate thickness becomes
21
small compared with the wavelength. This, together with the moderate relative
permittivity of, for example, ceramic substrates results in antennas with an
inherently high radiation quality factor. The characteristic features of these
antennas include a narrow impedance BW and low to moderate radiation
efficiency (Brebels et al. 2004, Li et al. 2004, Sang-Hyuk Wi et al. 2006).
Antennas operating below 5 GHz are thus difficult to integrate into
transceiver ICs and packages. These antennas are implemented as separate
components and externally interconnected to the front-end. However, in many
applications, such as mobile terminals, antennas are mounted into the housing of
the device.
1.3 Mobile terminal antennas
A constant increase of the number of wireless communication standards being
applied in a mobile terminal has created a need for internal antennas operating at
ten or more distinct frequency bands in the frequency range of 13 MHz–6 GHz
(Fig. 2). Moreover, multiple-input multiple-output (MIMO) systems are emerging.
In MIMO systems, data is transferred simultaneously via multiple antennas. This
will further increase the number of antennas in the future.
Fig. 2. A principal view showing wireless communication standards of a future mobile
phone.
At same time, there is a trend toward highly integrated multifunctional devices
that are slimmer, smaller, and lighter than those existing currently. Thus, these
devices need to accommodate an increasing number of antennas in a
simultaneously decreasing available volume. The devices appear in various form
22
factors, e.g., slider, swivel and clamshell, and antennas should perform equally
efficiently in open and closed states. The size and shape of a device are also
important factors from the antennas’ electrical performance viewpoint.
It has been shown that the radiation of a mobile terminal is a combination of
the radiation contributed by the antenna and the metallic chassis of a device
((Arkko & Lehtola 2001, Vainikainen et al. 2002). The chassis consists of a
multilayer PCB that houses the device’s electronics, EMC shields, battery, camera,
and many other metallic parts. Typically, the largest dimension of the chassis is in
the range of 70–150 mm. At the lower end of the frequency range (13 MHz–6
GHz), the wavelength is significantly larger than the typical chassis size.
An antenna operates near its characteristic resonant frequency and couples
power to free space and to characteristic chassis wave modes. In particular, at the
lower frequencies (< 1 GHz), even 90% of the radiation may originate from
chassis wave modes. The antenna makes only a minor contribution to the
radiation. For increasing operating frequency, the radiation of the chassis modes
diminishes. The contribution is about half of the overall radiation near 2 GHz. At
still higher frequencies, the radiation of the antenna becomes predominant
(Vainikainen et al. 2002). Thus, the combination of antenna and chassis should be
considered as a single radiating structure, which is significantly larger than the
resonating antenna element alone. The theoretical minimum radiation quality
factor for the antenna-chassis combination is thus smaller compared with a
calculation in which only the antenna is accounted for. Electrical performance,
especially the impedance BW depends strongly on the dimensions of the chassis
as well as the coupling of electromagnetic energy to it. An emerged understanding
on radiation mechanisms of mobile terminals has enabled new antenna concepts
based on more purposeful utilizations of the chassis wave modes (Holopainen et
al. 2006, Schroeder et al. 2006, Vainikainen et al. 2002, Villanen et al. 2006).
Another characteristic feature of mobile terminals is related to radio wave
propagation in an urban and indoor environment. A wave experiences multiple
modifications between the transmitter and the receiver. Owing to variations in a
multipath propagation environment and varying orientations of the device, a
mobile terminal antenna should exhibit omni-directional radiation, excluding the
power radiated toward the user’s head in the talk position. This makes the total
efficiency or the average gain, the most important figures-of-merit of mobile
terminal antennas. Polarization and the precise shape of the radiation pattern are
not often considered such important factors.
23
The total efficiency of the mobile terminal antenna is greatly affected by the
user. The proximity of hand and head deteriorates the total efficiency because of
power absorption in human tissue. In addition, user proximity causes a dielectric
loading, which in turn may have an impact on the antenna’s input impedance.
This may degrade the impedance matching and detune the resonant frequency
(Boyle et al. 2007a).
Owing to a potential health risk from the power absorption in body,
governmental regulating agencies have standardized mandatory safety regulations
that set maximum limits for RF power exposure. The parameter used to measure
an amount of the power a mobile phone radiates to the human body is called
Specific Absorption Rate (SAR). It is defined as SAR = σ|E2|/р, where σ is
conductivity of body tissue, E is electric field strength, and р is density of body
tissue. SAR has unit W/kg. In the US, the SAR limit set according the IEEE is
SAR < 1.6 W/Kg averaged over volume corresponding 1g of tissue (ANSI/IEEE
1999). In EU the limit is set according International Commission on Non-Ionizing
Radiation (ICNIRP) and, it is SAR < 2.0 W/Kg averaged over volume equal to 10
g of tissue (ICNIRP 1999). The SAR measurement procedures are also
standardized (European Std. 2001. IEEE Std 2003, IEC and CENELEC 2005).
1.4 Objectives and outline of the thesis
The objective of this thesis is to develop and research bandwidth enhanced
antennas for mobile terminals and multilayer ceramic packages. Special attention
has been paid to electrically frequency-tunable mobile terminal antennas (chapter
2) and multilayer ceramic package integrated microwave antennas (chapter 3).
The mobile terminal antennas in the chapter 2 are application driven and operate
at frequency bands allocated to existing telecommunication standards. The main
purpose of the antennas in chapter 3 is to demonstrate the technologies rather than
be specific for any individual application. The impedance BW of antennas
operating at the cellular bands (GSM850 (824–894 MHz), GMS900 (880–960
MHz), GSM1800 (1710–1880 MHz), GSM1900 (1850–1990 MHz) and UMTS
(1920–2170 MHz) is given against 3 dB and 6 dB return loss matching criteria.
Otherwise 10 dB matching criteria is used.
The antennas presented were designed and optimized with commercial EM
simulation software and empirical procedures and measured in free space.
Measurements accounting for the mobile phone user interactions or SAR values
are considered being out of the scope of the thesis. The antenna design and
24
optimization are thought to be generic information. Thus, detailed descriptions of
these processes, such as the simulation results, are left beyond the scope of the
thesis or are given only to a brief extent. The research topic of small antennas has
been the subject of intensive research in recent years. Relevant literature studies
are included in the beginning of each section and the results are discussed at the
end of both chapters.
Section 2.1 deals with frequency-tunable planar inverted-F antennas (PIFAs).
Sections 2.1.1 and 2.1.2 present two frequency tuning schemes for a dual-band
PIFA switched to operate in four frequency bands. In section 2.1.3, these tuning
methods are combined within a single dual-band antenna structure, which can be
switched to operate in eight frequency bands. All PIFAs operate at frequency
ranges appropriate to the five cellular bands. Section 2.2 deals with frequency-
tunable monopole antennas. Section 2.2.1 presents an implementation of a
compact varactor-tuned meander line monopole antenna for DVB-H signal
reception (470-702 MHz). The last section of the chapter, 2.2.2, describes the
tuning of a dual-band planar monopole antenna for operating in four cellular
bands.
In section 3.1, the use of a laser micro-machined embedded air cavity is
proposed to enable a size to bandwidth trade off for a 10 GHz microwave
microstrip antenna on a multilayer LTCC substrate. Section 3.2 describes the
implementation of an X-band surface mountable LTCC BGA antenna package, a
stacked patch microstrip antenna designed for optimal BW performance and
integrated into the package substrate. Besides the electrical design optimization, a
circuit model describing the antenna structure is presented. Section 3.3 presents a
frequency-tunable folded slot antenna utilizing an integrated BST varactor. The
antenna operates near 3 GHz and can be fabricated with a standard screen printing
process with silver conductors.
25
2 Frequency-tunable antennas for mobile terminals
In present mobile terminals, the volume reserved for the antenna is often
commercially restricted. According the fundamental physical limitations it is a
major challenge to make an antenna in the small available volume that covers a
single very wide frequency band or multiple frequency bands with high total
antenna efficiency. To some extent, the problem of covering wide bandwidths can
be overcome through the use of multiple antennas, but then, more space and
interconnections are required for the antennas, and mutual coupling between the
individual antennas may cause severe problems (Boyle & Massey 2006, Diallo et
al. 2006).
The current mobile terminal antennas are mainly of wide-band and multiple
band type. State-of-the-art cellular antennas may have a quad-band or penta-band
operation (Bhatti & Park 2007, Tzortzakakis & Langley 2007, Villanen et al.
2007, online 2008a, online 2008b). However, in many of these antennas, the
multiple band operation is achieved at the expense of increased antenna size
(Wong 2003, Mak et al. 2007). In addition, compromises in impedance matching
and/or radiation efficiency between the bands are often necessitated. This is
especially true for antennas of small devices that have a multiband operation at
lower frequencies, for example in the GSM850, GSM900 and DVB-H (470–702
MHz) bands. These antennas often utilize at least two relatively long quarter
wave resonators to create adjacent resonances and the maximum obtainable
impedance bandwidth is greatly reduced because of the small size of the chassis.
Antennas that have an electrically tunable operating frequency, together with
frequency-invariant radiation characteristics, would have clear technical benefits.
This kind of antenna will not cover all the frequency bands simultaneously, it may
provide several dynamically selectable narrow frequency bands and, within these
bands, exhibit higher efficiency than is achievable with conventional wide-band
and multiple band antenna solutions (Aberle et al. 2003). This approach may be
used to downsize the antennas while maintaining the operating bandwidth, or,
alternatively, to increase the usable bandwidth of the antenna without increasing
the physical size. In addition, antenna tuning can be used to compensate for
effects that occur when the operating frequencies of handset antennas are detuned
by the proximity of the user or by changes in the operating environment, such as
occur when opening and closing a clamshell or a swivel type mobile phone
(Sjoblom & Sjoland 2005, Boyle & Massey 2006, Jae-Hyoung Park et al. 2007).
26
In a wider context, reconfigurable antennas are used in space and military
applications and are proposed as one of the key enablers for the software defined
radios of the future (Aberle et al. 2003, Luo & Dillinger 2003, Yang & Rahmat-
Samii 2005 Christodoulou et al. 2006).
However, there are some conceivable drawbacks to frequency-tunable
antennas, which exhibit high RF power loss, poor linearity and high DC power
consumption of active tuning circuit components. It should also be noted that
these antennas are often complicated structures to design and implement, which
may have cost, reliability and RF front-end compatibility implications.
Withstanding these limitations, frequency tunable antennas provide significant
technical and economic advantages over their conventional counterparts.
2.1 Frequency- tunable dual-band planar inverted-F antennas
Planar inverted-F antennas (PIFAs) are among the most widely used antennas in
mobile terminals. This is due to their small size, low cost and ease of design and
manufacturing. Additionally, PIFAs can be easily modified for multiband
operation and concealed in mobile phone housings (Wong 2003). The PIFA
concept has been researched over decades (King et al. 1960, Taga Tsunekawa
1987, Tsunekawa 1989). Recently, PIFA structures covering the frequency range
of five and six telecommunication standards have been reported for applications
in mobile terminals and base stations. (Park et al. 2006, Sanz-Izquierdo et al.
2006, online 2008a, online 2008b).
A common approach for tuning the operating frequency of a PIFA is to
electrically reconfigure a short circuit connection with an external tuning circuit.
This has been used for single-band and dual-band antennas (Louhos & Pankinaho
1999, Karmakar et al. 2003, Karmakar 2004). Other previously reported tuning
methods include the use of an adjustable reactive component between a PIFA
patch and a ground plane (Louhos & Pankinaho 1999, Okabe & Takei 2001,
Panayi et al. 2001, Mak et al. 2007), or switched tuning stubs / reactive loads,
which have been applied in both single-band and dual-band PIFAs (Kivekäs et al.
2002, Ollikainen et al. 2002, Boyle & Steeneken 2007b). In addition, a dual-band
PIFA has been tuned to operate in three frequency bands by changing the feed
position using a RF switch (Mak et al. 2007).
However, in these tuning methods, the tuning circuit is separated from the
antenna and built either next to, or under, the antenna element. Tunable antennas
implemented in this way utilize separate DC wiring for controlling the state of the
27
RF switches or variable capacitors. These wires, like any metal objects in close
proximity of an antenna, may cause severe performance degradation, such as
deformation of the radiation pattern and undesired detrimental resonances
(Anagnostou et al. 2006). Furthermore, external tuning circuits increase the
overall size of antennas.
2.1.1 A frequency tuning method for a planar inverted-F antenna
The tuning method proposed in Paper I and (Komulainen et al. 2006) is based on
inserting a RF switch into an antenna element. Unlike the situations proposed by
Panaia et al. (2004) and Karmakar (2005), no separate DC control circuit for the
switch is used. The tuning circuit is integrated using reactive passive components
that form separate paths for the RF signal and DC control voltage. The DC
voltage is carried to the switch simultaneously with the RF signal. The proposed
tuning method has been demonstrated with a single-feed dual-band PIFA
operating in four distinct frequency bands (Fig. 3).
Fig. 3. Dual-band PIFA with an integrated tuning circuit. Dimensions are in mm.
[Paper I]
The antenna is a well-known PIFA type (Tarvas & Isohatala 2000, Kivekäs et al.
2002, Lindberg & Ojefors 2006,). The rectangular patch is slit to form two
separate paths for resonating currents, and this leads to dual-band operation. The
antenna element is built on a 0.8 mm thick R4003C printed circuit board
(εr = 3.38, tan δ = 0.0027 at 10 GHz) with 35 μm thick copper metallization. It
has a 9.2 mm thick air substrate and it is placed on top of a 45 mm × 110 mm
28
reverse side grounded R4003C PCB. This PCB mimics the chassis of a mobile
phone.
The short circuit point is separated from the rest of the antenna element
metallization by a narrow L-shaped slit. In order to generate a DC potential
difference across the diode, two capacitors (C = 100 pF) and an inductor
(L = 150 nH) are connected in parallel with the slit. The components form a
simple bias-T circuit. The capacitors, with low impedances at RF, act essentially
as a DC block and an RF short, and thus allow RF currents to flow freely in the
antenna element. The inductor with high RF impedance blocks the RF signal and
passes the DC. The DC voltage can be brought to the diode with the RF signal, so
that there is a DC potential difference between the feed and the short circuit pins.
Consequently, changing the polarity of the DC voltage to produce either a
forward bias (switch ON) or a reverse bias (switch OFF) condition for the diode
will control the switch impedance. The two capacitors were used in order to
minimize the impact of the bias-T circuit on the antennas resonant frequencies.
However, the antenna can be probably implemented also using a single capacitor.
The simultaneous frequency shift of both frequency bands can be explained
by using the simulator (Ansoft HFSS v. 10) to show the surface currents on the
antenna (Fig. 4). In the lower frequency band, a larger amount of current flows
through the diode in the ON state, and almost no current passes through the diode
in the OFF state. Hence, frequency tuning of the lower band is simply based on
modifying the length of the current path. The amount of current through the diode
is smaller in the upper band than in the lower band. This indicates that the
function of the switch is to load the antenna by changing the coupling between the
two resonators separated by the narrow slit. When the switch is ON, strong
surface currents occupy a larger area compared with that for the OFF state, thus,
the resonant frequency is lower in the ON state. The frequency shifts can also be
understood by the manner in which the switch is used for cutting the G-shaped slit
at the pre-defined position.
29
Fig. 4. Simulated surface current distribution plots: (a) in the lower band when the
switch is ON (890 MHz); (b) in the lower band when the switch is OFF (930 MHz); (c) in
the upper band when the switch is ON (1770 MHz); (d) in the upper band when the
switch is OFF (1960MHz). [Paper I]
An external bias-T component provided the DC control voltage for the switch
during the measurements. A Satimo Starlab near-field chamber was used for the
radiation pattern and total antenna efficiency measurements (Iversen et al. 2001).
The ON and OFF conditions for the switch were obtained using ±1 V (with a 20
mA forward current), respectively. The measured return losses of the tunable
antenna are presented in Fig. 5. When the switch is ON, the resonances are at 854
MHz and 1770 MHz. The antenna covers the frequency ranges of the GSM850
and GSM1800 bands within at least a 3 dB impedance bandwidth. The BWs for
the lower band in this state is 80 MHz (3 dB) and 45 MHz (6 dB). The
corresponding values for the upper band are 354 MHz (3 dB BW) and 198 MHz
(6 dB BW). When the switch is OFF, the resonant frequencies are 931 MHz and
2025 MHz. The GSM900 and UMTS bands are thus covered. The PCS1900 band
is between the operating frequencies, so that the RX and TX portions are
separated. The lower band has 3 dB and 6 dB BWs of 146 MHz and 85 MHz, and
corresponding values for the upper band are 418 MHz and 216 MHz, respectively.
The measured return loss of the reference antenna, not incorporating the
tuning circuit and additional slits at the component positions, showed well
30
impedance matched (the peak S11 better than 13 dB) resonances at 815 MHz and
1800 MHz. The lower band of the reference antenna showed 3 dB impedance BW
of 64 MHz and 6 dB impedance BW of 35 MHz. The corresponding values at
upper band were 380 MHz and 205 MHz, respectively.
Fig. 5. Measured return losses of the PIFA with the integrated tuning circuit. [Paper I]
The measured total efficiencies of the tunable antenna are presented in Fig. 6. The
results show peak values of 68% and 88% in the switched ON state and 65% and
85% in the OFF state. Thus, the antenna covers the frequency bands of GSM1800,
PCS1900 and UMTS with a 65% total efficiency margin. The total efficiency
coverage for the GSM900 and GSM850 bands were 50% and 40%, respectively.
The measured peak return loss values show that all the bands of the antenna
exhibit a good impedance match, with peak return losses better than 16 dB. Thus,
the maximum total antenna efficiencies are nearly equal to the maximum
radiation efficiencies, which are 69% and 88% in the switched ON state, and 65%
and 87% in the OFF state. No appreciable difference in the efficiencies between
the switched ON state and the switched OFF state was observed.
31
Fig. 6. Measured total antenna efficiencies of the PIFA with the integrated tuning
circuit. [Paper I]
The reference antenna showed peak radiation efficiency values of 84% and 95%
for the lower and upper bands, respectively. Radiation efficiency deterioration
caused by the tuning circuit can be estimated at less than 20% for the lower bands
and less than 10% for the upper bands. In addition, slight downward frequency
shifts of 40 MHz and 30 MHz were observed for the lower and upper resonant
frequencies compared with the tunable antenna in the ON state. Probable the
reason for the slight frequency shifts is related to the loading caused by the tuning
circuit components and the additional slit that separates the short circuit pin from
the rest of the antenna element metallization.
The measured far-field radiation patterns at the center operating frequencies
of both antennas are presented in Fig. 7. The patterns of the lower frequency
bands are near omni-directional and exhibit similarities with the pattern of a Z-
directed dipole antenna. The maximum lower band gain values of the tunable
antenna are 1.8 dBi in the ON state and 2.2 dBi in the OFF state. The
corresponding value for the reference antenna was 2.5 dBi. The antenna is able to
maintain the shape of the patterns despite the presence and use of the tuning
circuit. The patterns of the upper bands were found to be more directional, with
maximum gains of 4 dBi (switch ON), 3.6 dBi (switch OFF) and 4.3 dBi
(reference antenna). The upper band radiation patterns for the reference antenna
and the switched ON tunable antenna are quite similar. The use of the tuning
circuit, however, slightly tilts the pattern in the OFF state. This can be seen best
32
from the radiation plot in the XY plane. Nevertheless, the antenna is able to
maintain its efficiency and there is little degradation in the direction of radiation.
The essentially constant, dipole-like radiation patterns in the lower frequency
bands can be attributed to the dominant radiation contribution of the chassis wave
mode. At frequencies near 2 GHz, more radiation originates from the antenna
element, and thus the antenna is more directional, and a slight tilt in the upper
band patterns tends to occur when the antenna is tuned.
Fig. 7. Measured far-field radiation patterns at the center operating frequencies of the
PIFA with the integrated tuning and the reference PIFA without the tuning circuit.
[Paper I]
2.1.2 Use of a switchable parasitic antenna element
A similar dual-band PIFA can be tuned for a quad-band operation using a
switchable parasitic antenna element [II]. The principle has been previously
proposed for tuning the operating frequency of a PIFA mechanically by adjusting
the height of the parasitic element with a screw (Chiu et al. 2007). Furthermore, a
33
similar principle for tuning a dual band PIFA with a SPDT type of switch has
been patented (Milosavljevic 2005).
The dual-band PIFA (Fig. 8.) is similar to the one presented in the previous
section, except a parasitic antenna element is placed between the main element
and the ground plane. The parasitic element consists of vertical and horizontal
parts made of the R4003C PCB and a 0.3 mm thick copper plate, respectively.
Fig. 8. Frequency-tunable PIFA based on a switchable parasitic antenna element.
Dimensions are in mm. [Paper II]
The main antenna element is coupled electromagnetically to the parasitic patch
and the ground plane. Changing this coupling provides the basis for electrically
controlled frequency tuning. When the parasitic element is connected to the
ground potential, it loads the antenna capacitively. This loading shifts the resonant
frequencies downwards. It has been shown that the capacitive loading effect also
reduces the impedance BW (Rowell & Murch 1997, Chiu et al. 2007). When the
element is disconnected from the ground, the loading occurs more weakly and the
resonant frequencies are shifted upwards towards their initial positions. Both
resonant frequencies can be simultaneously tuned by either connecting or
disconnecting the parasitic element to the ground plane. A practical
implementation of the frequency tuning is made with two parallel RF PIN diode
switches. The switches are situated on the vertical parasitic element part. A
control voltage for the diodes is supplied through a thin (400 μm) DC line. The
line is connected to the upper part of the vertical PCB metallization through a 220
34
nH inductor. The lower end of the PCB is grounded. The inductor works as a RF
block and a DC short, and hence forces the RF signal to flow along the intended
route in the parasitic element.
BAR50-02V RF PIN diode switches and standard 0805-packaged passive
components were used in the fabricated antenna structure. The forward current of
the diodes was limited to 20 mA. The measured return losses of the tunable PIFA
are presented in Fig. 9. When the switch is ON, the resonances are at 845 MHz,
1765 MHz, and 2660 MHz. The 3 dB impedance BWs are (6 dB BW in brackets)
81 MHz (42 MHz), 182 MHz (105 MHz), and 194 MHz (not available) in the
switched ON state. The antenna thus covers the frequency ranges of GSM850 and
GSM1800 with a 3 dB impedance BW. The third band does not cover any specific
frequency range but, however, it is rather close to the 2.4 GHz WLAN band
(2400–2484 MHz). This resonance occurs when the parasitic element is shorted,
i.e., a quarter wave length resonator. While the switch is OFF, the resonances are
at 905 MHz and 1920 MHz. 106 MHz (61 MHz) and 353 MHz (191 MHz).The
antenna almost covers the GSM900, PCS1900, and UMTS bands with 3 dB
return loss margin.
Fig. 9. Measured return losses of the frequency-tunable PIFA based on the switchable
parasitic antenna element.
The measured return loss of the reference antenna not incorporating the parasitic
antenna element showed two bands at 935 MHz and 2040 MHz. The reference
antenna is otherwise similar to the tunable one, except that the parasitic element is
removed. The resonant frequencies are 90 MHz and 275 MHz higher compared
35
with the switched ON tunable antenna and 30 MHz and 120 MHz higher than in
the switched OFF state. The 3dB (6 dB) BWs are 106 MHz (62 MHz) and 450
MHz (239 MHz). As expected, the presence of the parasitic antenna element
shifts all the frequencies downwards. This means that the reference antenna
operating at the same frequencies would result in a larger size.
It can be seen that when the switches are OFF, the capacitive loading is rather
weak. The resonant frequencies and the impedance BWs are only slightly lower
compared with the reference antenna. When the parasitic element is switched to
the ground (switches ON), the loading effect occurs more strongly. The resonant
frequencies and the BW are now significantly lower compared with those of the
reference antenna.
The measured total efficiencies of the tunable antenna are presented in Fig.
10. The results show peak values of 52%, 81% and 51% in the switched ON state.
The corresponding values in the switched OFF state are 52% and 85%. The
antenna covers the GSM850 and GSM900 bands with a 25% total efficiency
margin. The efficiency coverage of the GSM1800, GSM1900, and UMTS are
35%, 65% and 40%, respectively. Except for the one at 2660 MHz, all the
frequency bands are well impedance matched (peak S11 better than 12 dB), thus
the radiation efficiencies at the center frequencies are almost equal to the total
efficiencies. The radiation efficiencies are 56% and 81% for the switched ON
state and 52% and 86% for the switched OFF state. No significant differences
between the switched ON and switched OFF state are observed.
The reference antenna showed peak total efficiency values of 66% and 90%.
These values are also the maximum radiation efficiencies. The deterioration of
radiation efficiency caused by the switchable parasitic antenna element is
estimated to be less than 20% in the lower bands and less than 10% in the upper
bands. There is a clear lower band efficiency difference (~ 14%) between the
reference antennas presented in the current and the previous sections. In the upper
band, the difference is smaller (~ 5%). The probable reasons for the difference are
tolerances in the assembly height or measurement related tolerances. It should
also be noted that the sizes of the antenna elements and the size of the reverse side
grounded PCBs as well as the resonant frequencies are different thus the results
are not fully comparable.
36
Fig. 10. Measured total antenna efficiency of the frequency-tunable PIFA based on the
switchable parasitic antenna element. [Paper II]
The measured radiation patterns of both antennas (Fig. 11.) are very similar to the
patterns of the antenna with the integrated tuning circuit. The antenna has
maximum lower band gain values of 0.7 dBi (switched ON), 0.9 dBi (switched
OFF) and 2.8 dBi (the reference antenna). The upper band pattern is more
directional. The maximum gain values are 3.9 dBi (switched ON), 4.1 dBi (switch
OFF) and 4.5 dBi (the reference antenna). The third frequency band in the
switched ON state (2660 MHz) showed a rather irregular pattern with a 1.4 dBi
maximum gain value. The maximum gain values at lower bands are about 1 dBi
lower when compared to the PIFA with the integrated tuning circuit presented in
the previous section. Since the pattern shapes at the lower bands are quite similar,
the gain difference can be explained with the lower total antenna efficiency
presented in this section. At the upper bands the maximum total antenna
efficiencies (80–90%) and maximum gain values (~ 4 dBi) are in similar
magnitude.
37
Fig. 11. Measured far-field radiation patterns of the frequency-tunable PIFA based on
the switchable parasitic antenna element and the reference PIFA, excluding the
parasitic antenna element. [Paper II]
2.1.3 Switching a dual-band planar inverted-F antenna to operate in eight frequency bands
By combining the two previously presented tuning circuits, a dual-band PIFA can
be switched to operate in eight frequency bands [III]. The antenna structure is
depicted in Fig. 12. The antenna element (38 mm x 21 mm) is on a 45 mm x 110
mm groundplane. The operation principle is straightforward. The switch in the
dual-band antenna element is used to generate four frequency bands. These four
38
bands can be further tuned by switching the parasitic antenna element. Antennas
of the two previous sections exhibited very similar frequency-tuning ratios. To
avoid overlapped bands in the eight band switchable antenna, modifications of the
structure were required. The height of the parasitic element was reduced by 0.5
mm and the diode position in the antenna element was changed in the manner that
larger frequency shifts in the lower and upper bands could be achieved. The size
and position of the parasitic antenna element were maintained.
Fig. 12. Eight band switchable dual-band PIFA. [Paper III]
The measured return losses are presented in Fig. 13. The results show four
separate lower frequency bands. In the upper frequency range, three separate
bands and one overlapping band are observed. The antenna covers the frequency
bands of 760–1075 MHz and 1530–2170 MHz with at least 3 dB return loss
margins. The frequency ranges of the GSM850, GSM900, GSM1800, PCS1900
and UMTS telecommunication standards are within these frequency ranges. The
ON-ON-state shows the narrowest impedance 3dB (6dB) BWs that are 92 MHz
(50 MHz) for lower band and 216 MHz (117 MHz) for the upper band. The
widest BWs, 177 MHz (106 MHz) at the lower band and 342 MHz (178 MHz) at
the upper band are obtained in OFF-OFF-state. This correlates with the results of
the previous chapter, where the capacitive loading of the parasitic antenna
element narrowed the BWs. The parasitic antenna element generates a well-
radiating resonance near 2600 MHz. This resonance is very similar to the one
described in the previous section, and for the sake of clarity it is not graphically
presented here. The measured total antenna efficiencies are presented in Fig. 14. The
maximum total efficiencies of the lower and upper bands are around 50% and
39
80% (excluding the overlapped band). The lower frequency limit of the
measurement system used was 800 MHz. Thus, it was not possible to properly
measure the efficiency of the lowest band. The overall total efficiency coverage in
the frequency range of ~ 750–1080 MHz is better than 25% and 1560–2120 MHz
is better than 50%. The size of the ground plane here is equal to the reference
antenna in section 2.1.1. The sizes of the antenna elements are also nearly equal.
The reference antenna showed well impedance matched resonances at 845 MHz
and 1900 MHz with corresponding peak radiation efficiencies of 82% and 93%.
The peak radiation efficiency deterioration caused by tuning can be estimated to
be approximately 35% in the lower bands and 15% in the upper bands.
Fig. 13. Measured return losses of the eight band switchable PIFA. [Paper III]
Fig. 14. Measured total efficiency of the eight band switchable PIFA. [Paper III]
40
The radiation patterns (Fig. 15) are very similar to the ones presented in the two
previous sections. The peak gain values at the lower bands range from -0.15 dBi
to 0.95 dBi. Only minor variations in the patterns of the antenna’s operating
modes can be observed. The upper band patterns are more directed (max. gains
from 2.5 dBi to 3.6 dBi) and larger pattern variations compared with the lower
bands can be observed. Finally the main results presented in three previous
sections are summarized in Table 1.
Fig. 15. Measured far-field radiation patterns and maximum gain values of the eight
band switchable PIFA. [Paper III]
41
Table 1. Result summary of the frequency-tunable dual-band PIFAs (Papers I-III).
Antenna,
Size [mm],
Gnd size
[mm]
Number of
tuning
states
BW lower
band 3dB
[MHz]
BW lower
band 6dB
[MHz]
BW upper
band 3dB
[MHz]
BW upper
band 6dB
[MHz]
Radiation
efficiency at
band center
LB*, UB**
Radiation
efficiency
drop vs. ref.
antenna***
LB,UB
Paper I
39.5×21×9.2
110×45
2
State 1
State 2
815-895
869-1015
831-876
893-978
1600-1954
1859-2283
1669-1867
1920-2145
69%, 88%
65%, 87%
-15%, -7%
-19%, -8%
Paper II
34×20×9.2
105×40
2
State 1
State 2
809-889
853-958
827-869
873-935
1678-1860
1770-2123
1714-1819
1834-2025
56%, 81%
52%, 86%
-10%, -9%
-14%, -4%
Paper III
21×38×9.2
110×45
4
State 1
State 2
State 3
State 4
764-856
790-898
844-1006
895-1072
780-830
810-876
884-966
927-1033
1534-1750
1595-1942
1753-1947
1825-2167
1580-1697
1666-1847
1783-1875
1877-2055
42%, 77 %
50%, 81%
49%, 61%
51%, 80%
-40%,-20%
-32%, -12%
-33%, -32%
-31%, -13%
* Lower band, ** Upper band, *** The reference antenna is described in text.
2.2 Frequency-tunable planar monopole antennas
Monopole antennas are widely used in mobile terminals. The first
implementations were of the whip type and protruded from the housing of the
device. Monopole antennas in a planar configuration have a low profile and can
be built into devices. Recently, planar monopole antennas capable of operations in
five and six bands have been presented (Freeman et al. 1992, Hyun-Jeong Lee et
al. 2007, Li et al. 2007, Thornell-Pers & Erlandsson 2007, Wu & Wong 2008).
From the frequency-tuning point of view, planar monopole antennas differ from
PIFAs. Planar monopoles do not have a ground plane beneath. Thus, a connection
to a ground cannot be done at an arbitrary position without using separate wires
like for ground plane antennas, such as PIFAs.
Frequency-tunable planar monopole antennas have been presented in the
literature. The most frequently used tuning approach is to switch between
different lengths of a monopole (Freeman et al. 1992, Zachou et al. 2006, Jung et
al. 2007, Thornell-Pers & Erlandsson 2007, Zhang et al. 2007). The tuning occurs
since the monopole length is approximately equal to a quarter wave length at the
operation frequency. Varactors have also been used between the antenna element
and the ground similarly as for PIFAs (Berg et al. 2007, Jung et al. 2007, Yoon et
42
al. 2008). Tuning circuit biasing schemes based on both separate DC wiring and
external bias T have been used.
2.2.1 Compact varactor-tuned meander line monopole antenna for DVB-H signal reception
Digital television services have become present-day technology in handheld
devices. The Digital Video Broadcasting-Handheld (DVB-H) standard utilizes the
frequency range of 470–702 MHz (ETSI 2005); hence, the reception antenna
should be able to cover a relative bandwidth of 40%. In addition, the free-space
wavelength at the lower end of the band is notably large (λ0 = 64 cm) compared
with the typical size of a handheld device. These facts form a major challenge to
the implementation of an internal DVB-H antenna in a small device.
In this frequency range, the chassis mode is the primary radiation mechanism.
The radiation properties, especially the impedance BW, depend strongly on the
size of the chassis. The use of non-resonant reactive coupling elements with
impedance matching circuits has been proposed for internal DVB-H antennas
(Holopainen et al. 2006, Wong et al. 2006). In this approach, the wave modes are
excited in the chassis without using a conventional resonating antenna element,
and the chassis acts as a wideband radiator. These solutions provide a compact
way to build internal DVB-H antennas. However, so far, they have been
implemented only in relatively large (75 mm x 130 mm and 40 mm x 200 mm)
bar or folder-type devices. It can be assumed that such wide-band radiators are
more difficult to implement in smaller devices.
DVB-H antennas for smaller devices have also been proposed. Choi et al.
(2006) presented a wide-band modified monopole antenna for an 85 mm x 40 mm
bar-type device. The size of the antenna element was 25 mm x 15 mm x 4 mm
(1.5 cm3) and according to the author’s best knowledge, it is the smallest self-
resonant DVB-H antenna presented in the literature.
The channel width in the DVB-H standard is only 8 MHz (ETSI 2005), and
therefore, a frequency-tunable narrowband antenna with a wide tuning range is
also usable. Such antennas have been implemented using RF switches and
variable capacitors (varactors) (Suzuki et al. 2006, Berg et al. 2007, Libo Huang
& Russer 2007, Milosavljevic 2007). However, these structures have larger
antenna elements compared the one presented by Choi et al. (2006).
In paper IV, a compact varactor-tuned meander line monopole antenna for the
DVB-H signal reception was presented. The antenna structure (Fig. 16) is built on
43
a 0.8 mm thick reverse-side grounded R4003C printed circuit board with 35 μm
thick copper metallization.
Fig. 16. (A) compact varactor-tuned meander line monopole antenna; (B) detailed view
of the antenna element. Dimensions are in mm. [Paper IV]
A short circuited matching stub similar to the one proposed by Sung & Feng
(2006) is used to increase the inductance and thus improve impedance matching.
A microstrip feed line (Z0 = 50 Ω), the matching stub and two tuning circuit
components are situated on the PCB horizontally, while the uniform meander line
section of the antenna is built on a vertically extended PCB end. The size of the
horizontal part of the antenna element is 34 mm x 7 mm x 1.8 mm, including the
component height of 1 mm. The size of the vertical section is 34 mm x 0.8 mm x
10 mm, and thus the overall volume is 0.7 cm3. The physical length of the
meander line section is ~ 140 mm. The overall length of the resonator is ~160 mm
which is 0.37 λ0 at 700 MHz and 0.25 λ0 at 470 MHz.
The main idea of the tuning circuit implementation is similar to that used
previously in PIFAs. Separate RF and DC signal paths are created to the antenna
structure. The varactor is placed between the end of the meander line section and
a grounded pad. The purpose of the varactor is to load the antenna with a voltage-
controlled capacitance (0.2–1.2 pF) and, hence, it tunes the antenna’s resonant
frequency. In order to create a DC voltage across the varactor, a DC block
44
capacitor is used as illustrated. The 100 pF capacitor acts as a DC block and an
RF short, isolating the DC ground potential of the short-circuited matching stub
from the rest of the antenna structure. Consequently, the DC voltage can be
supplied across the varactor with the RF signal.
A surface-mountable GaAs varactor (Metelics MVG-125-20-0805-2) and a
standard 0805-packaged capacitor were used in the fabricated antenna structure.
The measured return losses of the antenna with varactor reverse bias voltages
between 2.5 V and 17.5 V are presented in Fig. 17. The operating frequency was
continuously tuned over the DVB-H band. The narrowest 10 dB matching BW
(VSWR ≤ 2) was measured at 470 MHz, and it was 8 MHz. The widest measured
BW was 16 MHz, at 700 MHz. Thus, the antenna fulfils the required channel
width criteria (> 8 MHz).
The measured far-field radiation patterns in the XZ and XY planes at 470
MHz and 702 MHz are presented in Fig. 18. The patterns are omni-directional in
the XZ plane being similar to those of the Y-directed dipole antenna. The
essentially constant patterns can be attributed to the dominant radiation
contribution of the longitudinal dipole-like current distribution of the chassis
wave mode. This also explains that the radiation is linearly polarized along the Y
axis, and the cross polarization levels are significantly lower compared to the co-
polarization levels. The maximum gain as a function of frequency and the DVB-
H specification are presented in Fig. 19. The results show that the maximum gain
is -8.5 dBi and 0.5 dBi at 470 MHz and 702 MHz, respectively. These values are
1.5 dB and 7.5 dB better than required by the DVB-H standard.
45
Fig. 17. Measured return losses of the frequency-tunable meander line monopole
antenna with varactor bias voltages ranging from 2.5 V to 17.5 V. [Paper IV]
Fig. 18. Measured radiation patterns of the varactor-tuned meander line monopole
antenna in the (A) XZ plane (B) XY plane. [Paper IV]
The observed decrease in gain toward the lower edge of the DVB-H band can be
explained by a deterioration of the radiation efficiency, since the directivity is not
substantially changed, despite the tuned operating frequency. The probable
reasons for this are a weaker coupling to the ground plane at the lower
frequencies and more power loss in the varactor component. For further
improvements one needs to research whether the coupling to the ground plane or
the loss of the varactor forms the main source of losses in this structure.
46
Fig. 19. Measured maximum gain of the varactor-tuned meander line monopole
antenna and the DVB-H gain specification. [Paper IV]
2.2.2 Frequency-tunable planar monopole antenna for a quad-band operation
A frequency-tunable dual-band planar monopole antenna operating at cellular
frequencies is described in paper V. The antenna structure is similar to the DVB-H
antenna of the previous section. The antenna structure presented in Fig. 20 is built
on 40 mm x 100 mm x 0.8 mm R4003C PCB. The radiating element that
generates the upper band is a rectangular patch, while the lower bands utilize a
1.8 mm wide strip. Both resonators have two-dimensional forms achieved by
vertically extending the end of the PCB. In order to improve the impedance
matching of the lower band, a short-circuited stub with a length of 14 mm is used.
A BAR-50-02V RF PIN diode switch is placed on the longer resonator of the
antenna. The length of the resonator can be switched between 89 mm and 79 mm.
These values correspond to the free-space quarter-wavelengths at 850 MHz and
950 MHz.
The measured return losses of the antenna are presented in Fig. 21. The return
loss measured with the switch OFF shows resonances at 954 MHz and 1840 MHz.
The 3dB (and 6 dB) BW are 165 MHz (90 MHz) and 425 MHz (232 MHz) for
the lower and upper bands, respectively. In this state, the antenna covers the
frequency ranges of GSM900, GSM1800 and GSM1900 as is shown in Fig 21.
When the diode is switched ON, the antenna resonates at 855 MHz and 1815
MHz, covering the GSM850 (824–894 MHz) and GSM1800 bands. The 3 dB
47
(and 6dB) BW are 86 MHz (49 MHz) for the lower band and 318 MHz (164
MHz). The results show that the frequency tuning of the lower band also has an
impact on the bandwidth of the upper band. A probable reason for this is that the
lower band tuning changes the coupling between the resonators.
Fig. 20. Frequency-tunable dual-band planar monopole antenna A) the antenna
element, B) top view, C) bottom view. [Paper V]
Fig. 21. Measured return losses of the frequency-tunable dual-band monopole antenna.
[Paper V]
The radiation patterns measured at the center frequencies of the bands are
presented in Fig. 22. The lower bands show omni-directional patterns (the chassis
mode) with maximum gains of 2.2 dBi for the switched OFF state, and 0.9 dBi
48
for the switched ON state. The patterns of the upper bands prove to be more
directional, with a maximum gain of around 3 dBi for both states of the switch.
Despite the use of the tuning circuit, the upper band patterns maintain nearly
similar shapes and gain levels. The lower band maintains the pattern shape, but
systematic gain level deterioration of 1–1.5 dB can be observed between the
states. This can be attributed to the lowered radiation efficiency of the GSM850
band when the diode is ON. A probable reason for this is the increased power loss
in the diode that occurs in the ON state. This problem might be reduced by using
a RF diode switch with a lower on-state resistance. There is no measured
efficiency data available for this antenna structure.
Fig. 22. Measured radiation patterns of the lower bands of the frequency-tunable dual-
band planar monopole antenna in A) the YZ-plane, B) the XY-plane, and C) the XZ-
plane, and of the upper bands of the antenna in D) the YZ-plane, E) the XY-plane, and F)
the XZ-plane. [Paper V]
49
2.3 Discussions
The frequency tuning method proposed for PIFAs in paper I has several beneficial
features. The integrated tuning circuit allows more independent designs of the
antenna and the rest of the device when compared with external tuning circuit
techniques. This is because no parts of the tuning circuit, except for the bias-T for
combining RF and DC, are situated outside the antenna element. Moreover, the
method enables simultaneous tuning of several frequency bands. To some extent,
the amount of the simultaneous frequency-tuning of two or more bands can be
controlled by means of conventional antenna design and varying the switch
position in the antenna element. Besides, an inductor with a proper inductance
value can be added to the antenna element to control the tuning of bands more
independently. The inductor should act as a block (high impedance) at the higher
band and a short (low impedance) at the lower band (Yeh & Wong 2002). This
would be a particularly useful solution for antennas having a clear separation
between the bands, for example, for an antenna operating at the lower and upper
GSM bands.
In principle, the proposed concept can be extended to any kind of PIFA
structure. Replacing the RF diode with a very low loss RF MEMS switch may
further improve the electrical performance. In addition, RF varactors are potential
candidates for providing continuous frequency tuning over the desired frequency
band. The use of multiple switches or varactors in the same antenna element may
also be considered for yielding more versatile functions, such as a greater sub-
division of the overall frequency band, and also independent tuning of one or
more sub-bands in a multi-band antenna.
The frequency-tunable dual-band PIFA based on the switchable parasitic
antenna element in paper II was also implemented without increasing the overall
volume of the antenna. However, in many practical applications, the volume
beneath the antenna element is reserved for other functions, such as speakers. The
capacitive loading effect generated by the parasitic antenna element significantly
lowered both resonant frequencies. Thus, it can be used for downsizing the
antenna at the expense of a narrower impedance bandwidth. A radiating resonance
is excited on the parasitic antenna element. The resonant frequency is dependent
mainly on the element size. This resonance might be used for covering some
specific frequency band or used as an adjacent resonance to an already existing
band to improve bandwidth. This kind of adjustment would be very difficult to
50
properly implement, since it would have a significant effect on the frequency
tuning of the other bands.
In paper III, a combination of these two tuning methods was used to switch a
dual-band PIFA to operate in eight frequency bands (including one overlapped
band). The antenna covered 760–1075 MHz and 1530–2170 MHz effective
bandwidths with at least 3 dB return loss margins. The five cellular bands fall in
this range with a clear margin. This indicates an existing potential for downsizing
the antenna with a maintained operation in the cellular bands. Alternatively, the
covered bandwidth might be used for compensating detuning caused by the
proximity of a user or situations that occur when opening or closing a device of
clamshell or swivel type.
A valid performance comparison of different tuning methods for PIFAs is
difficult, because quite a limited amount of efficiency data is available. It should
also be noted, that in order to perform a useful frequency-tuning operation, an
antenna’s total efficiencies within the bands should be clearly separated. No
significant overlapping should occur. Nevertheless, by using switchable tuning
stubs, radiation efficiencies between 80–95% have been achieved (Kivekäs et al.
2002, Ollikainen et al. 2002). By using switchable feed position and switchable
ground connection, peak total efficiencies were achieved in the range of 40–60%
and 60–80% for the lower and upper GSM bands, respectively (Mak et al. 2007a).
Also, significant 50–60% drops in radiation efficiency, caused by the tuning
circuits, have been reported for different tuning methods (Louhos & Pankinaho
1999).
Frequency-tunable planar monopole antennas have not been researched as
much as PIFAs. An RF PIN diode tunable dual-band planar monopole operating
at both upper and lower cellular bands has been presented (Thornell-Pers &
Erlandsson 2007). The antenna consisted of two resonating strips. The tuning
circuit is arranged at the antenna’s feed in such a manner that the length of both
strips can vary with a single switch and the DC control voltage is supplied with
the RF signal, similarly as in paper V. The antenna provided penta-band operation
with peak total efficiency values of ~ 60% in the lower bands and 65–80% in the
upper bands.
DVB-H antennas in terminal devices are used for signal reception only.
Owing to its multiple narrow channels (8 MHz) over a wide frequency range at
rather low frequencies (470–702 MHz) and rather low gain specifications (-10 to -
7 dBi), a frequency-tunable antenna makes a very attractive solution for this
application. The reception antenna deals with inherently lower power levels as
51
compared with transmitter antennas, thus the requirements for the power handling
capability and linearity of the active tuning circuit are obviously lower than for
the ones used in transmitters. The antenna presented in paper IV is compact. The
antenna element occupies only a 0.7 cm3 volume. The antenna elements of both
frequency-tunable antennas and passive antennas proposed for DVB-H signal
reception are at least slightly larger than 0.7 cm3. However, in many of these
antennas, the BW and gain specification are fulfilled with clearer margins (Choi
et al. 2006, Holopainen et al. 2006, Wong et al. 2006, Milosavljevic 2007).
The frequency-tunable PIFAs in papers I–III showed equal radiation
efficiency deterioration compared with the corresponding reference antennas in
both the ON and OFF states of the switches. However, the planar monopole
antenna operating at the same frequency ranges in paper V showed a clear
radiation efficiency drop in the lower band between the switched ON (max. gain
0.9 dBi) and OFF (max. gain 2.2 dBi) states. This indicates that the RF PIN diode
switch exhibited an equal amount of loss for PIFAs in both states, whereas in the
planar monopole antenna, the amount of loss is significantly higher in switched
the ON state than in the switched OFF state.
General drawbacks for PIN diode switches are a reasonably high ON state
DC power consumption. The main reason for the use of diodes was a very limited
availability of commercial MEMS switches. More or less successful tunable-
antenna experiments, using RF MEMS switches have been reported (Panaia et al.
2004, Anagnostou et al. 2006, Boyle & Steeneken 2007b, Jae-Hyoung Park et al.
2007). In these experiments mostly unpackaged in-house prototype components
are used. Mak et al. (2007a, 2007b) compared a RF PIN diode switch, single pole
double throw (SPDT) GaAs switch and SPDT RF MEMS switch through an
implementation in two dual-band frequency-tunable antenna structures operating
at the cellular frequencies. Both antenna efficiency and high power properties
were studied. No significant differences between the switches were reported. In
fact, both papers concluded that the RF PIN diode switch was the most preferred
option for these applications. It should be emphasized that the purpose of these
studies was to compare three individual components, not the aforementioned
technologies. Nonetheless, these papers provide very useful information.
Essential system-level requirements, such as linearity, switching speed,
reliability, cost and front-end compatibility have to be also considered when
dealing with electrically tunable antennas. All of these issues are highly
dependent on the system specifications and the properties of the selected active
tuning circuit components.
52
53
3 Ceramic package integrated microwave antennas
Among the most often used constituent of SiP platforms for microwave and
millimeter-wave wireless communication applications is Low Temperature Co-
fired Ceramics (LTCC). The basic building block of the LTCC technology is a
ceramic green sheet. The process flow begins with via drilling and filling.
Individual sheets are then patterned with conductor, dielectric and resistive pastes
using the screen printing technique. The patterned sheets are stacked and
laminated. Following process steps are co-firing, inspecting and dicing. After co-
firing, the green sheets form a monolithic ceramic structure. The sintering
temperature of most of the material systems is near 900 °C enabling the use of
highly conductive metals, such as silver and gold as conductors.
The screen printing technology associated with the standard LTCC process
has traditionally been used to integrate various types of common passive
components, such as resistors, capacitors, and inductors, into packages. Critical
system-level microwave components such as power dividers, baluns, filters, and
antennas, can also be integrated into this type of structure (Heyen et al. 2003,
Seki et al. 2005, Jong-Hoon Lee et al. 2005a, Jong-Hoon Lee et al. 2005, Young
Chul Lee et al. 2005 b, Byoung Hwa Lee et al. 2006, Kangasvieri et al. 2006,
Dong Yun Jung et al. 2007). The typical way to integrate an antenna is to screen
print a microstrip antenna, or antenna array, onto the topmost layer of a multilayer
LTCC package. Besides these highly integrated packages, LTCC components and
sub-system packages are used in a traditional packaging approach, in which
passive components, single packaged ICs and sub-system packages are mounted
and reflow-soldered onto a common system board (PCB) using standard surface
mount device (SMD) assembly processes (Oppermann 2001, van Heijningen &
Gauthier 2004, Kangasvieri et al. 2007).
Most commercial LTCC green tape systems have a moderate relative
permittivity in the range of 5 to 20 (Sebastian & Jantunen 2008). Since the
substrate wavelength is proportional to εr-0.5, the use of ceramic substrate material
will reduce the antenna size when compared with implementing the antenna on
PCB or air dielectrics. However, at the same time, the radiation properties of
microstrip antennas, such as impedance BW and radiation efficiency are
adversely affected as the permittivity of the substrate increases (Pozar 1992).
54
3.1 Embedded air cavity backed microstrip antenna on an LTCC substrate
The effective dielectric constant seen by an antenna can be reduced by fabricating
an embedded air cavity in the substrate, beneath the radiating element [VI]
(Panther et al. 2005). This could be a particularly useful solution for antennas in
monolithic ceramic media operating at higher microwave and millimeter-wave
frequencies, where the resonant type antennas will naturally have a small physical
size. For such structures, the BW and ηrad could be traded for antenna size, which
would still retain dimensions of the order of millimetres. In this way, a multilayer
package could still contain integrated passive components that utilize the
moderate relative permittivity of the ceramic substrate, while providing a low
effective permittivity in the locality of the antenna radiating elements. A similar
technique is also adopted for improving the radiation properties of on-chip
antennas (Papapolymerou et al. 1998, Ojefors et al. 2007).
The rectangular microstrip patch antenna studied here incorporating an air
cavity in the substrate is illustrated in Fig. 23. An aperture coupled microstrip
antenna utilizes three layers of metallization and two dielectric substrates, making
it feasible for implementation using multilayer technology. A microstrip
transmission line on the bottom of the lower substrate is electromagnetically
coupled through an aperture in the common ground plane to the patch on top of
the upper substrate. The feed line is terminated by a quarter-wavelength open-
ended stub to maximize the amount of power coupled to the antenna.
Two aperture coupled microstrip patch antennas were designed to operate in
the fundamental TM01 mode at 10 GHz. To form a uniform monolithic structure, a
1 mm (0.03 λ0) thick DuPont 951 with material parameters εr = 7.8, and tan δ =
0.006 at 3 GHz, was selected as the material for both the upper and lower
substrates for both antennas. At 10 GHz εr can be assumed to be about the same,
and tan δ slightly higher than the values given by the manufacturer at 3 GHz. In
the first antenna, an air cavity with a height of 0.5 mm was positioned at the
bottom of the upper substrate. The second antenna without the air cavity was
fabricated on a bulk ceramic substrate, which resulted in similar geometrical
dimensions except for Px = 6.5, Py = 4.1, Ax = 3.5, and Ay = 0.4 (cf. fig. 23). The
fabricated cavity antenna was an assembly of two fired monolithic substrates
glued together. Altogether, the antenna was formed of 10 layers of DP951AX tape,
and the final thickness of the assembly was 2060 μm. A cross-section of the
antenna is shown in Fig. 24.
55
Fig. 23. Aperture coupled microstrip with an embedded air cavity. Px = 10.5, Py = 7, Cx =
14, Cy = 11, Ax = 5, Ay = 0.6, Lx = Ly =Ux = Uy = Gx = Gy = 30, Uz = Lz = 1, and Mx = 1.1, all
in mm. [Paper VI]
Fig. 24. A cross section image of the antenna showing the laser-micromachined cavity.
[Paper VI]
The measured return loss of the cavity antenna and the reference antenna are
presented in Fig. 25. The cavity antenna showed a 10% upward frequency shift to
an intended 10 GHz operation frequency. The shift was probably caused by the
somewhat cambered cavity roof on the electrically measured samples. The
measured 10 dB return loss BWs of the cavity antenna and the reference antenna
were 6.8% and 3.2%, respectively.
The measured far field radiation patterns (Fig. 26) showed maximum gains at
the corresponding center frequencies of 5.4 dBi for the cavity antenna and 3.2 dBi
for the bulk ceramic antenna. The significantly higher maximum gain value of the
cavity antenna suggests that the use of the cavity also improves radiation
efficiency by decreasing the dielectric losses and the radiation quality factor.
However, it should be noted that the radiation patterns (directivities) of the
antennas were not notably different and the amount of the radiation efficiency
increase cannot be accurately estimated from the maximum gain values.
56
Fig. 25. Measured return losses of aperture coupled microstrip antennas with and
without an embedded air cavity. [Paper VI]
Fig. 26. (A) Measured E-plane (YZ-plane) and (B) H-plane (XZ-plane) radiation patterns
for the antenna with the embedded air cavity. (C) Measured E-plane (YZ-plane) and (D)
H-plane (XZ-plane) radiation patterns for the bulk ceramic antenna. [Paper VI]
57
3.2 Surface-mountable LTCC-BGA antenna package
Paper VII described the design, modelling and implementation of a compact
surface mountable antenna package for applications in X-band wireless
communications. The passive antenna package consists of a stacked patch
microstrip antenna and a wideband vertical BGA-via transition path starting from
an organic motherboard and ending at the antenna element in the LTCC package.
A stacked patch configuration was selected to enhance the impedance BW of
a microstrip antenna (Waterhouse 2003) compared with a similar single patch
implementation. This is based on an excitation of two adjacent resonances. The
antenna here (Fig. 27a) consists of a driven lower patch embedded in the LTCC
and a parasitically coupled upper patch on the top of the package. The antenna is
fed by a vertical via transition, which connects a 50 Ω grounded co-planar
waveguide transmission line (GCPW) on the bottom surface of the package to the
lower patch through a circular opening in the ground plane.
The design and optimization of this kind of antenna is fairly straightforward
(Li et al. 2004). The feed position can be selected at the edge of the lower patch.
Both square patches share an equal size, which is half of the guided wavelength
(λg /2) at 10 GHz, and the same horizontal alignment. The design is mainly done
by optimizing the coupling between the patches. The vertical distance between
the ground plane and the lower patch (h1) is fixed and the distance between the
lower and upper patches (h2) is adjusted by selecting proper dielectric tape counts
and thicknesses. The commercial EM software Ansoft HFFS was used to find the
thicknesses, h1= 0.2 mm and h2 = 0.5 mm, which optimize 10-dB (VSWR=1.92)
BW.
Fig. 27b shows the equivalent circuit model of the GCPW-via transition-fed
stacked patch microstrip antenna. The circuit model has two parallel resonant
circuits, which describe the lower patch (L1, C1, and R1) and the upper parasitic
patch (L2, C2, and R2). The electromagnetic coupling between the patches is
modelled with a coupling capacitance (Cc) and a mutual inductance (M) (Anguera
et al. 2001, Li et al. 2004). Lp1 and Lp2 represent the distributed inductance and Rp
the resistance associated with the feeding via. Co and Cpad describe the
capacitances of the circular ground plane opening and a pad at the lower end of
the via (Wang et al. 2005). Finally, a universal transmission line block (Z0,
attenuation, phase) is used to model the GCPW line on the bottom surface of the
substrate. This block is not necessarily required in the antenna’s circuit model, but
it was used to link the presented circuit model with a model of a BGA transition.
58
Fig. 27. (a) Cross-sectional view of the GCPW-via transition-fed stacked patch
microstrip antenna; (b) circuit model of the structure. The GCPW line and gap widths
are 0.44 and 0.203. Dimensions are in mm. [Paper VII]
First-order approximations of the circuit model component values of the lower
patch, assuming an absence of the upper patch, were calculated using the
equations given in (Abboud et al. 1988, Wang et al. 2005). These values, L1=58
pH, C1=4.35 pF, R1=215 Ω, fr1=10 GHz (fr =1/2п√LC), Q1=59 (Q= 2пfrRC), and
Lp2=97 pH, were used as a starting point for optimization of the circuit model
component values. The subsequent fitting of the model response to the EM-
simulated S-parameter data was done with a RF circuit simulator (Ansoft
Designer). The best correspondence between the responses of the circuit model
and the EM simulation for the antenna structure was found with the values; Lp1=
225 pH, Lp2=90 pH, Rp=1 Ω, Co=75 fF, Cpad=30 fF, Cc=135 fF, M=3.6 pH, L1= 49
pH, C1= 5 pF, R1= 170 Ω, L2= 50 pH, C2= 4.85 pF, and R2=110 Ω. The
parameters for the transmission line block were defined to be 50 Ω, 0.15 dB/cm
and 45 deg. These values correspond to a physical line length of 1.75 mm at 10
GHz. In addition, resonant frequencies and quality factors, fr1=10.17 GHz, fr2=
10.22 GHz, Q1=54.3, and Q2=34.3 are thus obtained.
The EM-simulated and circuit-modelled input impedance loci of the GCPW-
via transition-fed stacked patch microstrip antenna are presented in Fig. 28.
59
Fig. 28. EM-simulated and circuit-modelled input impedance loci of the GCPW-via
transition-fed stacked patch microstrip antenna. [Paper VII]
It can be seen that the correspondence is good. The constant VSWR=1.92 circle
on the Smith chart shows that the BW is nearly optimized. In optimal BW
response, the locus should be at the center of the chart and as large as possible,
yet remaining inside the constant VSWR circle (Ollikainen & Vainikainen 2002).
The optimal relative bandwidth (BWopt) of dual-resonant patch antennas can be
estimated using Equation 5 (Ollikainen & Vainikainen 2002).
2
2212
12
22 111
4
11
QQQQS
SSBWopt ++⋅−−= (5)
Here, S denotes the maximum allowed VSWR value. By substituting the Q values
of the circuit model and S=1.92, a fractional BW of 0.062 is obtained.
Corresponding values obtained with the EM simulations and circuit model are
0.052 and 0.053, respectively. This proves that the deductions based on the
impedance loci on the Smith chart, as well as the magnitude of the circuit model
quality factors, are consistent. Finally, it is worth mentioning that the main factor
limiting the BW optimization of an antenna of this kind is that the dielectric tape
thicknesses usually have 100 μm increments, and thus, cannot be arbitrary
selected when dealing with commercial LTCC materials.
The fabricated and measured antenna package showed a 10 dB BW of 5.8%.
However, the measured return loss slightly exceeded the 10 dB margin, being 9
60
dB in the middle of the band. The radiation patterns at the two resonant
frequencies were almost identical and typical for a microstrip patch antenna. The
maximum gain of the antenna was about 4 dBi and the cross polarization level
was about 15 dB below the co-polarization level. It should also be noted that the
gain also accounts for the losses of the whole signal path of the package (~ 0.5
dB).
The impedance BW optimized stacked patch microstrip antenna presented in
this section was part of a surface mountable package assembly. The circuit model
with appropriate component values provided deeper insight into the electrical
behaviour of the modular structure and might also be used in the co-design of
antenna packages and complete transceiver systems with, e.g., SPICE-based
circuit simulation tools. A valid comparison between the aperture coupled
microstrip patches used in embedded air cavity experiments and the surface
mountable antenna package cannot be done. In spite of the same materials being
used, the horizontal sizes, the vertical profiles of the antennas as well as the
feeding techniques are different. Panther et al. (2005) showed that by using an
embedded air cavity between the patches of a stacked patch microstrip antenna, a
10 dB impedance BW close to 20% at 30 GHz can be achieved.
3.3 Integrated BST thick film varactor-tuned folded slot antenna
Barium Strontium Titanate (BST) based materials belongs to a well-studied non-
linear dielectric ceramics, which polarization properties, and thus their relative
permittivity, can be adjusted by a static external electric field. The BST materials
have two main phases, ferroelectric phase and paraelectric phase. The phase of
the material is depended on its characteristic Curie temperature (Tc), in which the
phase translation occurs. For RF- and microwave application BSTs are used in
their paraelectric phase (T > Tc). The reason for this is that in the ferroelectric
state (T < Tc) the material exhibits significantly higher loss because of its
hysteresis behaviour not existing in its paraelectric state (Gevorgian & Kollberg
2001).
BST in paraelectric state have found a wide range of applications in
tunable/reconfigurable RF and microwave components, such as filters, phase
shifters, varactors and frequency-selective surfaces (Gevorgian et al. 1998,
Miranda et al. 2000, Gevorgian & Kollberg 2001, Scheele et al. 2006, Scheele et
al. 2007). However, so far these materials have not been intensively used for
applications in small antennas. Jose et al. (1999) proposed a frequency-tunable
61
microstrip antenna based on BST substrates. In addition, a frequency-tunable
folded slot antenna based on a BST thin film layer has been proposed and
theoretically studied (Castro-Vilaro & Solis 2003). However, the antennas
presented lack a proper biasing and a characterization of radiation properties.
Our research group at the Microelectronics and Materials Physics
Laboratories has recently developed a low sintering temperature (~ 900 °C)
Barium Strontium titanate (BST) based thick film material (Tick et al. 2008a).
Paper VIII presents the dielectric properties of the BST thick films (εr, tan δ) in
the frequency range of 1–6 GHz and demonstrates the utilization of the material
as an integrated varactor in a frequency-tunable folded slot antenna.
The relative permittivity of the un-biased BST film was about 400 across the
frequency range, with 25% tunability for an electric field strength of 2.5 V/μm.
Loss tangents were 0.05 and 0.03 at 3.5 GHz, corresponding to the electric field
strengths of 0 and 2.5 V/μm. It was also shown that a ~ 40% tunability could be
obtained by increasing the bias field to 10 V/μm.
To demonstrate the use of BST thick films in frequency-tunable antennas, the
concept proposed by Holland et al. (2006) was adopted. In this approach, the
resonant frequency of a slot/co-planar patch antenna is tuned using a discrete
varactor component which is placed over the radiating slot. The structure of a
frequency-tunable folded slot antenna is depicted in Fig. 29. It consists of a FSA
operating near 3 GHz, which is designed on an alumina substrate (εr = 9.4), and
an integrated BST varactor. Simulation results showed varactor capacitance
values of 1.5 pF and 1.05 pF with BST layer relative permittivies of 400 and 270,
respectively.
62
Fig. 29. Schematic layout of (a) a folded slot antenna loaded with a BST varactor (b)
Side cross-sectional view of the BST varactor (c) Top view of the BST varactor (all
dimensions are in mm). [Paper VIII]
The tunable antenna and a reference antenna excluding the integrated varactor
were fabricated. An external coaxial bias T component was used to supply DC
bias voltage across the varactor with the RF signal. Far-field radiation patterns
were measured in an anechoic chamber using the same DC voltages. A Satimo
Starlab system was used for the total antenna efficiency measurements. However,
the DC bias was not used in the efficiency measurement.
The measured return losses of the antennas are presented in Fig. 30. The
center operating frequency of the unbiased tunable antenna is 3.18 GHz. When
200 V is applied, the center frequency shifts to 3.29 GHz (3.5%). The 10 dB BWs
are ~ 4% for all bias voltages. The reference antenna shows a center frequency of
3.73 GHz and a 5.5% 10 dB BW. Thus, the resonant frequency and BW of the
unbiased tunable antenna are 0.55 GHz and 1.5 % lower compared with those of
the reference antenna. The measured maximum total efficiencies (not graphically
presented) of the unbiased tunable antenna and the reference antenna were 44%
and 75%, respectively. Both antennas are well impedance-matched (peak return
loss better than 17 dB), thus, the maximum radiation efficiencies are nearly equal
to the maximum total efficiencies. Radiation efficiency deterioration caused by
the integrated varactor is about 30%. Slightly higher efficiencies may be expected
with increased DC voltage, since the εr and tan δ of the BST material decrease at
the same time.
63
Fig. 30. Measured return losses of the integrated BST varactor-tuned FSA and the
reference FSA without the varactor. (Paper VIII).
The measured far-field radiation patterns of the tunable antenna and reference
antenna are presented in Fig. 31. The patterns are near omni-directional in the XY
plane and bidirectional in the XZ plane. The pattern shapes of the tunable antenna
and the reference antenna are almost identical. The peak gain values of the
antennas are 1.25 dBi (0V), 1.85 dBi (200V) and 3.7 dBi (reference antenna). The
presence and use of the varactor does not substantially change the shape of the
radiation pattern, but the efficiency deterioration is seen as a decrease in gain
levels. Similar effects were also obtained in (Holland et al. 2006), where a
discrete semiconductor varactor was used.
64
Fig. 31. Measured far-field radiation patterns of the BST varactor-tuned FSA and the
reference FSA without the varactor; a) in the XZ plane; (b) in the XY plane; (c) in the YZ
plane. [Paper VIII]
3.4 Discussions
The 10 dB impedance bandwidth of an aperture coupled microstrip antenna
operating near 10 GHz was increased from 3.2% to 6.8% by fabricating an
embedded air cavity beneath the antenna. The cavity antenna still maintained a
low vertical profile (1 mm) and a reasonable small horizontal size (7 mm x 10.5
mm) for SoP assemblies. The work presented by Panther et. al. in (Panther et al.
2005) further validates the usefulness of the idea of using a substrate embedded
air cavity.
Laser-micromachining was chosen for the cavity preparation, since it enabled
cavities of arbitrary shapes and depths to be formed independently of the tape
thickness, while still providing a quick turnaround time between designs. The
cavity was laser-machined after lamination, thus avoiding the complicated
processing steps normally faced when laminating a green body with cavities
(Panther et al. 2005, Vaed et al. 2004). The antenna with the embedded air cavity
65
was an assembly of two separately fired ceramic pieces. The laser micro-
machining has later been used for fabricating truly monolithic co-fired LTCC
substrate integrated waveguides operating at frequencies up to 150 GHz (Tick et
al. 2008a). It is evident that the advantages of using the embedded cavities will be
further emphasized if the operation frequencies of the components increase.
Low sintering temperature BST thick film material contains several beneficial
features. LTCC compatibility of the material has been proven by using the BST
paste as a via filling material to form a substrate embedded varactor (Tick et al.
2008c) and co-firing the material inside a LTCC substrate using pressure aided
sintering (Tick et al. 2008d). In this manner, the functionality of LTCC packages
can be increased. In the experiments, over 34% tunability of εr for an electric field
gradient of 4 V/μm electric field has been reported.
Although the tunability of the fabricated folded slot antenna with an
integrated BST varactor was only 3.5%, and despite the radiation efficiency
deterioration caused by the varactor (31%), the results give an idea of the key
advantages of this technology: Compatibility with silver metallization, ability to
produce 25 µm thin films, and potential for low-cost volume production.
Altogether, this is believed to be one of the very first frequency-tunable antennas
based on an integrated paraelectric material, including proper biasing and
characterization of the radiation properties.
However, there are some conceivable drawbacks with the use of non-linear
BST material in frequency-tunable antennas. A typical electric field gradient
required to adjust the permittivity up to 50% is in the range of 2–6 V/μm, thus,
reasonably high DC voltages are required for appropriated tuning when dealing
with thick films. Furthermore, moderate tunability, moderate to high dielectric
losses and a high temperature dependence of dielectric properties should also be
accounted for. However, the material might be used for some other applications,
e.g., sensors in low-cost highly integrated multifunctional ceramic packages.
66
67
4 Conclusions
The main target of this thesis has been to develop and research bandwidth
enhanced antennas for mobile terminal and multilayer ceramic packages. The
research methods included the design, implementation and measurement of small
antennas. Literature surveys were given for each sub-topic of the thesis, and the
results were discussed with respect to the relevant data available in the open
literature.
The first part of the thesis was devoted to frequency-tunable mobile terminal
antennas based on discrete semiconductor components. The essential merits of the
work and the key results are concluded as follows:
– A novel frequency-tuning method for planar inverted-F antennas was
developed. In this method, the tuning circuit was built into the antenna
element by using discrete reactive passive components to form separate paths
for the RF signal and the DC voltage, the latter being used to control the state
of the active component. The DC voltage was applied to the antenna with the
RF signal, and the tuning circuit was completely integrated. The effectiveness
of the tuning method was demonstrated with a single-feed dual-band PIFA
tuned with a RF PIN diode switch to operate in four frequency bands. The
antenna was built on 45 mm x 110 mm grounded PCB and it covered the
frequency ranges of the GSM1800, PCS1900 and UMTS telecommunication
standards with a 65% total antenna efficiency margin. The total efficiency
coverage of the GSM900 and GSM850 bands were 50% and 40%,
respectively. The impact of the tuning circuit on the antenna’s radiation
pattern and efficiency was investigated by fabricating and measuring a
reference antenna without the tuning circuit. It was shown that the tuning
circuit reduces the radiation efficiency by less than 20% in the lower bands
and less than 10% in the upper bands. The radiation patterns of the lower
frequency bands remained essentially unchanged, despite the presence and
use of the tuning circuit. However, in the upper band, a slight tilt in the
pattern occurred when the antenna was tuned.
– A frequency-tunable dual-band planar inverted-F antenna based on a
switchable parasitic antenna element was presented. The antenna was built on
40 mm x 105 mm grounded PCB. The antenna covered the frequency ranges
of the GSM850 and GSM900 telecommunication standards with a 25% total
efficiency margin. The total efficiency coverage of the GSM1800, GSM1900,
68
and UMTS were 35%, 65% and 40%, respectively. The impact of the
capacitive loading effect of the parasitic antenna element was experimentally
studied by comparison with a reference antenna that did not have the parasitic
element. The results proved that the effect shifted the resonant frequencies
clearly downwards, which can be used to miniaturize the antenna’s size.
Moreover, it was shown that the loading effect narrows the impedance BWs,
especially when the parasitic element was switched to the ground potential.
The decrease in radiation efficiency caused by the presence and the switching
of the parasitic element was found to be less than 20% in the lower bands and
less than 10% in the upper bands. It was also found that frequency tuning
does not substantially change the shape of the radiation patterns.
– A frequency-tunable dual-band planar inverted-F antenna capable of
operating in eight frequency bands was implemented. Two separately
controllable tuning circuits consisting of three RF PIN diode switches and
four discrete passive components were built on the main antenna element and
a parasitic antenna element, which was situated between the main antenna
element and the ground plane. By changing the state of the switches, four
different operating modes, resulting in altogether eight different frequency
bands for the antenna, were created. The antenna built on 45 mm x 110 mm
PCB covered the 750-1080 MHz band with 25% and the 1560-2270 MHz
with 50% total efficiency margins. It was shown that the tuning circuit
reduces the radiation efficiency by less than 35% in the lower bands and less
than 15% in the upper bands, compared with a reference antenna not
incorporating any tuning circuitry. Slight variations were observed in the
radiation patterns, especially for the upper frequency bands.
– A varactor-tuned meander line monopole antenna for digital television signal
reception was presented. The antenna has omni-directional radiation pattern
and covers the frequency range of the DVB-H standard with requisite
impedance bandwidth and gain level. The main advantage is that the antenna
element occupies a very small volume (0.7 cm3) and can be used in 40 mm x
100 mm x 10 mm devices.
– A frequency-reconfigurable dual-band monopole antenna covering the
GSM850, GSM900, GSM1800 and GSM1900 bands within a 3 dB
impedance bandwidth was presented. The antenna was built on 40 mm x 100
mm PCB. The lower band of the antenna could be switched between the two
lower cellular bands with a RF PIN diode switch, whereas the upper band
inherently covered the two upper bands. The tuning circuit was integrated
69
directly into the antenna structure with no need for separate DC wiring. The
antenna provided maximum gain values of 2.2 dBi, 0.9 dBi, 3 dBi and 3 dBi,
corresponding with the GSM850, GSM900, GSM1800, and GSM1900 bands,
respectively. The antenna presented was one of very few frequency-tunable
planar monopole antennas operating at practical frequency bands presented in
the literature.
The second part of the thesis focused on multilayer ceramic package integrated
antennas. The essential merits of the work and the key results are concluded as
follows:
– A method to enable a size to impedance BW trade-off of a microwave
microstrip patch antenna on multilayer ceramic substrate using an embedded
air cavity beneath the radiating element was investigated. This would be a
particularly useful solution for antennas in higher microwave and millimeter-
wave system in packages (SiPs) where the resonant type antennas will
naturally have a small physical size. In this way, a multilayer package could
still contain integrated passive components that utilize the moderate relative
permittivity of the ceramic substrate. Laser-micromachining was chosen for
the cavity preparation, since it enabled cavities of arbitrary shapes and depths
to be formed independently of the LTCC tape thickness, while still providing
a fast turnaround time between designs. The measured BWs of aperture
coupled rectangular microstrip patch antennas on a LTCC substrate, operating
at near 10 GHz with (7.5 mm x 11 mm x 1mm) and without the air cavity (4.1
mm x 6.5 mm x 1 mm) were 6.8 % and 3.2 %, respectively. The measured far
field radiation patterns showed maximum gains at the center frequency of 5.4
dBi for the cavity antenna and 3.2 dBi for the bulk ceramic antenna.
– An implementation of a compact surface-mountable LTCC-BGA antenna
package feasible for use in X-band wireless transceiver systems was
described. The package assembly, operating near 10 GHz, was composed of
an electromagnetically shielded vertical BGA-via transition path starting
from an organic mother board and ending at the BW optimized stacked patch
microstrip antenna. A circuit model describing the antenna was developed to
gain a deeper insight into its electrical behaviour. The fabricated package was
12 mm x 12 mm x 1.4 mm in size and provided 5.8 % of 10 dB impedance
BW and 4 dBi maximum gain. In addition, the circuit model developed
agreed well with the EM-simulations, demonstrating the validity of the model.
70
– The properties of low-sintering-temperature BST thick films at the frequency
range of 1–6 GHz were reported and a frequency-tunable antenna
demonstrating this integration possibility was presented. In the range of 1–6
GHz, the film has a relative permittivity of 400 with 25% tunability using a
static electric field gradient of 2.5 V/μm. Depending on the biasing voltage,
the loss tangent value at 3.5 GHz is less than 0.05. The demonstrator selected
was a folded slot antenna operating near 3 GHz tuned with an integrated BST
varactor. Although the tunability of the fabricated folded slot antenna with an
integrated BST varactor was only 3.5%, and efficiency deterioration
compared to the reference antenna excluding the varactor was ~ 30%, the
results give an idea of the key advantages of this technology: compatibility
with silver metallization, the potential for LTCC compatibility, ability to
produce 25 μm thin films, and potential for low-cost volume production. This
is believed to be one of the very first frequency-tunable antennas based on an
integrated non-linear dielectric material including, a proper biasing and
characterization of radiation properties.
71
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Original papers
I Komulainen M, Berg M, Jantunen H, Salonen E & Free C (2008) A Frequency Tuning Method for a Planar Inverted-F Antenna. IEEE Transactions on Antennas & Propagation 56(4): 944–950.
II Komulainen M, Palukuru VK & Jantunen H (2007) Frequency-Tunable Dual-Band Planar Inverted-F Antenna Based on a Switchable Parasitic Antenna Element. Frequenz –Journal of RF engineering and telecommunications 61(9–10): 207–212.
III Komulainen M & Jantunen H (2008) Switching a dual-band planar inverted-F antenna to operate in eight frequency bands. Proceedings of Loughborough Antennas and Propagation Conference: 85–88.
IV Komulainen M, Berg M, Jantunen H & Salonen E (2007) Compact varactor-tuned meander line monopole antenna for DVB-H signal reception. IET Electronics Letters 43(24): 1324–1326.
V Komulainen M, Berg M, Palukuru VK, Jantunen H & Salonen E (2007) Frequency-Reconfigurable Dual-Band Monopole Antenna for Mobile Handsets. IEEE Antennas & Propagation International Symposium Digest: 3289–3292.
VI Komulainen M, Mähönen J, Tick T, Berg M, Jantunen H, Henry M, Free C & Salonen E (2007) Embedded air cavity backed microstrip antenna on an LTCC substrate. Journal of European Ceramic Society 27(8–9): 2881–2885.
VII Komulainen M, Kangasvieri T, Jäntti J & Jantunen H (2008) Compact Surface-Mountable LTCC-BGA Antenna Package for X-band Applications. International Journal of Microwave and Optical Technology 3(4): 451–459.
VIII Palukuru V K, Komulainen M, Tick T, Peräntie J & Jantunen H (2008) Low Sintering-Temperature Ferroelectric-Thick films: RF Properties and an Application in a Frequency-Tunable Folded Slot Antenna. IEEE Antennas and Wireless Propagation Letters 7: 461–464.
Reprinted with permission from IEEE (I, III, IV, V and VIII), Frequenz (II),
Elsevier B.V. (VI) and ISRAMT/IJMOT (VII).
Original publications are not included in the electronic version of the disserdation.
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