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This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIA.2017.2682793, IEEETransactions on Industry Applications

Bidirectional Current-Fed-Half-Bridge (C)(LC) –

(LC) Configuration for Inductive Wireless Power

Transfer System

Suvendu Samanta, Student Member IEEE, Akshay K. Rathore, Senior Member IEEE,

and Duleepa J. Thrimawithana, Member IEEE

Abstract -- This paper contributes to analysis and development

of a new power electronics system for bidirectional wireless power

transfer. The major focus is analysis and implementation of a new

current-fed resonant topology with current-sharing and voltage

doubling features. A new bidirectional wireless power transfer

system with current-fed half-bridge voltage doubler circuit is

proposed and analyzed with series-parallel and series resonant

networks. Traditionally used parallel L-C resonant tank in

transmitter circuit with current-fed WPT topology causes higher

voltage stress across the inverter devices to compensate the

reactive power consumed by the loosely coupled coil. In the

proposed topology, this is mitigated by adding a suitably designed

capacitor in series with the transmitter coil thus developing a

series-parallel CLC tank. Detailed analysis and design is reported

for both, grid-to-vehicle and vehicle-to-grid operations. The

power flow is controlled through variable frequency modulation.

Soft-switching of the devices is obtained irrespective of the load

current. A proof-of-concept experimental hardware prototype

rated at 1.2kW is developed and tested. Experimental results are

presented to verify the analysis and demonstrate the performance

of the system with bidirectional power flow.

Index Terms—Wireless power transfer, Voltage doubler,

Current-fed converter

I. INTRODUCTION

ireless power transfer (WPT) systems are capable of

transferring power over a large distance without any physical

contact. Various WPT applications are electric vehicles (EV)

[1]-[20], electronic gadgets, lighting, material handling and

biomedical implants. Capacitive WPT is usually implemented

for low power applications whereas the wireless inductive

power transfer (IPT) technology has been developed for both

low and medium power applications [8], [21].

Recently, research and development on IPT technology

for EV battery recharging application has grown tremendously

[1]. The IPT technology promises a convenient and safer way

of recharging the EV batteries. Ongoing research has proved

that the opportunistic charging with IPT technology reduces

the requirement on battery capacity significantly for a given

travel distance [8], [11] and is promising for local

transportation. A typical Inductive WPT system is shown in

Fig. 1. Generally, the front-end grid side ac-dc stage is chosen

as boost-type power factor correction (PFC) rectifier. For a

mid-range (100-300 mm) WPT, the IPT frequency is generally

selected around 20 kHz to 150 kHz to balance the converter

size, efficiency and cost [10], [19]. Voltage source inverter

(VSI) topologies in the inversion stage are commonly used and

reported in literature mainly because the ac-dc PFC output is a

voltage source [1]-[3]-[5], [11]. To reduce the number of

power conversion stages, direct ac-ac converters have also

been reported in [14].

AC-DC

Active

Rectifier

AC-DC

Coverter

200-300mm Air-gap

Transmitter

Coil

Receiver

Coil

120/230V

60/50Hz

Battery

Bank

On-BoardOff-Board

DC-AC

Convert

-er

Compensat

-ion/

Resonance

Network

Compensat

-ion/

Resonance

Network

Fig. 1. General IPT power conversion stages for EV battery charging

To compensate the reactive power in IPT system, the

simple method is to connect capacitor in both transmitter coil

(TC) and receiver coil (RC) circuits. It develops four types of

compensation networks such as series-series, series-parallel,

parallel-series and parallel-parallel. Transmitter side series

compensation is quite common when the inversion stage is

VSI topology [1]-[3], [11], [16], [21]. In addition, VSI

topologies with LCL [4], [10], [13] and CLCL [5], [10]

compensation networks have also been used for improved

power factor and better performance.

Transmitter side parallel compensation with current

source inverter (CSI) topologies is reported [12], [13]. The

merit of parallel resonant tank is that the capacitor provides the

required reactive power to the coil without flowing through the

inverter switches. In addition, the parallel capacitor provides

much lower impedance to the higher order harmonics and

hence, the coil voltage and current profiles are almost

harmonics free. However, at medium power level, the

requirement of higher voltage rated inverter devices is a major

limitation of transmitter side parallel LC tank. This is because

the parallel capacitor alone provides high volume of reactive

power consumed by the TC. To overcome this issue, in this

paper, a new IPT topology with current-fed converter is

proposed and analyzed. A capacitor is added in series with the

TC to develop CLC tank that reduce the voltage stress across

the inverter switches. Proposed IPT topology is capable of

conducting bidirectional power, thus enabling both grid-to-

vehicle (G2V) and vehicle-to-grid (V2G) operations.

This is the first attempt to implement bidirectional IPT

with current-fed topology with current-sharing voltage doubler

configuration [22]. This is an enhanced version of the paper

presented in [22] with additional results and detailed analysis

and design. DC link inductor provides natural short circuit

protection and also limits the peak and circulating current

through the components. Current sharing (half-bridge)

configuration further reduces the average and peak current

through the components resulting into reduced conduction

losses. Current-fed circuit also offers voltage gain and the

voltage doubler add 2x additional gain. Proposed converter is

analyzed and detailed design procedure is reported.

W



It should be noted that the focus and contribution of this

research is on power electronics, not on the coils or magnetics

design. The objective of this paper is to analyze a bidirectional

current-sharing voltage doubler current-fed topology with new

CLC series-parallel resonant network. The objectives are

realized in various sections. Section II introduces the proposed

bidirectional topology and explains detailed G2Voperation.

Section III studies V2G operation in detail. Systematic design

and components’ selection of the converter are presented in

Section IV. Section V demonstrates experimental results for

both G2V and V2G operations.

II. PROPOSED BIDIRECTIONAL IPT TOPOLOGY

Fig. 2 shows the proposed bidirectional IPT topology where

the transmitter side high-frequency (HF) inverter is current-fed

half-bridge and receiver side HF rectifier is a voltage doubler.

During G2V operation, i.e., EV battery recharging operation,

the switches S1 and S2 are modulated while the switches S3

and S4 are kept permanently off. Compared with conventional

parallel LC tank network, an extra capacitor is added in series

with the TC to reduce the effect of high leakage of the coil to

form CLC series-parallel resonant circuit. The detailed

comparison of parallel LC and CLC tank network with the

inverter as half bridge and full-bridge current-fed converter is

presented in [23].

Transmitter Receiver

Off-Board

S2S1

Cp

vd

id

Ld1L1 i1

jωMi2

v1

Cs

vsvi

S3 S4

Ld2

ii

On-Board

vo

io

Co1

C2L2i2

jωMi1

v2

S5

S6

Co2

vr

Fig. 2. Bidirectional WPT topology using current-fed half bridge converter.

An appropriate size capacitor is connected in series with

receiver coil to compensate the reactive power consumed by

the receiver coil. The series compensation network is most

common in IPT application due to simple structure and load

independent resonance. However, series compensation both in

the transmitter and receiver sides leads to insatiably during no

load condition. However, in this paper only the receiver side

compensation is series LC type; therefore, this instability issue

does not arise here. The detail design considerations of series-

series compensated IPT topology are elaborately described in

[24]-[29]. During G2V operation the voltage doubler network

is used as an uncontrolled rectifier whereas during V2G

operation, this converter acts as an inverter. The voltage across

and current through the switches are named as vS1 ~ vS6 and iS1

~ iS6, respectively and these signals for their body diodes are

named as vD1 ~ vD6 and iD1 ~ iD6, respectively.

A) Steady state operation of G2V

To explain the steady-state operation of the proposed

converter, the switches S1 and S2 are operating at fixed duty

cycle and power is controlled by variable frequency

modulation. Ideally the duty cycle of S1 and S2 are 0.5 and

their gating signals are complimentary. However, to make sure

the continuity of stiff dc link current, Id, a slight overlap in

gating signals of S1 and S2 is given. The operating power

factor at the output of the half bridge current fed inverter is

considered to be lagging to achieve zero-voltage switching

(ZVS) at device turn-on. However, soft-switching at device

turn-off is also possible if this power factor at inverter output is

leading. The steady state analysis, operating waveforms and

equivalent circuits for G2V operation of this converter are

reported in [30]. This paper reports steady state operation and

analysis for V2G operation and it is discussed next.

B) Steady state operation of V2G

During V2G operation the vehicle side converter acts as a

voltage-fed half-bridge inverter and grid side converter acts as

a current-doubler rectifier. To achieve soft switching of the

inverter switches irrespective load, the duty cycle of vehicle

side converter is kept fixed to 0.5 and power flow is controlled

by varying switching frequency. A slight dead band is always

maintained between switches S5 and S6 such that the dc bus

voltage, vo never gets shorted. Grid side current fed converter

switches S3 and S4 are kept on permanently and body diodes

D1 and D2 act as rectifier diodes. Fig. 3 shows the V2G

operating waveforms during steady state and Fig. 4 shows

equivalent circuit diagram of each switching intervals. During

steady state the operating power factor of the vehicle side

converter is considered to be lagging to achieve ZVS turn on

of switches S5 and S6.

Interval I (t0-t1): In this interval, switch S5 is closed and S6

is open. A positive voltage of 0.5Vo appears at the output of

vehicle side converter and due to presence of series resonant

tank a sinusoidal current flows through the receiver coil as

shown in Fig. 3 and 4a. In this duration the voltage and current

expressions of RC side components are given as

𝑣𝑆6 = 𝑣𝑜1 + 𝑣𝑜2, (1)

𝑖𝑆5 = 𝑖2, (2)

𝐶𝑜1𝑑𝑣𝑜1

𝑑𝑡= 𝑖𝑜−𝑖2, 𝐶𝑜2

𝑑𝑣𝑜2

𝑑𝑡= 𝑖𝑜, (3)

where, vo1 and vo2 are the voltages across capacitors Co1 and

Co2. During this interval ac side voltage of grid side converter

is positive and body diode D2 is forward biased. Thus the dc

link inductor current, Id1 passes through the TC tank network

as shown in Fig. 4a equivalent circuit. The TC side voltage and

current expressions are given as

𝑖𝐷2 , 𝑖𝑆4 = 𝑖𝑑1 + 𝑖𝑑2, (4)

𝐿𝑑1𝑑𝑖𝑑1

𝑑𝑡= 𝑣𝑑 − 𝑣𝑖, 𝐿𝑑2

𝑑𝑖𝑑2

𝑑𝑡= 𝑣𝑑, (5)

where, id1 and id2 are currents through inductors Ld1 and Ld2.

Interval II (t1-t2-t3): Interval t1-t2 is dead band period of the

switches S5 and S6. Because the power factor is lagging the

RC current, I2 does not change polarity at instant t1. Thus at t1

instant I2 starts flowing through the body diode of S6 as shown

in Fig. 4b. After the dead time i.e. at t2 instant the switch S6 is

turned on at zero voltage. However, the body diode of S6 i.e.

D6 keeps on conducting till the coil current I2 changes polarity

at instant t3. Throughout this interval the RC side voltage and

current expressions are given as

𝑣𝑆5 = 𝑣𝑜1 + 𝑣𝑜2, (6)

𝑖𝐷6 = 𝑖2, (7)

𝐶𝑜1𝑑𝑣𝑜1

𝑑𝑡= 𝑖𝑜, 𝐶𝑜2

𝑑𝑣𝑜2

𝑑𝑡= 𝑖2 + 𝑖𝑜, (8)



S6t

S5

id

t4t3 t5

t6t2 t7 t8 t9

iS5=i2

S5

i2

vr

vS5

iS5

ii

vi

is3

t1

S6

vS6

iS6

0.5vo

vo

0.5id

vo

iS6=-i2

vS1+vS3

t0

Fig. 3. Steady state waveforms of the converter during vehicle to grid operation.

Interval III (t3-t4-t5): At instant t3 the polarity of RC current, I2

changes and switch S6 starts conducting. For the interval t3-t4

the equivalent circuit is shown in Fig. 4c. Throughout the

period t0-t4, the body diode D2 of switch S2 conducts because

the magnitude of voltage Vi is positive. At instant t4 the

polarity of voltage Vi changes and becomes negative. Thus the

body diode D2 is reverse biased and D1 becomes forward

biased and takes complete dc link current Id. From instant t4

onwards the TC side switch voltage and current expressions

are given as

𝑖𝑆3 , 𝑖𝐷1 = 𝑖𝑑1 + 𝑖𝑑2, (9)

𝐿𝑑1𝑑𝑖𝑑1

𝑑𝑡= 𝑣𝑑, 𝐿𝑑2

𝑑𝑖𝑑2

𝑑𝑡= 𝑣𝑑 − 𝑣𝑖 . (10)

At instant t5 switch S6 is turned off and dead time of S5 and S6

begins. The switch S5 turns on at instant t6 when voltage

across it is zero, thus this switch also achieves ZVS at turn on.

This sequence of operation repeats in each switching cycle.

III. DESIGN PROCEDURE AND CONSIDERATIONS

A) Component ratings of G2V operation

Fig. 5a shows ac side equivalent circuit of the proposed

converter where the input and output are modeled as current

source and voltage source, respectively. The detailed design

equations, ZVS conditions and components’ rating for G2V

operation are reported in [30]. To achieve ZVS at device turn

on during G2V operation the current-fed converter output

power factor has to be lagging. If this power factor is leading

then the current-fed converter switches experiences soft

switching at device turn-off.

Cp

id

id1L1 i1

Cs

vi

D1 D2

id2

iiC2

L2i2

vr

(a)

(b)

(c)

vo

io

Co1S5

Co2

S6

id

C2L2i2

vr

vo

io

Co1S5

Co2

S6

id

C2L2i2

vr

vo

io

Co1S5

Co2

S6

Cp

id

id1L1 i1

Cs

vi

D1 D2

id2

ii

Cp

id1L1 i1

Cs

vi

D1 D2

id2

ii

Fig. 4. Equivalent circuit diagrams for different switching intervals during V2G operation (a) t0-t1 interval, (b) t1-t2- t3 interval and (c) t3-t4 interval.

Transmitter Coil Receiver Coil

Cp

C2

vr

iiL1i1 i2

jωMi2 jωMi1

ip

viv2

vs

v1

Cs

vc2

Transmitter Coil Receiver Coil

Cp

C2

vr

iiL1-M L2-Mi1 i2

M

ip

viv2

vs

v1

Cs

vc2

iM

(a)

(b)

Fig. 5. AC side equivalent circuit of the proposed topology. The wireless coil

is modelled as (a) coupled inductor (b) transformer.

To derive the voltage gain of the converter during G2V

operation, a transformer equivalent circuit of the tank network

is drawn as shown in Fig. 5b. Form Fig. 5b the voltage across

magnetizing inductance is derived as

𝑉𝑀 = 𝑉𝑟 − 𝐼2𝑍2 = 𝑉𝑟 −𝑉𝑟

𝑅𝑒𝑜𝑍2 , (11)

where, Reo is equivalent load resistance at the input of voltage

doubler circuit. Applying power balance at the input and



output of voltage doubler circuit, Reo in terms of output

resistance is calculated as

𝑅𝑒𝑜 = 2

𝜋2 .𝑉𝑜

𝐼𝑜=

2

𝜋2 𝑅𝑜, (12)

𝑍2 = 𝑠(𝐿2 − 𝑀) +1

𝑠𝐶2, (13)

and 𝑠 = 𝑗𝜔. Therefore, using (11) the current fed converter

output voltage is derived as

𝑉𝑖 = 𝑉𝑀 + 𝐼1𝑍1 = 𝑉𝑟 (1 +𝑍2

𝑅𝑒𝑜) + 𝑉𝑟

1

𝑠𝑀(1 +

𝑍2

𝑅𝑒𝑜) +

1

𝑅 𝑍1,

(14)

where,

𝑍1 = 𝑠(𝐿1 − 𝑀) +1

𝑠𝐶𝑠, (15)

During G2V operation the voltage gain expression is (Vr/Vi).

However, the input to the tank network is actually Ii. Thus, the

gain of the converter is (Vr/Ii) and it is derived as

𝐼𝑖 = 𝑠𝐶𝑝𝑉𝑖 + 𝐼1 = 𝑉𝑖 [(1 + 𝑠𝐶𝑝𝑍1) 1

𝑠𝑀(1 +

𝑍2

𝑅𝑒𝑜) +

1

𝑅 +

𝑠𝐶𝑝 (1 +𝑍2

𝑅𝑒𝑜)] . (16)

Fig. 6 shows the gain (Vr/Ii) and phase (∠Vr/Ii) plot of the

proposed converter during G2V operation.

Gai

n (

Vr/

I i)

in d

BP

has

e in

deg

Ro = 95 Ω

0

90

135

45

-45

36

40

28

32

20

24

-90

-135

Ro = 200 Ω

Ro = 150 Ω

Ro = 125 Ω

104.61 104.63104.65 104.67 104.69 104.71 104.73 104.75

Frequency in Hz

104.77

Ro = 95 Ω

Ro = 200 ΩRo = 150 Ω

Ro = 125 Ω

Fig. 6. Gain (Vr/Ii) and phase (∠Vr/Ii) plot of the proposed converter during G2V operation.

B) Derivation of component ratings of V2G operation

The ac side equivalent circuit of the converter remains same

during V2G operation as shown in Fig. 5a. Applying power

balance and considering active power flows in ac side due to

fundamental component only, rms values voltage and current

expressions are given as

𝐼𝑖 = 2√2

𝜋.

𝐼𝑑

2=

√2

𝜋. 𝐼𝑑 , (17)

𝑉𝑖 = 𝜋

√2. 𝑉𝑑, (18)

𝑅𝑒𝑖 = 𝑉𝑖

−𝐼𝑖=

𝜋2

2.

𝑉𝑑

−𝐼𝑑 , (19)

where, Rei is the equivalent load resistance during V2G

operation. Like the G2V operation, in V2G operation also the

rectifier side i.e. TC side converter input voltage Vi and Ii are

in same phase and their phasors are considered as reference

phasor. Using (18) the parallel capacitor, Cp current is derived

as

𝐼𝑝 =𝑉𝑖

(1

𝑗𝜔𝐶𝑝)

= 𝑗𝜋

√2𝜔𝐶𝑝𝑉𝑑. (20)

Applying KCL and using (17) and (20) the TC current and

series capacitor, Cs voltage are derived as

𝐼1 = −√2

𝜋. 𝐼𝑑 − 𝑗

𝜋

√2𝜔𝐶𝑝𝑉𝑑 , (21)

𝑉𝑠 = −𝜋

√2

𝐶𝑝

𝐶𝑠𝑉𝑑 + 𝑗

√2

𝜋

𝐼𝑑

𝜔𝐶𝑠 . (22)

Applying KVL at TC tank network the voltage across TC and

current through the RC are derived as

𝑉1 = −𝜋

√2𝑉𝑑 (1 +

𝐶𝑝

𝐶𝑠) − 𝑗

√2

𝜋

𝐼𝑑

𝜔𝐶𝑠 , (23)

𝐼2 =1

𝜔𝑀[

√2

𝜋𝐼𝑑 (𝜔𝐿1 −

1

𝜔𝐶𝑠) − 𝑗

𝜋

√2𝑉𝑑 (1 +

𝐶𝑝

𝐶𝑠− 𝜔2𝐿1𝐶𝑝)] .

(24)

Using (21) and (24) and applying KVL at the receiver side the

rms value of voltage across RC is derived as

𝑉2 =𝜋

√2𝑉𝑑 [𝜔2𝑀𝐶𝑝 +

𝐿2

𝑀 (1 +

𝐶𝑝

𝐶𝑠− 𝜔2𝐿1𝐶𝑝)] +

𝑗√2

𝜋𝐼𝑑 [

𝐿2

𝑀(𝜔𝐿1 −

1

𝜔𝐶𝑠) − 𝜔𝑀] .

(25)

Using (24) the RMS voltage across capacitor C2 is derived as

𝑉𝐶2 = −1

𝜔2𝑀𝐶2[

𝜋

√2𝑉𝑑 (1 +

𝐶𝑝

𝐶𝑠− 𝜔2𝐿1𝐶𝑝) + 𝑗

√2

𝜋𝐼𝑑 (𝜔𝐿1 −

1

𝜔𝐶𝑠)]. (26)

The peak blocking voltage of voltage doubler switches are

same as given in (24). The rms current rating of this switches

are given as

𝐼𝑆5 , 𝐼𝑆6 =1

√2𝐼2 , (27)

During this V2G operation the peak blocking voltage of

current fed converter switches are same as peak of ac side

voltage vi. The body diode average current of these switches

are given as

𝐼𝐷1 , 𝐼𝐷2

= 0.5𝐼𝑑 . (28)

Again, Fig. 5b transformer equivalent circuit is used to

derive voltage gain during V2G operation. The voltage across

magnetizing impedance branch is calculated as

𝑉𝑀 = 𝑉𝑖 + 𝐼1𝑍1 = 𝑉𝑖 + 𝑉𝑖𝑍1 (1

𝑅𝑒𝑖+ 𝑠𝐶𝑝). (29)

The voltage at the ac side of voltage doubler circuit is

calculated as

𝑉𝑟 = 𝑉𝑀 + 𝐼2𝑍2 = 𝑉𝑖 [(1 + 𝑍1𝑌) (1 +𝑍2

𝑠𝑀) + 𝑍2𝑌], (30)

where,

𝑌 =1

𝑅𝑒𝑖+ 𝑠𝐶𝑝. (31)

Fig. 7 shows the voltage gain (Vi/Vr) and phase (∠Vi/Vr) plot

during V2G operation for different loads.



G

ain

(V

i/V

r) i

n d

BP

has

e in

deg

0

45

-45

-90

15

20

5

10

0

-135

-180104.71

104.73 104.79104.75

Frequency in Hz

104.77

Rei=20

Rei=24

Rei=28

Rei=32

Rei=20

Rei=24Rei=28

Rei=32

Fig. 7. Voltage gain (Vi/Vr) and phase (∠Vi/Vr) plot of the proposed

converter during V2G operation.

C) ZVS conditions during V2G operation

To achieve soft switching of all the inverter switches

irrespective of load, variable frequency fixed duty cycle

control technique is used during V2G operation. From Fig. 5a

equivalent circuit the impedance at the ac side of voltage

doubler circuit i.e. Vr/I2 is derived as

Zero Phase angle line

Lagging power factor

soft-switching at device

turn-on

Leading power factor

Soft-switching at

device turn-off

Rei=20

Rei=24

Rei=28

104.71 104.73 104.75 104.77 104.79

Frequency in Hz

15

20

25

30

0

45

-45

Imp

edan

ce i

n o

hm

, Z

oP

ow

er f

acto

r an

gle

in

deg

, φ

o

UPF line

soft switching both at

turn-on and turn-off

Rei=32

Fig. 8. Plot of impedance (|𝒁𝒐|) and phase angle (∠𝒁𝒐) plot for different

Vd/Id ratio.

𝑍𝑜 =𝑉𝑟

𝐼2=

1

𝑗𝜔𝐶2+ 𝑗𝜔(𝐿2 − 𝑀) + 𝑗𝜔𝑀// [𝑗𝜔(𝐿1 − 𝑀) +

1

𝑗𝜔𝐶𝑠+ (𝑅𝑒𝑖//

1

𝑗𝜔𝐶𝑝)] . (32)

Fig. 8 shows the plot of the magnitude and phase of this

impedance for different equivalent load impedances. The

power injected from the voltage doubler circuit is given as,

𝑃𝑣2𝑔 = 𝑉𝑟

2

𝑅𝑒(𝑍𝑜)=

𝑉𝑟2

|𝑍𝑜|𝑐𝑜𝑠𝜑𝑜=

1

|𝑍𝑜|𝑐𝑜𝑠𝜑𝑜(

√2

𝜋𝑉𝑜)

2

. (33)

From Fig. 8, it is clear that around the resonant point the power

flow is maximum and power flow reduces when operating

frequency deviates from the resonant point. Keeping the

operating switching frequency higher than the resonance

frequency ensures lagging power factor operation. This leads

to ZVS turn on of the switches S5 and S6 irrespective of load.

D) Design example

To implement this bidirectional converter, appropriate

values of tank capacitors and coil inductance are required to be

determined. The total power transfer between the coupled coils

is derived as

𝑆 = 𝑗𝜔𝑀𝐼1𝐼2 = 𝑗𝜔𝑀 ×𝜋

√2𝐼𝑜 ×

1

𝜔𝑀[

𝜋

√2𝐼𝑜 (𝜔𝐿2 −

1

𝜔𝐶2) −

𝑗√2

𝜋𝑉𝑜] (34)

TABLE 1: SPECIFICATIONS FOR THE DESIGN EXAMPLE

Parameters Selected Values

Rated output power 1.2 kW

Switching frequency range 40 kHz - 60 kHz

Rated switching frequency 50 kHz

CSI input dc current, Id 7.0 A

Output/ battery voltage, vo 325 V (rated)

TABLE 2: CIRCUIT PARAMETERS

Components G2V

Rating

V2G

Rating

Selected devices/

Prototype Rating

Mutual

inductance

42 µH 42 µH 43 µH

TC current,

self-inductance

11A

211 µH

12

211 µH

12 A

211 µH

RC current,

self-inductance

8A

211 µH

10A

211 µH

12 A

211 µH

MOSFET

(CSI side)

Body diode

533V peak

4.9A RMS

-

530V peak

-

3.5A(Avg.)

Part no.-SCT2160KE

1.2 kV, 22 A, Rdson=

208mΩ

MOSFET

(vehicle side)

Body diode

325V peak

-

3.7A(avg.)

325V peak

8.4A RMS

-

Part no.-15EWL06FNTR

600V, 15A, Vf=1.05 V @

15 A

Capacitor, Cs 96 nF,

360 V RMS

96 nF,

390 V RMS

500V ac Cornel Doubler

film, 100 nF

Capacitor, Cp 96 nF,

375 V RMS

96 nF,

377 V RMS

500V ac Cornel Doubler

film, 100 nF

Capacitor, C2 48 nF,

530 V RMS

48 nF,

660 V RMS

2×500V ac Cornel

Doubler film, 48 nF

DC link

inductor Ld1, Ld2

1.7 mH

1.7 mH

1.2 mH

1.2 mH

1.2 mH

1.2 mH

Output caps,

Co1, Co2

224µF,

224µF

224µF,

224µF

300 µF, 300 µF

At close to resonant point the part [𝜔𝐿2 − 1/𝜔𝐶2] in (34)

becomes close to zero and complex part becomes zero. In this

condition the real power transfer from TC to RC is given as

𝑃 =𝜋

√2𝜔𝑀𝐼𝑜𝐼1 (35)

For easier explanation a design example of a 1.2 kW IPT

system is presented and this system is used for experimental

verification. The specifications of the target system are listed

in Table 1. In the 1.2kW experimental prototype the battery

voltage and rated charging currents are considered 325V and

3.7A respectively. Also the switching frequency corresponding



to the resonant point is considered approximately 50 kHz.

From (35), it is clear that to design the converter, I1 (RMS) has

to be considered. For the prototype it is considered as 11A

(RMS). Thus for 1.2kW power the required mutual inductance,

M is calculated to be 42 µH. From G2V operation in [30] it is

clear that with the given M the magnitude of TC voltage, V1

increase with higher value of TC self-inductance, L1. Also the

magnitude of Vi increases with the increase of TC self-

inductance where Vi directly determines switch voltage rating.

This indicates that the lower value the self-inductance i.e.

higher coupling factor leads to lower coil voltage. However,

there needs to be a trade- off between IPT coil size and coil

voltage because higher TC to RC coupling indicates larger coil

size. In the prototype IPT coil with 20% coupling is used and

this determines self-inductance of TC around 211µH. Since the

operating principle of the coupled coils are similar to two

winding transformer, therefore, TC to RC turns ratio near unity

provides better performance. In the porotype, TC to RC turns

ratio is selected to be unity.

The TC and RC side tank capacitors are calculated using

following expression derived in [30] as

𝜔𝑜1 =1

√𝐿1(𝐶𝑝𝐶𝑠

𝐶𝑝+𝐶𝑠)

, 𝜔𝑜2 =1

√𝐶2𝐿2 . (36)

Table 2 lists the selected circuit parameter values for both

during G2V and V2G operations. The voltage and current

ratings of the switches and body diodes are calculated from

[30] and (27), (28) for both G2V and V2G operations

respectively. The inductances of dc link inductors at close to

unity power factor of CSI is given as

𝐿𝑑1 , 𝐿𝑑2 =𝑉𝑑×𝑇𝑠

2Δ𝐼𝑑1 , (37)

where, 𝑇𝑠 = 1/𝑓𝑠 and Δ𝐼𝑑1 is peak to peak current ripple in

inductor Ld1 current. Also, the capacitances of output

capacitors are calculated as

𝐶𝑜1 , 𝐶𝑜2 =𝑇𝑠/2× 𝐼𝑜

Δ𝑉𝑜1 , (38)

where, Δ𝑉𝑜1 peak to peak voltage ripple across capacitor Co1.

The dc link inductor values Ld1 and Ld2 are calculated

considering a 20% ripple in inductor current and output

capacitors Co1 and Co2 are calculated considering 5% voltage

ripple at output voltage, Vo.

IV. EXPERIMENTAL RESULTS

A proof-of-concept hardware prototype rated at 1.2kW is

developed to verify the analysis and operation of the proposed

converter. Selected components are listed in Table 2. Fig. 9

shows the picture of the prototype. SKHI 61(R) is used as

MOSFET gate driver and DSP TMS320F28335 is used to

implement variable frequency digital control. During G2V

operation, an electronic load set at fixed voltage mode is used

to emulate 325V EV battery. During V2G operation this part is

replace by a dc power supply.

Fig. 10 shows experimental results of G2V operation for

900W power output and corresponding switching frequency of

the inverter is 46.6 kHz. The duty cycle of the inverter devices

is kept fixed to 0.52. Fig. 10a shows the profiles of the voltage

across and current through the coils. It is clear that the TC and

RC get almost pure sinusoidal voltage and current. Fig. 10b

shows voltage profiles of TC side tank network. From these

waveforms, it is clear that inverter output voltage, Vi is a

fraction of TC voltage (V1) where Vi directly determines the

switch voltage rating of the current-fed converter. Thus, this is

a major advantage of the proposed converter over simple

parallel LC tank at TC side that inverter switches get lower

voltage stress. It is clear that this is achieved because of the

presence of the series capacitor Cs.

Fig. 9. Photograph of 1.2kW experimental set-up.

v1

i1

i2

v2

v1

vs

vi

vi

-i2

vr

ii S1

vS1+vS3

vi

ii

S1

(a) (b)

(c) (d)

ZVS at device turn-on

Fig. 10. Experimental results of G2V operation at fs=46.6 kHz, Po=900W,

Vo=325V (a) TC and RC currents and voltages, i1 [10A/div], v1 [1.0 kV/div], i2

[10A/div], v2 [1.0 kV/div]; (b) Voltage across TC side tank components, v1, vi, vs [500V/div]; (c) Voltages and currents at inverter output and rectifier input, vi

[500V/div], ii [5A/ div], vr [100V/div], i2 [10A/ div]; (d) ZVS turn on of switch

S1, vs2+vs4 [500V/div], vi [500V/div], ii [5A/ div].

Fig. 10c shows the profiles of ac side voltages and currents of

both side converters during G2V operations. From these results

it is clear that the power factor of the TC side converter is

slightly lagging and this is suitable for ZVS turn-on of the

switches. Also RC side rectifier voltage and current profiles

are in the same phase and this ensures soft recovery of the

diodes D5 and D6. Fig. 10d confirms ZVS turn-on of the TC

side inverter switches because of lagging power factor

operation. At the instant when S1or S2 is turned-on, the

voltage across the corresponding leg switches i.e. S1-S3 or S2-

S4 are slightly negative. Switches S3 or S4 blocks this voltage

and switches S1 or S2 are turned-on at zero voltage.

Fig. 11 shows experiment results of G2V operation for

975W at switching frequency of 50 kHz. From [30], it is clear

that this operating region is in leading power factor region.



Fig. 11 validates this operation and the inverter switches S1

and S2 experience soft-switching at turn-off due to leading

power factor at inverter output. This operation is significant if

the inverter switches are selected as IGBTs to ensure zero turn

off loss. Fig. 12 shows efficiency curve for G2V operation.

The transmitter and receiver coil unloaded coupling factors are

250 and 200, respectively i.e. the internal resistances are 0.26Ω

and 0.32Ω, respectively. At rated load, coil-to-coil power loss

is around 4.8% of rated power. This forms more than 60% of

overall power loss. Therefore, major efficiency improvement is

possible by using optimum sized and shaped coils.

(a) (b)

(c) (d)

v1

i1

i2

v2

v1

vs

vi

vi

-i2

vr

ii S1

ii

S1

vS1+vS3

vi

Soft switching at turn-off

Fig. 11. Experimental results of G2V operation at fs=50 kHz, Po=975W,

Vo=325V (a) TC and RC currents and voltages, i1 [10A/div], v1 [1.0 kV/div], i2 [10A/div], v2 [1.0 kV/div]; (b) Voltage across TC side tank components, v1, vi,

vs [500V/div]; (c) Voltages and currents at inverter output and rectifier input, vi

[500V/div], ii [5A/ div], vr [100V/div], i2 [10A/ div]; (d) ZVS turn on of switch

S1, vs2+vs4 [500V/div], vi [500V/div], ii [5A/ div].

Fig. 13 and 14 shows the experimental results of V2G

operations for 950W and 760W power fed back to grid and

corresponding switching frequencies are 55.5 kHz and 57.5,

kHz respectively. During V2G operation, the vehicle side

voltage doubler circuit operates as voltage-fed inverter with a

fixed duty cycle 0.5. Both the Fig. 13a and 14a shows that the

coil voltages and currents are almost harmonic free. Fig. 13b

and 14b shows the voltage profiles of TC (V1), series capacitor

(Vs), and parallel capacitor (Vi). Form these results, it is clear

that parallel capacitor voltage, Vi is a fraction of TC voltage,

V1. Since, the voltage Vi directly determines the voltage rating

of the converter switches; therefore, the proposed topology is

capable of reducing switch voltage during V2G operation.

Fig. 12. Efficiency curve for G2V operation.

(a) (b)

(c) (d)

v1

i1

i2

v2

vi

-i2

vr

ii

v1

vs

vi

S5

id

vS5

-i2

vr


Fig. 13. Experimental results of V2G operation at fs=55.5 kHz, Po=950W,

Vo=325V (a) TC and RC currents and voltages, i1 [10A/div], v1 [1.0 kV/div], i2

[10A/div], v2 [1.0 kV/div]; (b) Voltage across TC side tank components, v1, vi, vs [500V/div] id [2A/div]; (c) Voltages and currents at inverter output and

rectifier input, vi [500V/div], ii [5A/ div], vr [200V/div], i2 [5A/ div]; (d) vs5

[200V/div], vr [200V/div], i2 [10A/ div].

(a) (b)

(c) (d)

v1

i1

i2

v2

vi

-i2

vr

ii

v1

vs

vi

S5

id

vS5

-i2

vr


Fig. 14. Experimental results of V2G operation at fs=57.5 kHz, Po=760W,

Vo=325V a) TC and RC currents and voltages, i1 [10A/div], v1 [1.0 kV/div], i2

[10A/div], v2 [1.0 kV/div]; (b) Voltage across TC side tank components, v1, vi, vs [500V/div] id [2A/div]; (c) Voltages and currents at inverter output and

rectifier input, vi [500V/div], ii [5A/ div], vr [200V/div], i2 [10A/ div]; (d)

Voltages and currents at inverter output and rectifier input, vs5 [200V/div], vr [200V/div], i2 [10A/ div].

Fig. 15. Efficiency curve for V2G operation.

50

60

70

80

90

100

400 600 800 1000 1200

Eff

icie

ncy

in

%

Power output in watt 70

75

80

85

90

95

100

400 600 800 1000 1200

Eff

icie

ncy

in

%

Power Output in watt



From Fig. 13c and 14c, it is clear that the ac side voltage and

current profiles of TC side converter are in the same phase due

to uncontrolled rectification. Also, ac side voltage and current

profiles of voltage doubler is slightly lagging and this is

suitable for ZVS turn-on of the switches S5 and S6. Fig. 13d

and 14d shows the ZVS turn-on of inverter switch S5. Similar

to switch S5 switch S6 also turns on at zero voltage.

Fig. 15 shows efficiency curve for V2G operation. Similar

to G2V operation, the wireless coil-to-coil power transfer loss

contributes major power loss in the converter circuit. The

proof-of-concept hardware is not optimized for packaging,

volume, and components. Therefore, the obtained efficiency is

close to 92%. Compare with the maximum reported efficiency

95% [31]-[32], this converter efficiency is less. Efficiency can

be farther improved by using improved quality and shaped

coils. The experimental results match closely with the

analytically predicted waveforms.

V. CONCLUSIONS

The contribution and focus of this paper is to propose,

analyze, and develop a new power electronics system for

wireless power transfer with G2V and V2G capability. A new

current-fed topology with bidirectional ability and current-

sharing and voltage doubling features has been proposed. The

proposed topology is analyzed with a new series-parallel CLC

tank network. The proposed tank network reduces the device

rating of grid side devices and permits the use of devices with

low on-state resistance and cost compared to traditional series

LC tank. Bidirectional inductive WPT is designed and

developed using proposed current-fed circuit and CLC tank

configuration. This is the first attempt to implement

bidirectional IPT with current-fed circuit and CLC tank and

demonstrate G2V and V2G operation. Keeping inverter output

power factor lagging, soft-switching turn-on of the inverter

switches is always ensured irrespective of load variation.

Complete mathematical analysis and systematic design is

reported. However, coils or magnetics design is not the focus

of the paper. Experimental results verify the reported analysis

and design and demonstrate the operation and performance.

REFERENCES

[1] J. M. Miller, O. C. Onar, and M. Chinthavali, "Primary-Side Power Flow

Control of Wireless Power Transfer for Electric Vehicle

Charging," Emerging and Selected Topics in Power Electronics, IEEE Journal of , vol.3, no.1, pp.147,162, March 2015.

[2] A. Khaligh, and S. Dusmez, "Comprehensive Topological Analysis of Conductive and Inductive Charging Solutions for Plug-In Electric

Vehicles," Vehicular Technology, IEEE Transactions on, vol.61, no.8,

pp.3475, 3489, Oct. 2012.

[3] G. Buja, M. Bertoluzzo and K. N. Mude, "Design and Experimentation

of WPT Charger for Electric City Car," in IEEE Transactions on

Industrial Electronics, vol. 62, no. 12, pp. 7436-7447, Dec. 2015.

[4] U. K. Madawala, and D. J. Thrimawithana, "A bidirectional inductive

power interface for electric vehicles in V2G systems," Industrial

Electronics, IEEE Transactions on , vol.58, no.10, pp.4789,4796, Oct. 2011.

[5] D. J. Thrimawithana and U. K. Madawala, "A Generalized Steady-State

Model for Bidirectional IPT Systems," in IEEE Transactions on Power Electronics, vol. 28, no. 10, pp. 4681-4689, Oct. 2013.

[6] K. Colak, E. Asa, M. Bojarski, D. Czarkowski and O. C. Onar, "A Novel

Phase-Shift Control of Semibridgeless Active Rectifier for Wireless Power Transfer," in IEEE Transactions on Power Electronics, vol. 30,

no. 11, pp. 6288-6297, Nov. 2015.

[7] B. Peschiera, K. Aditya, and S. S. Williamson, "Asymmetrical voltage-cancellation control for a series-series fixed-frequency inductive power

transfer system," Industrial Electronics Society, IECON 2014 - 40th

Annual Conference of the IEEE , vol., no., pp.2971,2977, Oct. 29 2014-

Nov. 1 2014.

[8] F. Musavi, and W. Eberle, "Overview of wireless power transfer

technologies for electric vehicle battery charging," Power Electronics,

IET , vol.7, no.1, pp.60,66, January 2014.

[9] J. M. Miller, O. C. Onar, C. White, S. Campbell, "Demonstrating

Dynamic Wireless Charging of an Electric Vehicle: The Benefit of

Electrochemical Capacitor Smoothing," in IEEE Power Electron. Magazine, vol. 1, no. 1, pp. 12-24, Mar 2014.

[10] S. Li, and C. Mi, "Wireless power transfer for electric vehicle

applications," Emerging and Selected Topics in Power Electronics, IEEE Journal of , vol.3, no.1, pp.4,17, March 2015.

[11] J. Shin et al., "Design and Implementation of Shaped Magnetic-

Resonance-Based Wireless Power Transfer System for Roadway-Powered Moving Electric Vehicles," in IEEE Transactions on Industrial

Electronics, vol. 61, no. 3, pp. 1179-1192, March 2014.

[12] S. Samanta, A. K. Rathore, "Analysis and design of current-fed (L)(C) (LC) converter for inductive wireless power transfer (IWPT)," in Energy

Conversion Congress and Exposition (ECCE), 2015 IEEE , vol., no.,

pp.5724-5731, 20-24 Sept. 2015.

[13] S. Samanta and A. K. Rathore, "A New Current-Fed CLC Transmitter

and LC Receiver Topology for Inductive Wireless Power Transfer

Application: Analysis, Design, and Experimental Results," in IEEE Transactions on Transportation Electrification, vol. 1, no. 4, pp. 357-

368, Dec. 2015.

[14] D. J. Thrimawithana, and U. K. Madawala, "A novel matrix converter

based bi-directional IPT power interface for V2G applications," Energy

Conference and Exhibition (EnergyCon), 2010 IEEE International, vol., no., pp.495,500, 18-22 Dec. 2010.

[15] K. Aditya, and S. S. Williamson, "Comparative study of Series-Series

and Series-Parallel compensation topologies for electric vehicle charging," Industrial Electronics (ISIE), 2014 IEEE 23rd International

Symposium on , vol., no., pp.426,430, 1-4 June 2014.

[16] K. N. Mude, M. Bertoluzzo, and G. Buja, "Inductive characteristics of different coupling setups for wireless charging of an electric city-car,"

Electric Vehicle Conference (IEVC), 2014 IEEE International , vol., no.,

pp.1,7, 17-19 Dec. 2014.

[17] P. Ning, J. M. Miller, Omer C. Onar, C. P. White, and L. D. Marlino, "A

compact wireless charging system development," APEC, 2013 Twenty-

Eighth Annual IEEE , vol., no., pp.3045,3050, 17-21 March 2013.

[18] K. Aditya, and S. S. Williamson, "Design considerations for loosely

coupled inductive power transfer (IPT) system for electric vehicle battery

charging - A comprehensive review," ITEC, 2014 IEEE , vol., no., pp.1,6, 15-18 June 2014.

[19] H. Z. Z. Beh, G. A. Covic and J. T. Boys, "Wireless Fleet Charging

System for Electric Bicycles," in IEEE Journal of Emerging and Selected Topics in Power Electronics, vol. 3, no. 1, pp. 75-86, March 2015.

[20] M. K. Naik, M. Bertoluzzo, and G. Buja, "Design of a contactless battery

charging system," AFRICON, 2013, vol., no., pp.1,6, 9-12 Sept. 2013.

[21] S. K. Mishra, R. Adda, S. Sekhar, A. Joshi and A. K. Rathore, "Power

transfer using portable surfaces in capacitively coupled power transfer

technology," in IET Power Electronics, vol. 9, no. 5, pp. 997-1008, 4 20 2016.

[22] S. Samanta and A. K. Rathore, “Bidirectional Wireless Power Transfer

Topology Using Current-Fed Half Bridge CLC Transmitter LC Receiver Configuration for Medium Power Applications,” IEEE IAS Annual

Meeting, Portland, USA, 2-6 Oct 2016.

[23] S. Samanta and A. K. Rathore, "A comparison and performance evaluation of L-C and C-C-L compensation schemes on CSI based

inductive WPT application," 2016 IEEE 25th International Symposium

on Industrial Electronics (ISIE), Santa Clara, CA, 2016, pp. 817-822.

[24] I. I. Nam, R. A. Dougal and E. Santi, "Novel Unity-Gain Frequency

Tracking Control of Series–Series Resonant Converter to Improve Efficiency and Receiver Positioning Flexibility in Wireless Charging of

Portable Electronics," in IEEE Transactions on Industry Applications,

vol. 51, no. 1, pp. 385-397, Jan.-Feb. 2015.

[25] K. E. Koh, T. C. Beh, T. Imura and Y. Hori, "Impedance Matching and

Power Division Using Impedance Inverter for Wireless Power Transfer via Magnetic Resonant Coupling," in IEEE Transactions on Industry

Applications, vol. 50, no. 3, pp. 2061-2070, May-June 2014.

[26] J. Y. Lee, H. Y. Shen and K. C. Chan, "Design and Implementation of

Removable and Closed-Shape Dual-Ring Pickup for Contactless Linear

Inductive Power Track System," in IEEE Transactions on Industry Applications, vol. 50, no. 6, pp. 4036-4046, Nov.-Dec. 2014.



[27] V. Cirimele, S. G. Rosu, P. Guglielmi and F. Freschi, "Performance

evaluation of wireless power transfer systems for electric vehicles using the opposition method," 2015 IEEE 1st International Forum on Research

and Technologies for Society and Industry Leveraging a better

tomorrow.

[28] D. Bavastro et al., "Design of Wireless Power Transmission for a Charge

While Driving System," in IEEE Transactions on Magnetics, vol. 50, no. 2, pp. 965-968, Feb. 2014.

[29] M. Khalilian, S. G. Rosu, V. Cirimele, P. Guglielmi and R. Ruffo, "Load identification in dynamic wireless power transfer system utilizing current

injection in the transmitting coil," 2016 IEEE Wireless Power Transfer

Conference (WPTC), Aveiro, 2016, pp. 1-4.

[30] S. Samanta and A. K. Rathore, "A new inductive wireless power transfer

topology using current-fed half bridge CLC transmitter LC receiver configuration," 2016 IEEE Energy Conversion Congress and Exposition

(ECCE), Milwaukee, WI, USA, 2016, pp. 1-8.

[31] S. H. Lee and R. D. Lorenz, "Development and Validation of Model for

95%-Efficiency 220-W Wireless Power Transfer Over a 30-cm Air Gap,"

in IEEE Transactions on Industry Applications, vol. 47, no. 6, pp. 2495-2504, Nov.-Dec. 2011.

[32] S. G. Rosu, M. Khalilian, V. Cirimele and P. Guglielmi, "A dynamic wireless charging system for electric vehicles based on DC/AC

converters with SiC MOSFET-IGBT switches and resonant gate-drive,"

IECON 2016 - 42nd Annual Conference of the IEEE Industrial Electronics Society, Florence, Italy, 2016, pp. 4465-4470.

Suvendu Samanta (S’16) received the B.E. degree from

IIEST Shibpur (formerly Bengal Engineering and

Science University), Howrah, India, in 2009 and the M.Tech. degree from Indian Institute of Technology

Kanpur, India, in 2013, both in Electrical Engineering.

He is presently working towards the PhD degree in the Department of Electrical and Computer Engineering

(ECE), Concordia University, Montreal, QC, Canada. He

was with Coal India Ltd. as operation and maintenance engineer of Electrical and Mechanical section from 2009-2011. Also from April 2014-April 2016 he

worked as research Engineer in the ECE department, National University of

Singapore, Singapore. His research interests include modeling and control of power electronic converters especially in wireless power transfer applications.

Akshay Kumar Rathore (M’05 - SM’12) received the M.Tech. degree from the Indian Institute of

Technology (BHU), Varanasi, India, in 2003. He

received the Ph.D. degree from the University of Victoria, Victoria, BC, Canada, in 2008. He had two

subsequent Postdoctoral Research Appointments with

the University of Wuppertal, Germany, and University of Illinois at Chicago, IL, USA. From

November 2010 to February 2016, he was an

Assistant Professor in the Department of Electrical and Computer Engineering, National University of Singapore, Singapore. He is currently an Associate

Professor at the Department of Electrical and Computer Engineering,

Concordia University, Montreal, QC, Canada. He has published more than 160 research papers in international journals and conferences including 55 IEEE

Transactions. His research is mainly focused on current-fed converters and

multilevel inverters. He is leading the area of current-fed power electronics and contributed to analysis, design, and development of new classes of

current-fed soft-switching converters.

Dr. Rathore is an Associate Editor of IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, IEEE TRANSACTIONS ON INDUSTRIAL

ELECTRONICS, IEEE TRANSACTIONS ON TRANSPORTATION

ELECTRIFICATION, IEEE TRANSACTIONS ON SUSTAINABLE ENERGY, and IEEE JOURNAL OF EMERGING SELECTED TOPICS IN

POWER ELECTRONICS. Dr. Rathore is Editor-in-Chief of IEEE Industrial

Electronics Technology News (ITeN). He is Paper Review Chair of IEEE Transactions on Industry Applications for Industrial Automation and Control.

He is a Distinguished Lecturer and Executive Board Member-at-Large of IEEE

Industry Applications Society. He received the Gold Medal during his M.Tech. degree for securing highest academic standing among all electrical engineering

specializations. He was a recipient of University Ph.D. Fellowship, NSERC

research assistantship and Thouvenelle Graduate Scholarship during his PhD. He received 2013 IEEE IAS Andrew W. Smith Outstanding Young Member

Achievement Award and 2014 Isao Takahashi Power Electronics Award.

Duleepa J. Thrimawithana (M’09) received his BE

in Electrical Engineering (with First Class Honors) in 2005 and his Ph.D. in power electronics in 2009

from The University of Auckland, Auckland, New

Zealand. From 2005 to 2008, he worked in collaboration with Tru-Test Ltd. in Auckland as a

Research Engineer in the areas of power converters

and high-voltage pulse generator design. He joined the Department of Electrical and Computer

Engineering at The University of Auckland in 2009

where he currently works as a Senior Lecturer. He also serves as the Chairman of the Joint Chapter of IEEE Industrial Electronics and Industrial Applications

Society, New Zealand (North). He has co-authored over 100 international

journal and conference publications, and holds 8 patent families on wireless power transfer technologies with several pending. In recognition of his

outstanding contributions to engineering as an early career researcher, Dr.

Thrimawithana received the Jim and Hazel D. Lord Fellowship in 2014. His main research areas include wireless power transfer, power electronics and

renewable energy.

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This article has been accepted for publication in a future issue of …msrprojectshyd.com/upload/academicprojects/770f1a1406e8... · 2017-09-26 · reactive power consumed by the