Ultrasonic Transducers Generating, detecting & processing ultrasonic signals This page is provided to provide a few handy hints to project students grappling with ultrasonics Last Update 27/5/08 Ultrasonics is the production of sound waves above the frequency of human hearing and can be used in a variety of applications such as, sonic rulers, proximity detectors, movement detectors, liquid level measurement. Ultrasonics is used in medicine and robotics, security devices, laboratory and industrial cleaners and a host of other applications. Many in-depth studies of ultrasonic transducer equivalent circuits have been produced but these studies are far beyond the scope of these pages.

Ultrasonic Transducers

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Page 1: Ultrasonic Transducers

Ultrasonic Transducers

Generating, detecting & processing ultrasonic signals

This page is provided to provide a few handy hints to project students grappling with ultrasonics

Last Update 27/5/08

Ultrasonics is the production of sound waves above the frequency of human hearing and can be used in a variety of applications such as, sonic rulers, proximity detectors, movement detectors, liquid level measurement. Ultrasonics is used in medicine and robotics, security devices, laboratory and industrial cleaners and a host of other applications. Many in-depth studies of ultrasonic transducer equivalent circuits have been produced but these studies are far beyond the scope of these pages.

Fig 1

A simplified model of the series and parallel equivalent circuits of an ultrasonic transducer.

from:- http://www.mpi-ultrasonics.com/transducers1.html

There are two main types of transducers used to transmit ultrasonic signals. They are the Piezo type and the electrostatic type. It is even possible to send ultrasonic signals using a

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conventional high frequency electromagnetic speaker (tweeter). This discussion will centre around the piezo type of transducer. The electrostatic type may be included at a later stage

and we will look at using conventional speakers as well.

Below is a specification for a typical piezo type ultrasonic transducer Tx/Rx pair. Note that the main difference between a receiving and transmitting transducer is the impedance. The

transmitter generally has a low impedance and a receiver has a higher impedance. Some ultrasonics transducers can be used as both receiver and transmitter however, when

designing circuits for these devices take careful note of the impedance and design the circuitry to suit.




Fig 2

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Part 1. Driving ultrasonic transducers

Piezo-electric ultrasonic output transducers are devices that have a complex impedance . They require a rather high voltage drive (~tens of Volts) to achieve maximum output and must be driven at their resonant frequency for maximum power output. The transducers can also be driven at low voltage by paralleled inverter gates or power amplifiers. This method is quite adequate for short range signals.

Any output transducer must be characterized to determine what load it presents to a driving circuit under the operating conditions. An ultrasonic transducer resonates at a particular frequency and at resonance the current and voltage are in phase. This results in a load that looks primarily resistive. This resistance must be known reliably for future calculations to be made. Often the load at resonance is given in the specifications. If it is not, it has to be measured.

Transducer Load Resistance at resonance

To measure the resistance of a Tx transducer similar to the one described above, at resonance, a 40kHz sine wave drive was applied with the drive voltage being observed on a CRO and a current probe being used to measure drive current. The resistance of the transducer was calculated from these measurements and was found to be ~ 500 Ohms . This is as expected from the spec. sheet.

Low voltage/low power

Low voltage/low power drive of transducers can be achieved by using inverting gates like 74HC04. This method is commonly used and is recommended for short range devices. The circuit below drive the transducer with pulses whose amplitude is twice that of the power supply

Reference: http://www.interq.or.jp/japan/se-inoue/e_sonic1_3.htm#3

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Fig 3

Power Output Using gate drivers: During each cycle there is an average of 9V across the load. The average power delivered is around (VxV)/R= 81/500 = 162mW

A standard kind of circuit for this drive technique can be found from Texas Instrument and uses a MPS430 uP


Higher voltage/Higher power

Higher voltage/Higher power drive can be achieved in a number of different ways:-

High voltage supply and amplifier. (expensive and complex) A Load matching transformer and transformer driver( Good for power efficiency

and high output, cheap but may be bulky) Inductive flyback technique (Not as high power as transformer but cheap, simple &

fairly easy to construct)

1. High voltage amplifier

We will not cover this method at this stage as it requires high voltage power supplies. The methods employed here will be for low voltage (e.g. 9V) battery operation only.

2. Matching transformer and transformer driver

To achieve the high driving voltages required a small transformer may be used.

"Off the Shelf" Audio Transformer

A small audio output transformer may be used here as they are cheap , available and work at low ultrasonic frequencies (~40kHz). These transformers are designed to match an amplifier with an output impedance of ~1.5k to a speaker of nominally 8 Ohm. If we turn it around we can use it to drive a higher impedance (U/S transducer) load with a low voltage, low impedance amplifier. Commonly, these transformer have a winding ratio of around 15:1. The relevant relationships are:

Es=Ep(Ns/Np) and Zp=Zs (Np/Ns)^2

Where Es=secondary voltage, Ep=primary voltage, Zp=source impedance, Zs=load impedance. Np=Primary turns, Ns=secondary turns.

(Note Zp & Zs does not refer to the inductive reactance of the windings XL [a common mistake] but to the impedance of the [Thevenin] driving source and load)

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Example: We wish to drive the transducer described above with an resonant impedance of 500 Ohm to continuous 20V rms with a 15:1 transformer (assume it is ideal), What is driving voltage, amplifier output impedance and power supply requirements?

Ep=Es x (1/15)= 1.3Vrms

Impedance looking into the primary is Zp=500 x 4.4E-3= 2.2 Ohm

Current = 1.3/2.2= 0.6 Amp(rms)

Power=0.6^2 x 2.2 = 0.8 Watt(rms)

Driving voltage p-p= 2 x 1.3X1.414= 2.8V

The disadvantage of this now obvious. To use this kind of transformer efficiently we have to provide a power supply of 2.8Volt and capable of delivering 0.6 Amp RMS. Also, it may be noted on experimentation capacitive effects in the windings may cause problems at 40KHz.

What if we have a different supply voltage and/or the necessary driving current is unachievable?

In this case we will have to design and build our own transformer.

Custom transformer design.

Ferrite materials for use at ultrasonic frequencies, associated bobbins and clips etc. are common and cheap but the design must be carried out carefully. The transformer is of a conventional type designed for square wave operation.

The design parameters are:


Resistive load of transducer at resonance= 500 Ohm

Max drive voltage on transducer = 20V RMS therefore (as determined earlier) power input to transducer (Po)=0.8 Watt

Step 1. Determine transformer winding ratio:

(as before) Es=Ep(Ns/Np)

Es/Ep=20/9 = 2 (note that this is a lot different from the ratio in example above)

Step 2. Select a transformer core size and material (Philips core specifications.)

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At 40kHz virtually any ferrite core material can be used as this is a common frequency used for switched mode power transformers.

We choose a common ferrite material (3F3)

The size and shape of the core should be appropriate for the required power throughput.

A low profile core that is easy to construct is the EFD type. For less that 5 Watt virtually any EFD core will suffice.

We will use an EFD15 core (mainly because that is what I've got!). A smaller core could be used like an EFD 10

Step3. Calculate number of primary and secondary windings

We can use the (square wave modified) transformer equation derived from Faraday's laws to determine the number of turns for a particular core.

N=(V. t)/(B.Ae)


N = number of turns (of a particular winding)

V = voltage across turns when current flowing (in Volts)

Bdelta= Max. change in flux density in core (in Tesla)

t= "on" time of driving pulse (in uSec) we choose a maximum “on” time as half the period, so t=T/2

Ae = Effective core area (in mm^2)

For an EFD15 core of grade 3F3 we can see from the specification sheet that:-

Ae = 15 mm^2, Al = 700, Bdelta is around 250mT(250E-3) for 3F3 material.

for 40Khz T =1 /40E3 = 25uS

t = T/2 = 12.5uS which is the maximum “on” time for a square wave at 40kHz

Therefore Ns=(9x12.5)/(250E-3x15)= Ns= 30 Turns

Ns/Np=2 Therefore Np=15 Turns.

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Note that we could get the same driving voltage by use any combination of windings with a ratio of 2:1. The purpose of using the equation is the efficient use of the transformer core material without causing magnetic saturating of the core the material and hence risking stress on the driving transistors. When transformer cores saturate the driving current rises sharply.

Step 4. Calculate the winding inductances (for simulation purposes only.)

For the the purpose of designing a single frequency matching transformer the inductance value of a winding is irrelevant but to simulate the drive circuits in SwitcherCadIII©, the inductance of the windings must be known and a coupling co-efficient must be assumed to enable a simulation of the transformer. It is also useful to calculate winding inductance so that it can be compared with a measured value.

From the specs for the core chosen we see that:-

L=N^2.AL (nH)

Lp =15^2x700=157500nH = 157uH (Primary winding inductance)

Ls =30^2x700=630000nH = 630uH (Secondary winding inductance)

The coupling co-efficient of k= 0.9 is commonly used in SwitcherCadIII©

Step 5. Choose wires sizes

For max efficiency and minimum losses the copper losses in the winding must equal the core loss. As this is a low power transformer we will leave out the calculation of the wire size as this can introduce complexities that will produce little advantage. For this design we will use 0.25mm diameter enamel coated copper wire for both primary and secondary windings.

Push-Pull transformer driving circuits

There are a number of options for driving the transformer primary including a full bridge topography:-.

The full bridge design has the advantage of using the primary winding to its maximum advantage by pushing and pulling current during alternative halves of the driving waveform. The disadvantage is that four devices are needed as well as some control logic. In this simulated design we will use complementary pairs of transistors as the switching elements. In the first half cycle Q2 & Q3 are on, in the second half cycle Q2 and Q3 are on. At least one device on each side of the bridge need to be off when there is no drive.

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Fig 4


Fig 5

From the results of the simulation we can see that the output across the load is 39V pk-pk (or 18V RMS) which is a little less than calculated due to switching losses in the

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transistors. This is acceptable as we chose the maximum value to start with and a little less voltage across the transducer will allow us a safety margin. The current in the switches is around 400mA pk which means that a 1 Amp transistor pair like a BD139/BD140 is appropriate.

A variation on this classic full bridge push-pull configuration requires two primary windings but allows the use of only two switching devices. As both device source terminals are ground referred we can conveniently use two N channel mosfet transistors here

Fig 6

The simulation results show the required output voltage (with losses) and ~400mA peak current through each device on the half cycle. As this design requires only the addition of 15 primary turns it is quite attractive for simplicity and convenience.

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Fig 7

Power output using transformer and a push-pull driver: During each cycle there is an average of ~20V RMS across the load. The average power delivered is (VxV)/R= 400/500

= 800mW which in this case is the maximum allowable for the transducer used in these examples.

Other possible driving configurations include half bridge capacitively coupled circuits or the use of an audio power amplifier on split or single rails. None of these configurations has

any substantial advantage over the ones shown above.

3. Inductive Flyback Method

Inductive flyback is a method that can give some impressive performance with a low parts count. The driving circuit is quite cheap and straightforward in design.

This the simulation circuit .

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Fig 8

How it works:

When the transistor is turned on the current in the inductor will rise linearly and energy is stored in an induced magnetic field. When the transistor is turned off the inductor produces a "flyback" voltage of opposite polarity in an attempt to maintain the flow of current and the stored energy is "dumped" into the transducer. This process produces a high voltage across the ultrasonic transducer. It must be kept in mind that the flyback voltage must not exceed the maximum collector voltage of the transistor or the peak drive voltage of the transducer. It has been noted, during experiments, that both the transducer and transistor are reasonably hardy but remember that a piezo electric device like an ultrasonic transducer can be destroyed instantaneously if the maximum voltage is grossly exceeded.

Determine the time the transistor is turned on. As previously determined, for 40kHz drive using a square wave this is T/2 = 0.5x (1/40k) = 12.5uS.

Note that the inductor does not have to be driven by a symmetric "square" wave but the "on" time for the transistor must be known and the frequency must be appropriate to the the resonant frequency of the transducer. More power will be delivered to the load with greater "on" times, but this will result in a higher flyback voltage which could damage the transistor or the transducer. Somewhere a compromise must be made.

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We then determine a maximum driving voltage. The specifications say 60V p-p is safe. This usually refers to a square wave pulse and it has been found experimentally that the instantaneous peak voltage can be be somewhat higher without doing damage to the transducer. Let us choose 70Vpk-pk. To deliver this voltage we will need a peak inductor driving current of I= V/R = (70+9)/500 =~ 0.16Amp.

The value of the inductance can be calculated by using a relationship from Faraday's law's

V=L di/dt

Where: V=Applied voltage,L = Inductor value (in Henrys), di/dt = the rate of change of current in the inductor

Let us assume for the sake of mathematical simplicity that the coil resistance is fairly low. (<<1.0 Ohm) this would make the current rise in the coil close to linear.

From this relationship the rise of current in L1 is di/dt = 0.16/12.5E-6 = 12.8kV/sec Using a supply voltage of 9V then V=9, hence L1 = 9/12.8k = 0.7mH (preferred

value is 0.68mH)

A simulation of this circuit reveals the following results:- The peak inductor current is as expected (blue trace) with a flyback voltage across the transducer of around 80V-9V=71V (red trace). Practically, a flyback voltage of a little less than this may be expected with this circuit considering a typical coil resistance. If this voltage is too high for the transducer then the peak current should be lowered and hence the value of the inductor increased. Alternatively the transistor "on" time can be reduced to adjust the drive voltage down to the required level. The switching MOS transistor should be at least a 120V type.

Page 13: Ultrasonic Transducers

Fig 9


Making your own inductor.

Designing and winding a high quality power inductor is fairly straight forward. All you have to do is choose an appropriate sized core that will not saturate at this flux density level

and use the AL of the core to calculate number of turns.

e.g. Using the EFD15 core with 3F3 material used above with an AL of 700nH, Ae=15mm^2

(You could use a EFD10 for a physically smaller inductor)

L=N^2.AL (nH)

For a 700uH inductance, N=Square root of (700E3/700)= 31 Turns

Check for saturation Using N=(V. ton)/(Bchange.Ae)

Power output using flyback: Assume duty cycle of 0.5, Peak current = 160mA, efficiency= 0.75 (power delivered from coil to load). The average power delivered is (Ipk x V x D x 0.75)/2= 0.16x9x0.5x0.75/2= 270mW

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Testing the transmitter circuit

To test if the transmitter is working it should be driven by a continuous 40kHz pulse at the previously determined duty cycle. The receiver transducer should produce about 500mV peak to peak or better at about 0.5 metre (at resonance ~40kHz). This Rx signal should be a sine wave. View this received signal with a CRO attached to the Rx transducer and move the transducer backwards and forwards to observe the signal phase change caused by the delay in sound transmission. Varying the driving frequency should show definite "peaking" of the received signal around 40kHz .

Be careful to enclose these transmitting circuits in a metal box as broadband RF noise can be generated by the large switching signals. Keep the leads to the output transducer short. When not in box tune in to the noise on an AM radio.

Part 2. Detecting an ultrasonic signal

Ultrasonic receiving transducers, like the transmitting transducers, have a complex impedance.

Determining transducer source impedance at resonance

No matter how complex, the source impedance can be reduced to a Thevenin or Norton equivalent circuit. The source impedance can be determined from the specifications of the device but as these are sometimes hard to obtain or if incomplete information is presented then the value can be measured.

Method. (Using a Murata MA40A3R receiving transducer.)

a. If you don't have a working transmitter then drive the Rx transducer at the resonant frequency with a sine wave generator and measure the peak current and driving voltage with a CRO. The Thevenin resistance can be determined by using Rt=Vin/Iin. At resonance the voltage and current should be in phase. If you haven't got a current probe then place a variable resistance in series with the transducer and increase the value until the voltage across the transducer is half that of the source voltage. The value of the resistor is equal to the Thevenin resistance. (Don't forget to take the driving source resistance into account.)

b. If you have a transmitter working tape the Rx transducer directly against the transmitter face. Observe the output on a CRO. Decrease the load resistance across the Rx transducer until the output peak signal is half that of the unloaded signal. This resistance is equivalent to the Thevenin resistance.

Page 15: Ultrasonic Transducers

The 40kHz signal can be amplified by an AC amplifier. The amplifier could be designed using an op-amp or transistors. A designer could be tempted to follow the amplifier with filters of various combinations, high pass, low pass or bandpass but there is not much justification for this complexity as high Q and therefore narrow bandpass has already been achieved by the resonance processes and interference from out of band signals should not be a problem in most cases. Other signal processing for intelligence recovery can follow this. A voltage gain of between 5 & 60 is usually more than sufficient for the amplifier depending on the transmitted distance required. In some applications and for carrying out short range experiments an amplifier and other analogue circuitry is not even necessary, especially if one of the higher powered transmitting methods is used. The signal can be applied directly to a comparator for direct conversion into a series of pulses for digital processing. If an op-amp is used then choose a high slew rate type.

Below is one approach to an amplifier designed for ultrasound using 2 BJT's in the front end configured as a differential pair. The purpose of using this configuration is:-

a. To easily match the impedance of the transducer by using feedback. This is done for maximum signal power transfer at resonance.The simulated input impedance of this amp. is around 3.7k Ohms.

b. To provide maximum gain around 40KHz. The purpose of this is not to provide a high Q filter action but to limit the upper and lower frequency response of the amplifier to attenuate signals from common interference sources e.g. mains (50Hz, 100Hz), AM radio 530kHz->1.6MHz) etc. The -3dB bandwidth is around 330kHz.

c. To provide a differential output to the comparator and provide easy thresh-hold adjustment. This can be done by increasing R9 with a trimpot.

Page 16: Ultrasonic Transducers

Fig 10 An ultrasound pre-amp with feedback and comparator


Fig 11 Comparator input voltages and output current


Fig 12. Input voltage and current

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Fig 13. The frequency response of the amplifier.


The actual circuit waveforms

Signal on collector of Q1 or Q2

Page 18: Ultrasonic Transducers

As can be seen the receiver transducer oscillates well beyond 8 cycles after it has been energized by incoming 8 40kHz sound pulses.



Part 3. Measuring distance

Absolute measurement using "time of flight"

A short burst of ultrasound can be used to measure distance and detect movement by measuring the "time of flight" of the burst. The burst needs to be long enough for the resonant circuits to reach full amplitude but short enough for the minimum measuring distance to be acceptable. A good compromise has been found to be around 8-16 cycles for a 40kHz device.

If ever there was an application crying out for a micro controller it is this one, as generating a repetitive burst of 8 cycles at 40kHz is a fairly straightforward task for the programmer using the internal timers. There may be times however, where a micro controller does not need to be used, when the it is too busy processing other signals or performing calculations or when the circuit needs to be cheap. In this case a state machine can be used. A PAL/GAL could do the job or it could be achieved using discrete logic. Below is such a circuit based on a CD4060 binary counter and a 74HCT107A (or similar) negative edge triggered JK flip flop. This circuit can be used in conjunction with a receiver circuit described in the next section. A burst repetition frequency of 19.53Hz has been used here but this can be varied by choosing another frequency from the CD4060 outputs. An accurate 80kHz external oscillator can be made using another CD4060 and a 5.12MHz crystal.

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Below is a circuit for generating continuous ultrasound or 8 cycles of 40kHz pulses at 19.53Hz repetition rate. The use of a crystal ensures an accurate 80kHz reference for the state machine. A push pull centre tapped primary transformer is used to drive the transducer. (See above for transformer details). If you wish to use the flyback transducer drive technique then omit the AND gates (U3) and drive the switching transistor from output Q on U2B. If you wish to drive the transformer (or the transducer directly as in Fig 3.) using paralleled inverting gates take the driving signal from this point also. To transmit a larger number of 40kHz cycles use output Q5 for 16 cycles , Q6 for 32 cycles etc. to CK input of U2A


Fig 14© David Castles



Fig 18

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This output signal can be timed and displayed by further circuitry or the signal can be timed by a micro controller and converted to time or distance.

Q. See if you can you measure the speed of sound in air at your local ambient temp and air pressure using this setup

note: Speed of sound is 331.5 m/s at 0 °C at sea level

Small relative displacements

In the discussion above we saw how an ultrasonic Rx and Tx could be used as a range finder by using the time of flight of a pulse of ultrasonic energy. This is applicable for distances from a couple of hundred millimetres to many metres. What about measuring small displacements using ultrasound? This can be done but we must rethink how the devices are to be used.

Experiment & observation. Apply a continuous drive to the Tx transducer and observe the received signal on a CRO. You will notice that as you move the devices closer together and further apart there is a phase shift between the two sine waves.

Fig 19

What relative displacement does this 180° phase shift between the TX signal and Rx signal represent?

1 period(40kHz) = 1/f = 1/40E3 = 25E-6 sec

Velocity of sound is 331.5m/sec (remember this is a temperature dependent value)

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Distance sound travels in 1 period is 331.5x25E-6 = 8.2875 x 10E-3 metres = 8.2875 mm

A 180° phase shift would then represent a relative displacement of 360/180 x 8.2875 = 4.14375mm

If the displacement device is microprocessor controlled, then measuring this phase shift should not be a difficult matter. The received signal could be "squared" using a comparator and then logically ANDed or XORed with the transmitted signal. The resultant pulse width would represent the phase difference as time. This time could be measured and the relative displacement calculated.


Fig 20 © David Castles

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The diagram Fig. 16 above shows the transmit driving signal (at the top) logically combined with the received signal at various phase shifts using and AND function and an XOR function. The fine blue traces show the received signal and the red trace shows the resultant signal. The XOR function is the one classically used as a phase detector so we shall concentrate on this. The AND function is useful but as you can see from the diagram only produces half the number of pulses in a given displacement.

There two problems now to consider.

1. A phase shift of less that 360°.

Given you have established a reference point within a 360° shift it is a fairly simple matter to measure a change of XORed signal pulse width using a microprocessor or analogue circuitry. The problem lies in determining the direction of travel as a trend in pulse width change could mean one direction within one particular 180° boundary or the opposite direction within the other 180° boundary. This then requires some means of detecting in which direction the phase shift is moving. A clue to solving this can be found by observing in Fig 16 the direction of change of the XORed pulse on the rising/falling edge of the Tx pulse. From 0° to 108° the rising/falling edge of TX results in a rising edge of the XOR pulse. from 180° to 360° the rising/falling edge of Tx causes a falling edge of the XORed pulse. Using a microcontroller the direction of travel can be easily determined by recording the effect of a rising/falling edge on the Tx signal. This combined with the pulse width of the XORed signal can be used in an appropriate algorithm to determine the relative displacement.

2. A phase shift of more than 360°.

To measure displacements of more than 360° we have to consider what happens at the 0°/360° crossover point. If the pulse width reaches a minimum and increases then a 360 boundary has been crossed and some kind of count must be recorded to keep track of whole 360° phase shifts. A precise boundary can be recorded by detecting a change of direction of the XORed signal at the 0°/360° boundary. Knowing the quantity of complete wavelengths travelled and any partial phase displacement combined with direction means the total relative displacement can be calculated.


Circuits and information on this page may be freely used for educational purposes but material quoted in assignments or reports must be attributed to the author. (David Castles , Electronics Engineer, (retired) Dept. Electronic Eng. Latrobe University, Melbourne, Australia.


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The simulator used is SwitcherCADIII©(freeware from Linear Technology Download Site)