Wireless sub-THz communication systemwith high data rateS. Koenig1*, D. Lopez-Diaz2, J. Antes1,3, F. Boes1,3, R. Henneberger4, A. Leuther2, A. Tessmann2,

R. Schmogrow1,5, D. Hillerkuss1,5, R. Palmer1, T. Zwick1, C. Koos1, W. Freude1*, O. Ambacher2,

J. Leuthold1,5* and I. Kallfass2,3*

In communications, the frequency range 0.1–30 THz is essen-tially terra incognita. Recently, research has focused on thisterahertz gap, because the high carrier frequencies promiseunprecedented channel capacities1. Indeed, data rates of100 Gbit s21 were predicted2 for 2015. Here, we present, forthe first time, a single-input and single-output wireless com-munication system at 237.5 GHz for transmitting data over20 m at a data rate of 100 Gbit s21. This breakthrough resultsfrom combining terahertz photonics and electronics, wherebya narrow-band terahertz carrier is photonically generated bymixing comb lines of a mode-locked laser in a uni-travelling-carrier photodiode. The uni-travelling-carrier photodiodeoutput is then radiated over a beam-focusing antenna. Thesignal is received by a millimetre-wave monolithic integratedcircuit comprising novel terahertz mixers and amplifiers. Webelieve that this approach provides a path to scale wirelesscommunications to Tbit s21 rates over distances of >1 km.

Data rates in both fibre-optic and wireless communications havebeen increasing exponentially over recent decades. For the upcom-ing decade this trend seems to be unbroken, at least as far as fibre-optic communications is concerned. In wireless communications,however, the spectral resources are extremely limited because ofthe heavy use of today’s conventional frequency range up to60 GHz. Even with spectrally highly efficient quadrature amplitudemodulation (QAM) and the spatial diversity achieved with multiple-input and multiple-output (MIMO) technology, a significantcapacity enhancement to multi-gigabit or even terabit wireless trans-mission requires larger bandwidths, which are only available in thehigh millimetre-wave and terahertz region. Between 200 and300 GHz there is a transmission window with low atmosphericlosses3. In contrast to free-space optical links, millimetre-waveor terahertz transmission is much less affected by adverse weatherconditions like rain and fog4,5.

Here, we present for the first time a single-input single-output(SISO) wireless 100 Gbit s21 link with a carrier frequency of237.5 GHz. By combining state-of-the-art terahertz photonics andelectronics and by utilizing the large frequency range in the tera-hertz window between 200 and 300 GHz, we realize a wireless100 Gbit s21 link with SISO technology, that is, a link with onetransmit antenna and one receive antenna. To date, 100 Gbit s21

wireless links have only been demonstrated at lower carrier frequen-cies around 100 GHz (refs 6–8) over a wireless distance of �1 m,with a bit error ratio (BER) of �1 × 1023. Because of the limitedbandwidth, these systems relied on optical polarization multiplexingand spatial MIMO with more than one wireless transmitter and

receiver. Here, a 100 Gbit s21 wireless transmission capacity isachieved without resorting to MIMO technology.

We envisage various applications1,9,10 for such a high-capacitywireless link (Fig. 1). If an end-to-end fibre connection is absentand the deployment of a new fibre link is not economical, asmight be the case in difficult-to-access terrains and certain ruralareas (last mile problem), or if an already existing fibre connectionfails, a permanent or ad hoc wireless connection could help.Furthermore, we anticipate indoor applications, such as high-speed wireless data transfers between mobile terminals anddesktop computers. Figure 1 presents a schematic of our100 Gbit s21 wireless experiment embedded into an applicationscenario where an obstacle, here a broad river, is bridged by thewireless link. We first discuss the general system concept, andthen provide further details.

For the transmitter (Tx) we use a terahertz photonics technologyset-up (Fig. 1). We generate exceptionally pure and stable terahertzcarriers by heterodyning frequency-locked laser lines11. A controlunit contains a single mode-locked laser (MLL), selects the appro-priate frequency-locked comb lines, and modulates data on thecarrier lines. An optical fibre transmits the modulated carrierstogether with an unmodulated comb line, which acts as a remotelocal oscillator (LO), to a remote uni-travelling-carrier photodiode(UTC-PD). By photomixing the LO and the modulated carriers,radiofrequency signals are generated. Optical heterodyning hasalready been used in earlier works to implement multi-gigabit wire-less systems in the 60 GHz band12–14, in the W-band (75–110 GHz,refs 6–8,15,16), at 120 GHz (refs 17,18) and at carrier frequenciesbeyond 200 GHz (refs 19,20).

For the electronic in-phase/quadrature (IQ) receiver (Rx), we usea custom-developed, active millimetre-wave monolithic integratedcircuit (MMIC) with a radiofrequency bandwidth of 35 GHz(refs 21,22). This is, to the best of our knowledge, the first activebroadband IQ mixer at 237.5 GHz. The Rx comprises a low-noiseamplifier (LNA) and a subharmonic downconversion IQ mixer,and is realized in a metamorphic high electron mobility transistor(mHEMT) technology (Supplementary Section S5) featuring agate length of 35 nm and a cutoff frequency of more than900 GHz (refs 23,24). The complex data are directly downcon-verted to the baseband and separated into I and Q signals.Previous works6–8 in the W-band have illustrated the importanceof a high carrier frequency. However, to date, no direct downcon-version to baseband has been used due to a lack of IQ mixers cover-ing the full W-band. For carrier frequencies beyond 110 GHz(refs 17–20), simple on–off keying modulation and envelope

1Karlsruhe Institute of Technology (KIT), 76131 Karlsruhe, Germany, 2Fraunhofer Institute for Applied Solid-State Physics (IAF), 79108 Freiburg,Germany, 3University of Stuttgart, 70569 Stuttgart, Germany, 4Radiometer Physics GmbH, 53340 Meckenheim, Germany, 5ETH Zurich, 8092 Zurich,Switzerland. *e-mail: s.koenig@kit.edu; w.freude@kit.edu; juergleuthold@ethz.ch; ingmar.kallfass@ilh.uni-stuttgart.de

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detection with Schottky barrier diodes have comprised the stateof the art, and data rates up to 28 Gbit s21 over 0.5 m havebeen demonstrated20.

In our photonic terahertz Tx, a single MLL (Time BandwidthProducts ERGO XG) outputs a frequency comb with frequencyseparation DfMLL¼ 12.5 GHz. A programmable optical filter(Finisar waveshaper 4000E) selects the desired optical carriers fromthe comb: the LO at fLO¼ 193.138 THz (lLO¼ 1,552.22 nm), acentral carrier at f1¼ 193.3755 THz (l1¼ 1,550.31 nm) and twoadjacent carriers at f2,3¼ f1+DfMLL. The central carrier and thegroup of two adjacent carriers are each modulated with differentIQ data, provided by two independent multi-format transmitters25

MFTx 1 and MFTx 2. The modulated carriers and the LO areamplified, aligned in polarization, combined, and superimposedon a UTC-PD (Supplementary Section S4) similar to the oneshown in ref. 26 for photomixing. The photodiode current com-prises signals at the three intermediate frequencies f1,2,3 2 fLO thatare radiated from a horn antenna and focused by an asphericalplano-convex lens.

The receiver is equipped with a similar lens-and-horn antennaset-up, and is positioned at distances d¼ 5, 10, 20 and 40 m. Afterthe Rx antenna, a variable waveguide attenuator allows the inputpower to be adjusted for our MMIC Rx. Owing to the integratedLNA stage, the Rx ideally operates at an input power of only232 dBm. The downconverted I and Q baseband signals are digi-tized by an 80 GSa s21 analogue-to-digital converter (Agilent real-time oscilloscope DSO-X-93204A). Subsequent offline processing27,28

includes channel equalization, carrier recovery, filtering and signaldemodulation. (See Methods and Supplementary Sections S1, S2and S3 for more details on the Tx, the antennas and the Rx.)

We transmit two kinds of signals over the SISO link. First, in the‘single channel’ case (Fig. 2a), only the carrier f1 (Ch1) and theLO fLO are present. We generate QPSK, 8QAM and 16QAMsignals with a symbol rate of up to 25 GBd, resulting in bit ratesof up to 100 Gbit s21. Second, we discuss the ‘multiple channel’case (Fig. 2b), with two additional carriers f2,3 (Ch2, Ch3) and afixed spacing DfMLL¼ 12.5 GHz (Fig. 2b, right). We use sinc-likepulse shaping29 with a raised-cosine spectrum and a roll-off factorb¼ 0.35 in all channels. Symbol rates are 13 GBd in Ch1, and8 GBd in Ch2 and Ch3. Channel Ch1 and its neighbours overlapslightly. For our set-up with given comb line spacing and limitedsystem bandwidth, this represents the best tradeoff between digitalprocessing effort in the MFT, linear channel crosstalk and spectralefficiency. At the Tx, we shape the spectrum with the waveshaperto compensate the frequency response of the UTC-PD. With16QAM signalling in Ch1 and 8QAM signalling in Ch2 and Ch3,the aggregate bit rate is 100 Gbit s21.

We now discuss the single-channel and multiple-channel datatransmission experiments. Details on BER, error vector magnitude(EVM) and forward error correction (FEC) are given in the Methods.

Figure 3a shows the EVM of a 50 Gbit s21 single-channel QPSKsignal for different wireless transmission distances as a function ofthe relative Rx gain G¼ 1/LAtt, where LAtt is the loss added by theRx waveguide attenuator. The constellation diagram shows clearand distinct symbols. With increasing Rx gain, the EVM decreases(improves) until it starts to show a floor at 16% (corresponding toa residual BER of �1 × 1029), which can be seen for d¼ 5, 10and 20 m. This floor is due to the constant electronic Rx noisecontribution. For d¼ 40 m the available receiving power is insuffi-cient to reach this EVM floor.

Photonic THz transmitter

Electronic THz receiver

Atten. MMIC19.79 GHz E

O

Q

×60°

90°Control unit

UTC-PD

237.5 GHz

IQ input 2

IQ input 2

IQ input 1

MFTx 2

MFTx 2

MFTx 1

MLLWS LO

fLO

f3

f1

f2ΔfMLL =

12.5 GHz

Figure 1 | Prospective application scenario for a long-range, high-capacity wireless communication link at terahertz frequencies. Here, the wireless link

bridges a broad river in difficult-to-access terrain to provide high-speed internet access in remote and rural areas. The figure shows key elements of our

100 Gbit s21 SISO wireless link at 237.5 GHz carrier frequency. The wireless signal is generated photonically: a mode-locked laser in the control unit outputs

a frequency comb with frequency separation DfMLL¼ 12.5 GHz. Three adjacent comb lines f1, f2 and f3 and a line denoted LO are selected by a waveshaper

(WS). Two multi-format transmitters (MFTx1,2) encode different in-phase (I) and quadrature (Q) data onto the central carrier f1 and the group of two carriers

f2 and f3. The modulated data carriers f1, f2 and f3 and the unmodulated LO fLO are mixed by a remote UTC-PD. Three distinct millimetre-wave carriers

encoded with data at f2 2 fLO¼ 225 GHz, f1 2 fLO¼ 237.5 GHz and f3 2 fLO¼ 250 GHz are then radiated over a beam-focusing antenna, and received by a

second beam-focusing antenna (Atten.¼ variable attenuator). An electronic receiver comprising active MMICs acts as an optical transmitter via an electro-

optical converter (E/O) for transporting the data over a connecting fibre to any other optical receiver.

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From Fig. 3a it can also be seen that doubling the transmissiondistance in the far-field (10 . . . 20 . . . 40 m) requires four times thereceiving power, as would be expected from the free-space path loss.

For 8QAM and 16QAM signalling (Fig. 3b,c) and for d¼ 5 and10 m, the BER reaches a minimum and begins to increase if the

receiving power becomes larger. This behaviour is attributed tononlinearities of the active MMIC Rx that affect multilevel data inparticular. This can be seen when comparing to Fig. 3a, where theconstant-amplitude QPSK signal does not exhibit an EVM increasefor comparable receiving powers. For the 50 Gbit s21 16QAM

5 m

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Ch3Ch2 Ch1

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−20 −10 0

−20 −10 0 −20 −10 0

FEC7%

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FEC7%

FEC 7%FEC 7%FEC 7%

Figure 3 | Transmission of multi-gigabit wireless signals with data rates up to 100 Gbit s21 over distances of 5, 10, 20 and 40 m. a–e, EVM for received

QPSK signals (a,d) and BER for received QAM signals (b,c,e) as a function of the relative receiver (Rx) gain. The horizontal dashed line indicates the

threshold for error-free transmission when using second-generation hard-decision FEC with 7% overhead. Insets: optimum constellation diagrams for 20 m

wireless transmission. a, EVM for the single-channel 50 Gbit s21 QPSK signal. b, Measured BER for the single-channel 75 Gbit s21 8QAM.

c, 50 Gbit s21 (open squares) and 100 Gbit s21 (filled squares) 16QAM signals. d, EVM for 58 Gbit s21 multiple-channel transmission with QPSK in all three

channels. e, BER for 100 Gbit s21 multiple-channel transmission with 52 Gbit s21 16QAM in the centre channel and 24 Gbit s21 8QAM in the

neighbouring channels.

fLO

Blow-up: Multiple-channel signalLO + multiple-channel signalLO + single-channel signal

Frequency

−10

All roll-offs β = 0.35

−20

b

Ch1:13 GBd

Ch2:8 GBd

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f1

f3

Frequency (THz)193.25 193.50193.00

Frequency (THz)193.25 193.50193.00

f1 fLOΔf = 237.5 GHzΔf = 237.5 GHz

Figure 2 | Optical spectra before the UTC-PD for photonic terahertz signal generation. a, Single-channel signal configuration. The frequency spacing Df

between the unmodulated optical local oscillator fLO and the modulated data carrier f1 (12.5 GBd 16QAM or 25 GBd QPSK, 8QAM or 16QAM) defines the

generated wireless carrier frequency after optical heterodyning by the UTC-PD. b, Multiple-channel signal configuration. Three data carriers with channel

spacing DfMLL¼ 12.5 GHz are modulated with 13 GBd 16QAM (Ch1) and 8 GBd 8QAM (Ch2 and Ch3). Sinc-like pulses with a raised-cosine spectrum and a

roll-off factor b¼0.35 are transmitted in all three channels (right panel). Ch1 and the neighbouring Ch2 and Ch3 overlap slightly. The three channel powers

are intentionally made different to precompensate the frequency response of the UTC-PD.

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signal, we achieve an optimum BER of 3.7 × 1024 after 40 m. Forthe 100 Gbit s21 16QAM signal, the optimum BER for distancesup to 20 m remains below the FEC limit for 7% overhead.Figure 3d shows the EVM results for the 58 Gbit s21 multiple-channel signal transmission with sinc-like pulse shaping andQPSK modulation in all channels. The 13 GBd centre channelCh1 performs better than both 8 GBd outer channels Ch2 andCh3, because the limited receiver bandwidth cuts into the spectraof Ch2 and Ch3.

Figure 3e shows the results for the aggregate 100 Gbit s21 mul-tiple-channel transmission where Ch2 and Ch3 each carry an8 GBd 8QAM signal, and Ch1 carries a 13 GBd 16QAM signal.The same receiver bandwidth limitation applies as above, but thesignal-to-noise ratio required for transmitting the 16QAM signalis so much larger than for the 8QAM signals that the BER of thecentre channel Ch1 is worse.

Figure 4 presents the optimum BER from Fig. 3b,c,e as a functionof the wireless transmission distance. The results for a zero trans-mission distance are taken from electrical back-to-back measure-ments (Supplementary Section S3). Over distances of d¼ 5, 10and 20 m, the receiving power is sufficient for an error-correctedtransmission, while the BER for d¼ 40 m increases because of thelimited power budget.

In summary, we have demonstrated, for the first time, a SISOwireless link with a maximum data rate of 100 Gbit s21 bridging adistance of 20 m using up to three radiofrequency subcarriers near237.5 GHz. Over 40 m, a maximum data rate of 75 Gbit s21 wastransmitted with a single radiofrequency carrier at 237.5 GHz and8QAM signalling. Our results show that combining terahertz photo-nics and terahertz electronics could pave the way to high-capacitywireless communications using carrier frequencies in the terahertzgap. By exploiting optical polarization multiplexing and MIMOtechnology with two photonic wireless transmitters and twoMMIC receivers, we could boost the system’s performance to adata rate of 200 Gbit s21. A further scaling towards a Tbit s21

system could be achieved by taking advantage of optical wavelengthdivision multiplexing, frequency division multiplexing in the200–300 GHz band and spatial multiplexing with severalMIMO links in parallel. By using MMIC amplifiers23,24 at theUTC-PD output and high-gain Cassegrain antennas (.50 dBi) atboth the Tx and Rx side, considerably larger point-to-point dis-tances (.1 km) will be possible. To reach a reliable, compact,

precommercial system, the integration of the photonic and elec-tronic subcomponents is essential. Recently, the first monolithicIQ electrooptic modulator was demonstrated in gallium arsenide30,which is also the substrate material of the MMICs in this Letter.Noting this and anticipating further progress in the field, weforesee that the photonic and electronic subcomponents will beintegrated either on a monolithic or hybrid (heterogeneous31) plat-form, leading to compact terahertz transmitters and receivers forwireless communications at Tbit s21 rates.

MethodsTransmitter, antennas and receiver. The UTC-PD current comprises unwantedintercarrier mixing products at ( f3 2 f1)¼ ( f1 2 f2)¼ 12.5 GHz and ( f3 2 f2)¼25 GHz, which are filtered by the rectangular waveguide (TE10 mode cutofffrequency fc¼ 174 GHz) connecting the UTC-PD to the antenna. The UTC-PDis operated with a total optical input power of þ14 dBm, which leads to anaverage photodiode current of 6.5 mA and an output power of 213.5 dBmfor a single modulated terahertz channel at f1 2 fLO¼ 237.5 GHz. We estimatethat the single sideband (SSB) phase noise of the optically generated 237.5 GHzcarrier is 233.1, 266.4, 280.8 and 281.9 dBc Hz21 at offset frequencies of 100 Hz,1 kHz, 10 kHz and 100 kHz, respectively (see Supplementary Section S2 fordetails on phase noise characterization).

The combined transmitter–receiver lens-and-horn antenna gain is 86 dBi(compared to an isotropic antenna). The beamwidth is ,28, which suitswireless point-to-point connections. The small wavelength of �1 mm wouldallow the realization of very compact antenna arrays. For f¼ 237.5 GHz, d¼ 10 mand vacuum speed of light c, the free-space path loss is LFSPL¼ (4pdf/c)2,corresponding to 100 dB.

In the MMIC Rx, the LNA stage provides �30 dB gain and forwards the signal toan IQ mixer. The subharmonic mixer operates with a LO with half the frequency(118.75 GHz) of the transmitted centre carrier. This LO signal is derived via afrequency multiplier chain from a synthesizer operating at 19.79 GHz. Theestimated SSB phase noise of the Rx LO at 237.5 GHz is 244.4, 256.4, 256.4 and278.4 dBc Hz21 at offset frequencies of 100 Hz, 1 kHz, 10 kHz and 100 kHz,respectively. The complete IQ Rx is integrated on a chip area of 2.5 × 1 mm2 andpackaged into a split-block waveguide module. The packaged Rx module featuresa measured conversion gain of 3.8 dB and a noise figure of ,10 dB (refs 21,22).

BER, EVM and FEC. In our experiments, we digitized and recorded the receiveddata with a real-time oscilloscope (Agilent DSO-X-93204A) for offline digitalsignal processing and signal quality evaluation. The recording length was always80 ms. For a 25 GBd single-channel signal, a recording length of 80 ms correspondsto a total of 2 × 106 symbols, that is, we evaluated �4 × 106 received bits for the50 Gbit s21 QPSK signal, 6 × 106 bits for the 75 Gbit s21 8PSK signal and 8 × 106

bits for the 100 Gbit s21 16QAM signal.A widely accepted quality metric for a data signal is the BER. To determine

the BER, a known bit sequence must be transmitted. In our experiments, wetransmitted a pseudo-random bit sequence of length 215 2 1. At the receiver, wecompared the recorded bit sequence with the originally sent bit sequence andcounted the number of bit errors. This number divided by the number of comparedbits is then an estimate for the BER. However, if the signal quality is high, that is,if the BER is small, a significant amount of time (a very large number of recordings)is needed to count enough errors to determine a reliable BER value.

In our case, the quality of the received QPSK signals (Fig. 3a,d) was so good thattoo few or no errors were counted within the recording length of 80 ms. For allother signal formats, enough errors were counted within the recording length, andthe BER could be computed. For low-error rate QPSK signals we measured theEVM instead of the BER. The EVM is a standard metric for signal quality in wirelessand wireline communication systems. The EVM describes the effective distance ofthe received complex symbols from their ideal positions in the constellationdiagram. Under the assumptions of additive white Gaussian noise, data-aidedreception and quadratically arranged xQAM constellations, with log2(x) being aneven number (for example, x¼ 4 for QPSK), the BER can be estimated frommeasured EVM values32. The horizontal solid lines in Fig. 3a,d indicatecorresponding calculated BER values32 of 1 × 1029.

The horizontal dashed lines in Fig. 3 and Fig. 4 indicate a BER of 4.5 × 1023.A raw BER of 4.5 × 1023 is the threshold for error-free transmission when usingsecond-generation hard-decision FEC with 7% overhead33. If the FEC worksproperly, the corrected output BER becomes ,1 × 10215.

In our experimental set-up (Supplementary Section S1), we use a 3 dB fibrecoupler to combine the modulated signal(s) and the LO before photomixing. Asdescribed by Hirata and colleagues34, variations in temperature might causefluctuations in the optical path length difference between the signal path and the LOpath. This in turn could result in phase fluctuations of the generated terahertz signalon a timescale of several milliseconds to seconds. Thus, within the recording lengthof 80 ms, we did not see any significant influence of this phase effect on signalquality. However, error bursts would be an issue in practical systems. On the otherhand, Hirata and colleagues34 have shown that error bursts can be avoided if

Single-channel signals

06

4

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(BER

)a b

2

6

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) 2

10 20 30Wireless distance (m)

40 0 10 20 30Wireless distance (m)

40

16QAM, 100 Gbit s−1

FEC 7%

8QAM, 75 Gbit s−1

16QAM, 50 Gbit s−1

100 Gbit s−1

multiple-channel signal

Ch1: 16QAM52 Gbit s−1

Ch2: 8QAM24 Gbit s−1

Ch3: 8QAM24 Gbit s−1

Figure 4 | Optimum data transmission as a function of wireless

transmission distance. a, 8QAM and 16QAM single-channel transmission.

b, 100 Gbit s21 multiple-channel transmission. The results for a zero

transmission distance are taken from electrical back-to-back measurements

(Supplementary Section S3). For distances of 5, 10 and 20 m, the receiving

power suffices for error-free reception if an appropriate FEC with 7%

overhead is used. For a transmission distance of 40 m the BER increases

beyond that limit because of the limited power budget.

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photonic integrated structures and combiners are used instead of discrete fibre-based components.

Received 3 May 2013; accepted 5 September 2013;published online 13 October 2013

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AcknowledgementsThe authors thank NTT Electronics (NEL) for providing the UTC-PD for this experiment,and W. Schroeder from the Karlsruhe Institute of Technology (KIT) for the artwork inFig. 1. The authors also acknowledge support from the MILLILINK project(Millimeterwellen-Drahtlos-Links in optischen Kommunikationsnetzwerken) funded bythe German Federal Ministry of Research and Education (BMBF; grant 01BP1023), theKarlsruhe School of Optics & Photonics (KSOP), the Helmholtz International ResearchSchool for Teratronics (HIRST) at the Karlsruhe Institute of Technology (KIT) and theAgilent University Relation Program.

Author contributionsS.K. developed the concept, designed and performed the experiments, implemented thephotonic transmitter, characterized the MMIC receiver module, analysed the data andwrote the paper. D.L.-D. designed the MMIC receiver chip and characterized the MMICreceiver module. R.H. packaged the MMIC receiver chip and provided the horn antennas.A.T. simulated and designed the MMIC amplifiers. A.L. developed the 35 nm mHEMTMMIC technology. J.A., F.B., R.S., D.H. and R.P. assisted in performing the experimentsand analysing the data. T.Z., C.K., W.F., O.A., J.L. and I.K. developed the concept and wrotethe paper.

Additional informationSupplementary information is available in the online version of the paper. Reprints andpermissions information is available online at www.nature.com/reprints. Correspondence andrequests for materials should be addressed to S.K., W.F., J.L. and I.K.

Competing financial interestsThe authors declare no competing financial interests.

NATURE PHOTONICS DOI: 10.1038/NPHOTON.2013.275 LETTERS

NATURE PHOTONICS | VOL 7 | DECEMBER 2013 | www.nature.com/naturephotonics 981

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