frequency bands by appropriately adjusting the dimension of cut

wire pairs.

4. CONCLUSIONS

In conclusion, we propose a dual band directive patch antenna with

meta-superstrate made from the periodic cut wire pairs with differ-

ent lengths. The influence of the variation of distance between the

meta-superstrate and ground on the radiation performance is dis-

cussed according to the F-P theory. The tradeoffs and design con-

siderations of this value are investigated to obtain the optimal per-

formance. The measured results show that the designed antenna

gain is observably enhanced by about 8 dB at two frequency bands.

The higher directivity can be expected if the aperture size of the

meta-superstrate is made larger than that of our design.

ACKNOWLEDGMENTS

This work was supported by 973 Program of China (No.

2006CB302900) and scientific innovation funding for graduates of

Chinese Academy of Sciences.

REFERENCES

1. V.G. Veselago, The electrodynamics of substance with simultane-

ously negative value of e and l, Sov Phys Usp 10 (1968),

509–514.

2. M. Thevenot, A. Reineix, and B. Jecko, Directive photonic band-

gap antennas, IEEE Trans Microwave Theory Tech 47 (1999),

2115–2122.

3. C. Cheype, C. Serier, M. Thevenot, A. Reineix, and B. Jecko, An

electromagnetic bandgap resonator antenna, IEEE Trans Antennas

Propag 50 (2002), 1285–1290.

4. A.R. Weily, L. Horvath, K.P. Esselle, B.C. Sanders, and T.S. Bird,

A planar resonator antenna based on a woodpile EBG material,

IEEE Trans Antennas Propag 53 (2005), 216–223.

5. Y.J. Lee, J. Yeo, R. Mittra, and W.S. Park, Design of a frequency

selective surface (FSS) type superstrate for dual-band directivity

enhancement of microstrip patch antenna, IEEE Antennas Propag

Soc Int Symp 43 (2005), 462–467.

6. Y.J. Lee, J. Yeo, R. Mittra, and W.S. Park, Thin frequency selec-

tive surface (FSS) superstrate with different periodicities for dual-

band directivity enhancement, In: IEEE International Workshop on

Antenna Technology: Small Antennas and Novel Metamaterials

(IWAT), 2005, pp. 375–378.

7. D.H. Lee, Y.J. Lee, J. Yeo, R. Mittra, and W.S. Park, Design of novel

thin frequency selective surface superstrates for dual-band directivity

enhancement, IETMicrowave Antennas Propag 1 (2007), 248–254.

8. J. Hu, C.S. Yan, and Q.C. Lin, A new patch antenna with metama-

terial cover, J Zhejiang Univ Sci A 7 (2006), 89–94.

9. H.L. Xu, Y.G. Lv, X.G. Luo, and C.L. Du, Metamaterial super-

strate and electromagnetic band-gap substrate for high directive

antenna, Int J Infrared Milli Waves 29 (2008), 493–498.

10. S. Enoch, G. Tayeb, P. Sabouroux, and P. Vincont, A metamaterial

for directive emission, Phys Rev Lett 89 (2002), 213902.

11. G. Dolling, C. Enkrich, M. Wegener, J.F. Zhou, C.M. Soukoulis,

and S. Linden, Cut-wire pairs and plate pairs as magnetic atoms

for optical metamaterials, Opt Lett 30 (2005), 3198–3200.

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ing approach to left-handed material design, Opt Lett 31 (2006),

3620–3622.

13. C. Huang, Z. Zhao, Q. Feng, J. Cui, and X.G. Luo, Metamaterial

composed of wire pairs exhibiting dual band negative refraction,

Appl Phys B, accepted.

14. Y.J. Lee, J. Yeo, R. Mittra, and W.S. Park, Application of electro-

magnetic bandgap (EBG) superstrates with controllable defects for

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VC 2009 Wiley Periodicals, Inc.

A LOW-POWER CMOS DUAL-BAND RFRECEIVER FOR IEEE 802.15.4-BASEDSENSOR NODE APPLICATIONS

Trung-Kien Nguyen, Hoyong Kang, Nae-Soo Kim,and Cheol-Sig PyoRFID/USN Research Division, Electronics and TelecommunicationsResearch Institute (ETRI), Daejeon, Republic of Korea;Corresponding author: rfmicro.vn@gmail.com

Received 21 April 2009

ABSTRACT: This article presents the design and experimental results

of a low-power dual-band RF receiver front-end including a dual-bandlow-noise amplifier (LNA) and a downconversion mixer based on the

IEEE 802.15.4 standard for sensor node applications. A dual-band LNAwith two inputs is tuned to two resonant frequencies by controlling thevoltage on a switched MOS. The implemented RF receiver front-end

achieves a maximum voltage conversion gain of 31 and 21 dB, a noisefigure of 6 and 9 dB at the 868/915 MHz and 2.45 GHz bands,

Figure 6 Radiation patterns of the patch antenna with meta-superstrate

in the E plane at (a) 14.7 GHz (measured) and 14.5 GHz (simulated);

(b) 15.3 GHz (measured) and 15.4 GHz (simulated). [Color figure can

be viewed in the online issue, which is available at www.interscience.

wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 52, No. 1, January 2010 163

respectively. The RF receiver front-end dissipates a total of 3 mA

(including I/Q mixers) under a supply voltage of 1.8 V at both operationbands. VC 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 52:

163–166, 2010; Published online in Wiley InterScience

(www.interscience.wiley.com). DOI 10.1002/mop.24876

Key words: dual-band; low power; receiver; sensor node

1. INTRODUCTION

Recently, the ubiquitous sensor network is drawing a lot of

attention as a method for realizing a ubiquitous society. IEEE

802.15.4 was finally standardized in 2006 [1] as a low-rate wire-

less personal area network for low-complexity, low-cost, and

low-power short-range wireless connectivity among inexpensive

fixed, portable, and mobile devices. Moreover, the demand for

multiband RF radio transceiver implementation is rapidly

increasing.

Most of the conventional multiband receiver architectures are

achieved by using multiple individual receiving paths, which

increase the cost and power consumption [2]. To overcome

these disadvantages, a concurrent receiver architecture that can

simultaneous receive signals in a multiband has been developed

[3]. However, this architecture experiences a linearity problem

because the spurs in one band can corrupt signals in the other

band. Also, as there is only one input on a low-noise amplifier

(LNA), its input matching quality factor cannot be optimized for

both bands. Consequently, the gain, noise figure (NF), and line-

arity cannot be optimized.

In this article, a proposed RF receiver front-end is designed

by adopting a dual-band LNA with two separate inputs and

using one downconversion mixer as shown in Figure 1, in which

the first band will cover the 868 and 915 MHz bands as they are

very close together and have the same physical requirements,

and the second band will cover the 2.45 GHz band. This dual-

band receiver adopts a low-IF architecture with an intermediate

frequency (IF) of 2 MHz. The IF of 2 MHz has been chosen by

considering the 3-dB bandwidth of both operation frequency

bands. Also, the chosen IF frequency is well beyond the flicker

noise corner. The detailed design and experimental results of the

proposed RF receiver front-end are described in the following

sections.

2. CIRCUIT DESIGN

2.1. Low-Noise AmplifierIllustrated in Figure 2 is a simplified schematic of dual-band

LNA [4] that adopts a cascode amplifier configuration. Transis-

tors M1 and M2 are the LNA input where M1 is for the 868/915

MHz band input and M2 is for the 2.45 GHz band. When the

LNA is working in 868/915 MHz band, the PD-B1 (power-

down of the 1st band) and PD-B2 (power-down of the 2nd

band) are turned ON and OFF, and vice versa. The advantage of

this configuration is that the input matching of each band can be

optimized independently. This is an important aspect because

the LNA input matching network quality factor is a key factor

in determining the gain, noise, linearity, and sensitivity to com-

ponent variations [4].

To achieve simultaneous noise and input matching at low

power level, an external capacitor (Cex1 or Cex2) is added at

each input. This capacitor together with degeneration inductor

Ls will bring the noise optimum point close to the input match-

ing optimum point. Not only that, it also gives more freedom to

optimize the input matching [5]. An off-chip inductor (Lg1 or

Lg2) in series with each input tunes out the input capacitance.

With single-band LNAs, the size of transistor M3 is typically

chosen to be equal to that of M1 in order to minimize the para-

sitic capacitance associated with the layout wires at node X.

However, in this dual-band LNA, as two parallel input transis-

tors increase the capacitance at that node, the size of M3 should

therefore be different. From simulation it was found that when

the size of M3 is equal to 60% of M1, the NF in both bands will

be optimized.

A switched resonator formed by inductors L1, L2, and M4 is

utilized for band selection and gain control of the LNA [4, 6].

By controlling Vctrl from low to high voltage levels, the equiva-

lent inductance (Leq) will change from Leq1 to Leq2 (where Leq1

� L1 þ L2 and Leq2 � L2). Consequently, a switched resonator

will work at low and high bands.

The values of L1, L2, and M4 are chosen by considering the

LNA gain requirement. In this work, the design target for LNA

gain and current dissipation are 12 dB and 1 mA, respectively.

The inductor L2 is determined by tuning the circuit to 2.45

GHz. The size of M4 will be chosen so that the gain is at least

12 dB when turned ON. The inductor L1 is added to tune the

circuit at the 868/915 MHz band with M4 turned OFF.

The gain control function is set as the same with the band

selection. In other words, when M4 is ON, the switched resona-

tor will resonate at the 2.45 GHz band. When M4 is OFF, the

LNA output is tuned to the 868/915 MHz band, and the gain at

the 2.45 GHz band is decreased. The same concept is also appli-

cable for the 868/915 MHz band.Figure 1 Dual-band receiver architecture

Figure 2 Schematic of the dual-band LNA

164 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 52, No. 1, January 2010 DOI 10.1002/mop

2.2. I/Q Downconversion MixerAn I/Q downconversion mixer consists of two identical double-

balanced Gilbert cell type without a tail current as shown in Fig-

ure 3. The tail current is removed to improve the linearity

because of the limited voltage headroom. It is obvious that with-

out the tail current, the voltage headroom will be improved by

the Vdsat of a MOS transistor if a simple current source is used.

In Figure 3, transistors M1 and M2 form a differential pair with

the gate of M2 ac grounded by an on-chip capacitor (Cbypass).

This configuration helps to reduce the total power consumption

of the receiver because the single-ended LNA dissipates half of

the current of the differential amplifier. Transistors M3–M6 are

switching transistors driven by a differential LO signal. The LO

frequency is changed with the input frequency in order to keep

the IF at 2 MHz. The downconversion mixer has two gain

modes which are formed by resistors R1, R2, and PMOS trans-

mission gate Mc.

This mixer can operate at two frequency bands without

changing any device parameters, because the required gain and

NF are different at two frequency bands as described later.

3. EXPERIMENTAL RESULTS

The dual-band RF receiver front-end was fabricated based on

TSMC 0.18 lm CMOS technology with a die-size of 1.2 mm2

including ESD pads. The testing board was built by directly bond-

ing the die on a four-layer FR4 substrate as shown in Figure 4. A

differential to single-ended conversion process at the LO ports and

output IF ports were done by a commercial passive balun.

Figure 5 shows the measured voltage conversion gain and

NF of the implemented RF receiver front-end. As can be seen

from Figure 5, the implemented RF receiver front-end shows a

maximum voltage conversion gain of 31.5 and 22 dB at the

868/915 MHz and the 2.45 GHz bands, respectively. The

obtained NF results are 6 and 9 dB at low and high bands,

respectively. As mentioned in the IEEE 802.15.4 standard [1],

the required sensitivity of the 868/915 MHz and the 2.45 GHz

bands are �92 and �85 dBm, respectively. Assuming the low-

IF section will be shared for both frequency bands, the required

Figure 3 Schematic of the downconversion mixer

Figure 4 Implemented RF receiver front-end test board. [Color figure

can be viewed in the online issue, which is available at www.

interscience.wiley.com]

Figure 5 Measured voltage conversion gain and NF of the imple-

mented RF receiver front-end. [Color figure can be viewed in the online

issue, which is available at www.interscience.wiley.com]

TABLE 1 Summary of RF Receiver Performances

868/915 MHz Band 2.450 GHz Band

Simulated Measured Simulated Measured

Maximum gain (dB) 32 31.5 23 22

Minimum gain (dB) 13 12 12 11.5

NF at 2 MHz (dB) 5.7 6.0 8.5 9

P-1 dB at high

gain (dBm)

�25 �25 �20 �20

P-1 dB at low

gain (dBm)

�10 �5 �10 �5

HP3 at high

gain (dBm)

�16 �15 �16 �15

Input return loss (dB) �20 �14 �15 �11

Supply voltage (V) 1.8 1.8

Current dissipation

(mA)

3.0 3.0

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 52, No. 1, January 2010 165

gain and NF of the RF front-end operating in the 2.45 GHz

band are 10 and 3 dB lower than that in the 868/915 MHz band.

Therefore, the implemented RF receiver can satisfy the required

specifications for a multiband IEEE 802.15.4 standard.

The other measured and simulated results are summarized in

Table 1. From Table 1 we can see that the measured results are

very close to the simulated results.

4. CONCLUSION

A low-power, dual-band RF receiver front-end for IEEE

802.15.4 standard-based sensor node applications is reported and

fabricated in 0.18 lm CMOS technology. The RF receiver with

a 1.2 mm2 die size consumes 3 mA under a supply voltage of

1.8 V at all bands. The RF receiver uses a low-IF architecture

with two separate LNA inputs for the 868/915 MHz and the

2.450 GHz bands and a single-ended input downconversion

mixer. The RF receiver shows a maximum conversion gain of

31.5 and 22 dB, and an NF of 6 and 9 dB at the 868/915 MHz

and 2.45 GHz bands, respectively.

ACKNOWLEDGMENTS

This work was supported by the IT R&D program of MIC/IITA

(2005-S-106-2, Development of Sensor Tag and Sensor Node

Technologies for RFID/USN), Republic of Korea.

REFERENCES

1. IEEE 802.15.4, Wireless medium access control (MAC) and physi-

cal layer (PHY) specifications for low-rate wireless personal area

networks (WPANs), IEEE, New York, 2006.

2. M. Zargari, et al., A single-chip dual-band tri-mode CMOS trans-

ceiver for IEEE 802.11a/b/g WLAN, IEEE J Solid State Circ 39

(2004), 2239–2249.

3. H. Hashemi and A. Hajimiri, Concurrent dual-band CMOS low

noise amplifiers and receiver architectures, In: VLSI Circuits Sym-

posium Digest of Technical Papers, 2001, pp. 247–250.

4. Z. Li, R. Quintal, and K.K. O, A dual-band CMOS front-end with

two gain modes for wireless LAN applications, IEEE J Solid State

Circ 39 (2004), 2069–2073.

5. T.-K. Nguyen, et al., CMOS Low noise amplifier design optimiza-

tion techniques, IEEE Trans Microwave Theory Tech 52 (2004),

1433–1442.

6. S.-M. Yim and K.K. O, Demonstration of a switched resonator

concept in a dual-band monolithic CMOS LC-tuned VCO, In: Pro-

ceedings of IEEE Custom Integrated Circuits Conference, 2001,

pp. 205–208.

VC 2009 Wiley Periodicals, Inc.

COMPACT SUB-WAVELENGTHMICROSTRIP BAND-REJECT FILTERBASED ON INTER-DIGITALCAPACITANCE LOADED LOOPRESONATORS

Yi Peng and Wen-Xun ZhangState Key Laboratory of Millimeter Waves, Southeast University,Nanjing 210096, People’s Republic of China; Correspondingauthor: yipeng@seu.edu.cn

Received 27 April 2009

ABSTRACT: In this article, the microstrip band-reject filters based oninter-digital capacitance loaded loop resonators (IDCLLRs) are

proposed by using meandered microstip-line, which enhances the

coupling to the resonators and provides more structural parameters for

flexible design. First, an IDCLLR-based single-stage meanderedprototype of band-reject filter has been designed and fabricated. The

measured frequency response shows 2.2% relative stop-bandwidth (�10dB) with �21 dB maximal insertion-loss at 3.34 GHz. Furthermore, theresonant frequencies versus inter-digital length are studied. Finally, an

IDCLLR-based three-stage meandered prototype of band-reject filterwas designed and fabricated too, the measured relative stop-bandwidth

(�10 dB) is 6.2% and the insertion-loss at center frequency approaches�42 dB. VC 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett

52: 166–169, 2010; Published online in Wiley InterScience

(www.interscience.wiley.com). DOI 10.1002/mop.24885

Key words: meandered microstrip; band-reject filter; the inter-digital

capacitance loaded loop resonator (IDCLLR); split ring resonator (SRR)

1. INTRODUCTION

The band-reject filter is an indispensable component in recent

communication systems for noise reduction and interference

rejection. Though various microstrip ring-resonators [1] had

been popularly used in practice, their dimensions in scale of full

resonant wavelength kg do not meet the demand of miniaturiza-

tion for mobile system applications. Recently, the sub-wave-

length resonators (SWRs), such as split ring resonators (SRRs)

[2] and related structures [3] have been introduced into the

band-reject filter [4–6]. SRR consisting of a pair of concentric

metallic rings with a split etched in opposite sides as Figure

1(a) was originally proposed by Pendry, its size can be much

smaller than its resonant wavelength due to its large effective

capacitance. Other SWRs may be derived from the basic geome-

try of the SRR, for example, broadside coupled-SRR (BC-SRR)

[7], capacitance loaded loop resonator (CLLR) [8], and spiral

resonator (SR) [3], etc. In this article, a novel SWR, namely

inter-digital capacitance loaded loop resonator (IDCLLR) as

shown in Figure 1(b) is proposed to design band-reject filter.

IDCLLR consists of only single metallic ring but with inter-

digital gap to enhance effective capacitance and provide more

designable parameters for structure minimization.

In the previous design of a SWR-based band-reject filter,

SWRs are always placed by both side of a straight microstrip

line as array [4–6]. The band-rejection behavior is resulted from

inductive coupling between line and SWRs: the forward current

on the line induces to the loops of SWRs at first, then coupled

back to the line due to resonance on loop, and results in back-

ward current on the line. However, the structure of multiple-

stage SWRs is necessary for a rigorous rejection response, since

the weak coupling in single-stage structure.

A stronger coupling mechanism must benefits to reduce the

stages and sizes of filter. In Ref. 9, an open loop rectangular res-

onator has been embedded into a U-bend of meandered micro-

strip line to achieve a better coupling between resonator and

transmission line, the similar structure is applied for IDCLLR-

based meandered band-reject filter in this article. A comparative

study for frequency responses between IDCLLR-based and

SRR-based single-stage meandered band-reject filter, and among

different inter-digital lengths are performed, respectively. In

addition, an IDCLLR-based three-stage meandered band-reject

filter has been fabricated to further enhance the maximal rejec-

tion level is presented.

2. SINGLE-STAGE IDCLLR-BASED MEANDEREDBAND-REJECT FILTER

The quasi-static resonant angular frequency x0 ¼ 1=ffiffiffiffiffiffiffiffiffiffi

L0C0

pof

IDCLLR depends on the total inductance L0 of the IDCLLR

166 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 52, No. 1, January 2010 DOI 10.1002/mop