A Wide-Band Power-Efficient Inductive Wireless Link for Implantable

IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—I: REGULAR PAPERS, VOL. 54, NO. 10, OCTOBER 2007 2211

A Wide-Band Power-Efficient InductiveWireless Link for Implantable Microelectronic

Devices Using Multiple CarriersMaysam Ghovanloo, Member, IEEE, and Suresh Atluri, Member, IEEE

Abstract—This paper presents a novel inductive link for wirelesstransmission of power and data to biomedical implantable micro-electronic devices using multiple carrier signals. Achieving higherdata bandwidth without compromising the power efficiency is thedriving force behind using multiple separate carriers. Two sepa-rate pairs of coils have been utilized for inductive power and for-ward data transmission, plus a pair of miniature antennas for backtelemetry. One major challenge, however, is to minimize the inter-ference among these carriers especially on the implantable side,where size and power are highly limited. Planar power coils withspiral shape are optimized in geometry to provide maximum cou-pling coefficient . The data coils are designed rectangular in shapeand wound across the power coils diameter to be oriented per-pendicular to the power coil planes. The goal is to maximize datacoils direct coupling, while minimize their cross-coupling with thepower coils. The effects of coils geometry, orientation, relative dis-tance, and misalignments on the coupling coefficients have beenmodeled and experimentally evaluated.

Index Terms—Coupling, implantable microelectronic devices(IMDs), inductive link, neuroprosthesis, planar spiral coils,telemetry.

I. INTRODUCTION

AN INDUCTIVE link between two magnetically coupledcoils that constitute a transformer is so far the only viable

solution to wirelessly energize high performance implantablemicroelectronic devices (IMDs) with high power requirementssuch as neuromuscular stimulators, cochlear implants, and vi-sual prostheses [1]–[7]. These IMDs are either battery-less andshould be continuously powered from an external portable bat-tery, or have miniature rechargeable batteries that should be in-ductively charged on a regular basis. In both cases, the inductivepower transmission should be very efficient to maximize batterylifetime, minimize the size of the battery, and eliminate over-heating of the surrounding tissue due to heat dissipation withinthe implant or within the surrounding tissue as a result of sur-passing the exposure limit to electromagnetic field [8].

IMDs that substitute sensory modalities such as vision andhearing also need a wide-band forward wireless link to transfersizeable amounts of real-time information from external artifi-cial sensors, such as a camera or microphone, to a large number

Manuscript received July 20, 2006; revised November 23, 2006 and March26, 2007. This paper was recommended by Associate Editor B. Zhao.

M. Ghovanloo is with the GT Bionics Laboratory, Department of Electricaland Computer Engineering, Georgia Institute of Technology, Atlanta, GA 30332USA (e-mail: [email protected]).

S. Atluri is with Integrated Device Technologies Incorporated, San Jose, CA95138 USA (e-mail: [email protected]).

Digital Object Identifier 10.1109/TCSI.2007.905187

of neurons in the central nervous system (CNS). These IMDsusually communicate with the CNS by driving tens to hundredsof stimulating sites through multiple parallel channels [4]–[7].As a result, wide-band data transmission is another requirementfor the inductive wireless link. The link should be robust enoughnot to be affected by patient’s motion artifacts, minor coil mis-alignments, and external noise and interference. To achieve this,a back telemetry link is needed to form a closed-loop implantpower regulation mechanism and improve data integrity throughhandshaking protocols [9]. The back telemetry link can alsobe used for measuring the stimulating sites impedance in situ,which is important in monitoring sites chemical stability overtime [10]. Back telemetry is also useful in recording the neuralresponse to stimulus pulses for accurate electrode placement,stimulation parameter adjustments, and for research purposes.

The wireless link operating frequency, also known as thecarrier frequency, is one of the most important parameters of theimplantable biomedical system. Traditionally, a single carrierfrequency has been used for inductive power and data transmis-sion from the outside world towards the implanted device [4],[11]–[15]. Achieving high power-transmission efficiency, highdata-transmission bandwidth, and coupling insensitivity usingthe traditional single carrier method would be very challenging,if not impossible, because of the conflicting constraints that areinvolved in achieving high performance in two or more of theabove system requirements. For example, increasing the carrierfrequency could result in a wider bandwidth for forward datatransmission. However, it degrades power transmission effi-ciency due to more power absorption and deposition in the tissue[16] and more power dissipation in the external and internalpower conditioning blocks [17]. Establishing a back telemetrymechanism by changing the secondary coil loading [load shiftkeying(LSK)] relies on the subsequent changes in the powercarrier amplitude or current on the primary side [9], [18]. Thismethod is popular in radio frequency identification (RFID) ap-plications, even though it results in loss of power efficiency [19].

Researchers have recently proposed utilizing two carriers toachieve a better performance by separating some of the afore-mentioned functions. In [20] and [21] the inductive link is usedfor power and forward data transmission, while a high frequencycarrier is utilized for back telemetry. In [22] and [23] a high fre-quency carrier is used for bidirectional telemetry and the induc-tive link is solely used for powering or recharging the implant.Finally in [24]–[26] the inductive link is used for power andback telemetry, while a separate link is proposed for forwarddata transmission.

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2212 IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—I: REGULAR PAPERS, VOL. 54, NO. 10, OCTOBER 2007

Fig. 1. Block diagram of a modern neuroprosthetic system with both wireless neural recording and stimulation capabilities, which utilizes the proposed multicar-rier wireless link with emphasis on the blocks that are directly related to this link. The upper, middle, and lower rows of blocks are dedicated to power, forwarddata, and back telemetry functions, respectively. The inductive part of the wireless link is enclosed in a dashed box.

In the rest of this paper, we describe a novel solution for uti-lizing multiple independent carrier signals to resolve the afore-mentioned conflicts between three major functions; power, for-ward data, and back telemetry, in design of a wireless linksfor high performance IMDs. Section II provides an overviewof an IMD system that utilizes this multicarrier wireless link.Section III reviews the theoretical basis and important consider-ations behind the design of the multicarrier inductive link alongwith numerous simulation results. Section IV examines the ef-fects of misalignment, which is an important aspect of the induc-tive links in IMDs. Measurement results with the multicarrierwireless link are depicted in Section V followed by concludingremarks in Section VI.

II. MULTICARRIER WIRELESS LINK

We propose utilizing three carrier signals at three differentfrequencies and amplitude levels which are properly selectedfor the specific functions expected from the wireless link [27].

a) A low-frequency MHz high-amplitudeV carrier for power transmission from the

external energy source (battery) to the implant.b) A medium-frequency MHz medium-

amplitude V carrier for forward datatransmission from the external sensor to the implant.

c) A high-frequency MHz low-amplitudeV carrier for back telemetry from the implant

to the external part of the system.Employing multiple carriers would help to effectively isolate

many of the conflicting requirements in design of the wirelesslink with only one carrier. In a multicarrier system, individualparameters of every carrier can be optimized for its associatedfunction(s) and requirements regardless of the other carriers. Onthe other hand, every individual carrier requires its dedicatedcoils or antennas, which could potentially add to the implant sizeand complexity. The main challenge in using multiple simulta-neous carriers, however, is the interference between differentcarrier signals. Among interference issues, the most importantone is eliminating the strong power carrier interference with datacarrier on the implantable receiver side, where the power budgetand size are extremely limited. To overcome this issue we haveadopted the following remedies.

1) Utilizing two individual pairs of coils dedicated to powerand forward data transmission. The geometry and orienta-

tion of these coils are selected such that they occupy a smallspace, direct coupling within each pair is maximized, andcross-coupling between the two pairs is minimized.

2) Instead of amplitude shift keying (ASK), which is themethod used in most similar applications [1], [4], [14],[20], [21], the data carrier is frequency shift keyed (FSK)to be more robust against noise and interference [15].

3) The forward data carrier frequency is chosen 5 10times higher than the power carrier frequency to pro-vide enough spacing between the carrier harmonic compo-nents in the frequency domain. This would allow reliabledetection of the forward data bits on the implant receiverside with proper tuning, filtering, and utilization of theFSK modulation and demodulation technique [15]. Sim-ilarly the back telemetry carrier frequency is selected10 50 times higher than .

4) The forward data carrier amplitude, , is adjustable onthe transmitter side based on the quality of the receiveddata on the implant side. can be increased in responseto an increase in the receiver bit-error rate (BER).

Fig. 1 shows the block diagram of a modern neuroprostheticsystem that utilizes the proposed multicarrier wireless link withemphasis on the blocks that are directly related to the link. Theseblocks provide a bidirectional wireless interface between the ex-ternal artificial sensors and internal microelectrodes that interactwith the neural tissue. Such prosthesis would be capable of bothstimulating the neural tissue through multiple channels at highrates as well as recording several channels of the neural responsethrough its back telemetry link, while maintaining high powertransmission efficiency. The upper row of blocks in Fig. 1 is re-sponsible for power transfer and conditioning across the skin. Aclass-E power amplifier that is controlled by the external con-trol unit drives , which is the primary power coil. Power isinduced into , which forms an LC tank circuit, resonatingat to boost the received power carrier. is followed by afull-wave rectifier and a low dropout regulator to supply the restof the implant [17], [28].

The middle row of blocks in Fig. 1 is dedicated to forwarddata transmission. Forward data is a serial data bit stream thatmainly includes various stimulus parameters such as site ad-dress, amplitude, phase, and timing information. Any digitalmodulation scheme that supports wide-band serial data trans-mission with robustness against noise and interference can be

GHOVANLOO AND ATLURI: WIDE-BAND POWER-EFFICIENT INDUCTIVE WIRELESS LINK FOR IMDS 2213

utilized in the forward data path. Unlike traditional single car-rier links, low quality factors and a high modulation indexes canbe chosen for this link to achieve low BER at high data rateswithout degrading the system power efficiency. Our choice wasa phase coherent frequency shift keying (pc-FSK) method de-scribed in [15]. Hence, consists of two frequencies,and , which represent logic “0” and logic “1”, respectively.The pc-FSK carrier passes through an adjustable gain stage thatdrives , the primary data coil. is part of a series-parallelLC tank circuit, which provides a wide-band low-Q inductivelink that passes the high-rate pc-FSK carrier with minimum dis-tortion [see Fig. 12(a), shown later] [15]. On the implant side,

picks up the forward data carrier signal, which then passesthrough a high-pass filter (HPF) to block any residual low-fre-quency components resulted from power carrier interfer-ence. Finally, a pc-FSK demodulator converts the data carrierinto a serial data bit stream and a constant frequency clock [15].

Those blocks in the lower row of Fig. 1 are involved in backtelemetry of various information such as the implant operationaland power supply status, site impedance values, handshakingsignals, and recorded neural response to the external part of thesystem. Once again, the designer is free to choose a proper mod-ulation scheme depending on the system requirement withoutaffecting other functions. We chose pulsewidth modulation(PWM) to improve the robustness of the back telemetry linkagainst noise and interference [29]. The PWM signal passesthrough a voltage controlled oscillator (VCO) that generates theback telemetry carrier in the industrial-scientific-medical(ISM) band. VCO is designed to operate at low voltages inorder to minimize implant power consumption. The VCOoutput drives a wide-band miniature patch antenna, , whichis tuned at [23]. The back telemetry signal in the ISM-bandis picked up outside of the body by an FM receiver equippedwith a larger patch antenna, . Finally, a PWM demodulator,which is a synchronized timer/counter, recovers the backtelemetry signal, while converting it to digitized values [29].

The rest of this article focuses on the inductive part of the pro-posed multicarrier wireless link, which is enclosed in a dashedbox in Fig. 1. A brief reference to a possible implementation ofthe microwave back telemetry link is included in Section V-C.

III. INDUCTIVE LINK GEOMETRIC CONSIDERATIONS

The inductive part of the developed wireless link, enclosedin a dashed box in Fig. 1, consists of four coils in two pairs:

for power and for forward data transmission.The self and mutual inductances among these coils are criticalparameters that affect the performance of the wireless link.

A. Self and Mutual Inductance

Self inductance is the ratio of the magnetic flux generatedin an area enclosed by a conductor loop to the current passingthrough the loop. According to [30], for the condition of a loopwith , where and are the radii of the wire and theloop, respectively, self inductance can be approximated by

(1)

In circular coils with turns, if the coil length is muchsmaller than , the self-inductance is approximately ,where is the self-inductance of a single-turn loop derivedin (1). Whereas, for the case of a planar spiral coil havingturns with different radii the overall self-inductance should be calculated from

(2)where if , and otherwise [30].

The mutual inductance between two conductor loops, ,depends on the proportion of the magnetic flux generated by oneloop that passes through the other loop (flux coupling). There-fore, highly depends on loops geometries, relative orien-tation, and magnetic properties of the medium. The mutual in-ductance of two circular coils with radii , , center to centerdistance , and lateral misalignment is

(3)

where and are the Bessel functions of the zeroth and firstorder, respectively [30].

The mutual inductance between two coils is often normalizedwith respect to their self inductance to reach a qualitative senseof how strongly they are coupled. The coupling coefficientbetween two coils with self inductances and is defined as

(4)

Unfortunately the closed-form analytical solutions for cal-culating self and mutual coil inductances either become toocomplicated to be practical in design of the inductive links orlose their accuracy for multiple coils with various geometriesand orientations. Therefore, one should either use tabulated pa-rameterized equations [31] or coil analysis software such asFastHenry-2 [32], HFSS, or Maxwell [33]. FastHenry-2 is aquasi-static EM solver for computing frequency-dependant selfand mutual inductances as well as resistances of arbitrary 3-Dconductive structures. To perform electromagnetic (EM) mod-eling, simulations, coil analysis, and optimizations we devel-oped a MATLAB code that generates 3-D geometry files oftwo or more coils for FastHenry-2. Impedance matrixes gener-ated by FastHenry-2 can then be used in LC circuits in SPICEfor time- and frequency-domain simulations on carrier signals.This flexible approach allowed us to optimize coil designs bysweeping the coils’ geometrical or electrical parameters and ob-serve the effects of parameter changes on the performance of theinductive wireless link.

B. Power Coils

The inductive link power transmission efficiency is thekey parameter in design of the power coils, which should be


Fig. 2. To maximize the coupling coefficient k between two identical planarspiral coils, the outer radius R , the inner radius R , the spacing betweentwo adjacent turns �, and the number of turns N should be selected such thatR =R = 0:34.

maximized. Other important factors are the coils size, sepa-ration, misalignment effects, and surgical considerations. Ac-cording to Ko et al. [12] in a simplified scenario, the parametersaffecting the link power efficiency are

(5)

where and are quality factors of the primary and sec-ondary coils, respectively ( and in Fig. 1), and is thecoupling coefficient between the two coils. and can beincreased by reducing coils parasitic resistance and forming in-ductive–capacitive (LC) tank circuits tuned at . However, themost effective parameter, , depends on the coils self and mu-tual inductances according to (4), which in turn depend on theirgeometry, separation, and orientation.

We chose circular, planar, spiral geometry for the power coils,which is one of the most widely used methods in design of theinductively powered IMDs. The spiral coils are fairly robustagainst lateral misalignments and their optimal design has beencovered in prior literature [30]. According to Zierhofer et al., aspiral coil can be described with three parameters: is theouter radius, is the inner radius, and is the spacing be-tween every two adjacent turns of the coil. Obviously, shouldalways be greater than , the diameter of the coil wire. Thenumber of turns in a spiral coil can then be calculated from

(6)

The closed-form equations suggest that the coupling coeffi-cient should not be a strong function of in circular coils sinceby increasing both numerator and denominator of (4) increaseat almost the same rate. In case of spiral coils, however, our sim-ulation results in Fig. 2 showed that as was increased, cou-pling coefficient kept on increasing until ,which is in agreement with in [30]. Therealso needs to be a compromise between and the coil wire di-ameter, , which also affects the coil parasitic resistance andeventually its factor.

Fig. 3. Coupling coefficient between two planar spiral coils with R =

10mm as a function of R and distance d between the coils. It can be seenthat there is an optimal value for R at every distance that maximizes k.

Another design parameter that can affect is the outerdiameter of and coils with respect to their distance .According to (3) increasing and equally help in in-creasing . However, there is a size constraint over ,which is imposed by the maximum allowable size of the im-plant. Therefore, a design guideline is to choose (in spiral coils) as large as the implant size allows. To find thebest , we limited to 10 mm and changedfrom 8 to 24 mm for different values of , while maintaining

. Fig. 3 shows how the optimalincreases with and changes in a wider range.

C. Data Coils

The geometry and orientation of the data coil pairis very important because the power carrier amplitude is usu-ally much larger than the data carrier amplitude on both sides.Therefore, unless the interference of the power carrier over datacarrier is adequately suppressed on the receiver side (across ),demodulation of the data carrier and detection of the forwarddata bits would not be possible. Geometry and orientation ofthe data coils should be designed in a way that they occupy asmall space, have a large direct coupling , and very smallcross-coupling with the power coil pair ( , , , andin Fig. 1).

Our approach is to take advantage of the relationship betweenand the relative orientation of two coaxial conductive loops

to minimize undesired mutual couplings. Soma et al. analyzedthe effects of coils misalignments in [34] and concluded that ifone of the coils is tilted by an angle , their mutual inductancereduces by a factor of

(7)

where is the relative center to center distance between thecoils, and is the angle between their planes. Therefore, parallelcoils provide maximum coupling while those at right angles willbe weakly coupled, especially if they are symmetrical as well.Fig. 4 is the result of a simulation that illustrates the variationof between two identical rectangular coils when one coil isrotated from 0 in parallel to the second coil (position A) by90 . The coils have minimum when they are at right angle


Fig. 4. Coupling coefficient between two identical rectangular 1-turn coils inparallel [A], perpendicular [B], and in the same plane [C] versus the relativeangle between their planes. The relative distance between coils is constant (10mm) in all three positions.

(position B). Now keeping the first coil at 90 , the second coilis rotated from 0 to . The rotations are done pivotal to oneside of each coil such that the original vertical distance betweenthe coils is not altered. It can be seen that improves when bothcoils are in the same plane (position C).

Another guideline in design of the data coils is that the receiverdata coil shouldbe implemented within the same space thatoccupies inside the implant. Thus, we chose to be rectangularin shape and wound across the diameter of to give it the max-imum possible length . The external data coilwas chosen with the same shape, wound across to be perpen-dicular to both power coils but parallel to in order to achieve afair amount of coupling between two parallel wires on thecloser edges of and rectangular coils. Thin wires can beused in construction of and since they do not need to carrylarge currents as opposed to and . It is important especiallyfor not to notably increase the size of the implant.

Fig. 5 shows the final configuration of the power and datacoils rendered in FastHenry-2. It can be seen that data coils inthis design are located in position C in Fig. 4. Even though datacoils could have a higher coupling if they were held in positionA, when considering the cross-coupling between each data coiland the power coils, position C proves to be the best option. Inthe model shown in Fig. 5, mm and the maximumdistance between the power coils is assumed to be about 10 mm.Therefore, we can conclude from Fig. 3 that the optimum radiusfor the external power coil is about 20 mm. The primaryand secondary data coils are then considered to be rectangularand measure 42 mm 5 mm and 21 mm 1 mm, respectively,wound across the diameter of and . Table I summarizethe self inductance as well as direct and cross-coupling coef-ficients between these coils, simulated by FastHenry-2. It canbe seen that , which according to (5) has a strong effect on

, is the largest coefficient and the desired data coils couplingcoefficient is significantly higher than all other undesiredcross-coupling coefficients.

Since data coils are wound around the high current power coils(see Fig. 5), yet another consideration in minimizing the power-data coils cross-coupling is to zero out the sum of power coil cur-rents that pass through the data coil loops. This would eliminate

Fig. 5. 3-D rendering of the inductive part of the multicarrier wireless link con-sisting of a pair of planar spiral coils in parallel for power transmission and apair of rectangular coils on the same plane for forward data transmission, whichare vertical to the power coil planes. Table I shows the self-inductance and mu-tual coupling between these coils.

TABLE ISELF-INDUCTANCE AND COUPLING COEFFICIENTS FOR

POWER AND DATA COILS

any electromotive force (EMF) that could be induced from thepower coils into the data coils. This is the opposite of what is de-sired in current-sense transformers [25]. This requirement wassatisfied by winding the data coils across the entire diameter ofthe power coils, making sure that they encompass equal numberof wires passing current in two opposite directions, thus zeroingthe net current and canceling out their interfering EMF.

D. Simultaneous Data and Power Circuit Simulation

To evaluate the interference between power and forward datacarriers in time and frequency domains, circuit simulationswere performed with the inductive wireless link LC-tank cir-cuits, shown in Fig. 1. In these simulations, we used the coilparameters of Table I. Power is transmitted at kHzand V using stagger tuning [13]. FSK modulation isused to transmit data at high rate using MHz and

-V data carrier as explained in [15]. The transient andfrequency responses of the inductive wireless link in Fig. 6 showthe insignificant effect of the power carrier interference with thereceived FSK data carrier despite the exaggerated amplitude ofthe power carrier. This is the result of proper coil design, largecarrier frequency separation, and bandpass filtering effect of thetuned LC-tanks. Further high-pass filtering on the received datacarrier can be added on-chip, as shown in Fig. 1.


Fig. 6. (a) Transient and (b) frequency responses of the inductive part of thewireless link in Fig. 1 showing the small power carrier interference on the re-ceived FSK data carrier as a result of the measures explained in Section III.

IV. EFFECTS OF COIL MISALIGNMENTS

Coil alignment in inductive links is usually enforced by usingpermanent magnets in the center of the internal and externalcoils as a mean to align the external coil with the implanted oneand hold it in place across the skin. The effects of coil misalign-ments are important in design of inductive links for implantabledevices because the implant user might remove the external coiland does not place it back in the exact same location. The coilsmight also move or vibrate as the user walks or jogs. Differenttypes of misalignments between two circular coils have beenanalyzed in [34]. Planar spiral coils consist of several circularcoils and follow the same trend with less sensitivity due to in-teractions between every turn in one coil and all other turns inthe other coil [30].

Fig. 7 shows simulation results using FastHenry-2 andMATLAB on how , , and change as a result of coilslateral misalignments along X- and Y-axes at constant

mm [36]. The spiral power coils show a high direct couplingwhen they are perfectly aligned. In Fig. 7(a), shows similarbell-shaped drops along X- and Y-axes misalignments due tothe power coils symmetry. On the other hand, and in

Fig. 7. Direct and cross-coupling coefficients between coils in Fig. 5 versusX-axis and Y-axis misalignments a d = 10 mm. (a) k . (b) k . (c) k .(d) Overlapping k and k at X = 0 mm to observe the range of acceptablemisalignments along Y-axis.

Fig. 7(b) and 7(c) shows that data coils are far less sensitive tomisalignments along the X-axis (along their length) compared to


Fig. 8. Measurement setup showing the inductive part of the multicarrier wire-less link with a pair of planar spiral coils for power transmission and a secondpair of coils wound across the power coils for forward data transmission. Thesecoils are made of Litz wire (Set-3 in Table II).

TABLE IIHANDMADE POWER AND DATA COILS SPECIFICATIONS

All inductances are in microhertz and all sizes are in millimeters.Width of the rectangular data coil,Length of the rectangular data coil.Number of Litz wire strands in power coil.Number of Litz wire strands in data coil.

the Y-axis (perpendicular to their X-Z plane). Because in X-axismisalignment, the distance between data coils does not changeand they still hold their symmetry with respect to the power coils.

It can be concluded from these simulations that Fig. 5 coilconfiguration can very well handle coil misalignments along theX-axis. However, along the Y-axis, the misalignments are accept-able only as long as . This can be seen more clearlyby overlapping and surfaces and looking at a certaincross-section of the overlapped curves. This is shown in Fig. 7(d),which indicates that with the coil dimensions used in these sim-ulations, lateral misalignments up to 2 mm in the Y directionare acceptable. In order to reduce data coils sensitivity against Ymisalignments, one possible solution is to add a second pair offorward data coils in Y-Z plane perpendicular to the first pair (and in X-Z plane) as well as the power coils ( and in X-Yplane). The second pair of data coils, which can alternatively beutilized for back telemetry, should be used for forward dataonly when the BER through the first pair is not satisfactory.

V. MEASUREMENT RESULTS

Three sets of prototype data and power coil pairs similar toFig. 5 were hand made using coil copper wire gauge 22 andLitz wire gauge 46 (40 strands for power and 7 strands fordata coils) for experimental measurements and comparison withsimulation results. Fig. 8 shows one of these test setups andTable II summarizes the geometrical and electrical specifica-tions of these coils. Using considerations in Section III, the ex-ternal coils inner and outer radii were designed for the optimalrelative distances, , of about 10 mm for sets 1 and 3, and 18 mm

Fig. 9. Network analyzer was used to measure fabricated coils direct and cross-coupling coefficients, while changing their relative distance d.

for Set-2, while the size of all internal coils was limited by theimplant size to 22 mm in diameter. The number of turns forthe data coils was increased to 3 from only 1 turn in Fig. 5 toimprove and to better cope with the coil misalignment ef-fects. However, we did not optimize the number of turns in datacoils. Coils self inductances were measured during fabricationusing a precision LCR meter at 100 kHz (Instek LCR-819).

A. Coupling Coefficient Measurements

A network analyzer (Agilent E5071B) was used to performtwo-port measurements on every pair of coils in the inductivelink from 300 kHz to 10 MHz and obtain the S-parameters, asshown in Fig. 9. Meanwhile, we changed the relative distance

between transmitter and receiver coils to observe and com-pare the effects of coils distance variations. The S-parameterswere then converted to Z-parameters using Agilent AdvancedDesign System (ADS). The two-port Z-matrix includes the selfand mutual inductances, which can be extracted from (8) at eachoperating frequency, and substituted in (4) to calculate

(8)

The measured and simulated (FastHenry-2) values of fordirect and cross-couplings between power and data coils areshown in Fig. 10, while is changed from 10 to 30 mm [36].It can be seen that there is a very good agreement between sim-ulated and measured values of and . Coupling betweenthe power coils, , is strong at mm equal to 0.228 and0.15 for sets 1 and 2, respectively. If and values are opti-mized for maximum power efficiency, it can be concluded from[30] that can be achieved for mm. Also asexpected, there is a sharper drop in for the smallest trans-mitter coil (Set-1) than the largest one (Set-2). While the largesttransmitter sends the EM power deeper into the tissue.

Fig. 7(c) shows that in order to achieve minimum cross-cou-pling, all coils should be symmetrical and axially aligned. How-ever, reaching close symmetries in handmade coils is difficult.It is possible through to compensate some of the effects of theseasymmetries with intentional misalignments to practically min-imize . Therefore, we adjusted the height of the receiver coilto experimentally find the Y-axis misalignment at whichwas minimal. Fig. 10(c) shows versus vertical misalignment


Fig. 10. Measured and simulated values of coupling coefficient between(a) power coils, k and (b) data coils, k in sets 1 and 2. (c) Y-axis misalign-ment between coils to find minimum cross-coupling for set-2 (d) measured andsimulated values of cross-coupling coefficient k between transmitter powercoil L and receiver data coil L .

Fig. 11. Measured direct and cross-coupling coefficients between set-3 powerand data coils in Table II, all on a semi-logarithmic scale.

for Set-2 with minimum achieved at mm, which fol-lows the same trend as Fig. 7(c) simulation. With these adjustedheights, Fig. 10(d) compares simulated and measured valuesfor sets 1 and 2. In these measurements, the fact that is verysmall within the best accuracy of our manual measurement setuphas resulted in some discrepancies between simulated and mea-sured values. Nevertheless, the measurement and simulation re-sults are in the same range.

To make comparison between coupling coefficients easier,Fig. 11 shows the results of measured direct and cross-couplingcoefficients between Set-3 power and data coils in Table II, allon a semi-logarithmic scale. The main point to be made here isthat as a result of adopting the novel coil configuration of Fig. 5,the undesired data-power cross-coupling, , in Fig. 10(d) and11 are at least one order of magnitude smaller than the desireddirect data coils coupling, , in Figs. 10(b) and 11. This cansignificantly help in reducing power carrier interference withthe received data signal on the implant side, especially whencombined with frequency separation of data and power carriers,frequency selective nature of the LC tank circuits, and high-passfiltering of the received data carrier.

B. Power and Forward Data Transfer Measurements

In order to demonstrate simultaneous power and data trans-mission using the multicarrier wireless link, we modified andused the power and data transmission/reception circuitry thatwe had previously developed for an implantable wireless mi-crostimulating system (Interestim-2B) [11]. In this experimentwe used coil Set-3, shown in Fig. 8 and specified in Table II,at mm. Interestim-2B (IS-2B) was originally designedto operate with a single FSK carrier at 5 and 10 MHz. There-fore, we had to disable the on-chip power supply and amplifythe received data carrier using an off-chip comparator. IS-2Bwas used in this test basically as an FSK data demodulator andsince the demodulator was not specifically designed for a mul-ticarrier setting, the results of this experiment are not represen-tative of the highest bandwidth that can be achieved using mul-tiple carrier wireless links. They can however, demonstrate thefeasibility and advantages of utilizing multiple carriers in highperformance implantable device applications.


Fig. 12. (a) Power and forward data transmission measurement setup(b) sample measured waveforms showing from bottom to top: the receivedpower carrier at 500 kHz, rectified and regulated �5-V dc from the powercarrier, received data carrier at 5/10 MHz, and recovered data.

On the transmitter side, a high efficiency class-E power am-plifier, tuned at kHz, drove from a dc supply

[37]–[39]. In order to reach V, as shownon the third trace from top in Fig. 12(b), had to be increasedto 10 V. In this condition 81 mA was being consumed by theclass-E, resulting in a total input power of 810 mW. Meanwhileon the receiver side, was tuned at with a parallel capac-itor and connected to a fast full-wave rectifier. The rectifierwas followed by a low dropout regulator, which converted thereceived power carrier, with peak to peak amplitude of 8V as shown on the lower trace in Fig. 12(b), to a 5-V regulateddc voltage . The regulator was loaded withand as well as IS-2B chip and the comparator asshown Fig. 12(a). The power delivered to the comparator andIS-2B in this condition was measured 125 mW, while another117 mW was being dissipated in . The result was an overallpower transmission efficiency of from the externaldc supply to the regulated power supply on the implant.

Data was transmitted using pc-FSK technique described in[15]. Logic values “1” and “0” were transmitted using

MHz and MHz, respectively. To provide a wide-band low-Q bandpass spectrum across the inductive data linkwithout using dissipative components, a combination of seriesand parallel LC tank circuits were used to generate zeros at

and across the data transmitter output. In this ex-periment, we used an Agilent 33250A function generator with50 output impedance and maximum modulation rate of 100kHz to generate the FSK data carrier. Since we had already de-signed based on coils geometrical considerations, describedin Section III, other and values of the FSK transmitter inFig. 12(a) were chosen based on the equations derived in [15].Either or could be used as in Fig. 1.

FSK data carrier with a peak to peak voltage of 10 V at thetransmitter output was applied to the series-parallel LC tankcombination while the parallel tank circuit was chosento transmit the data, as shown in Fig. 12(a). The received datacarrier should theoretically have the same amplitude at both

and frequencies if the tank circuits are perfectlytuned. However, due to nonidealities it was being slightly am-plitude modulated as can be seen on the 2nd trace from top inFig. 12(b). The amount of power carrier interference on the re-ceived data carrier without any high-pass filtering was about25 mV, which can be tolerated by the FSK demodulator. It canalso be easily removed using a high-pass filter with cutoff fre-quency around 2 MHz. The major cause of this interference isthe asymmetry resulting from coil construction and vertical mis-alignments, which can be reduced by more accurate coil fabri-cation or better alignment methods. The top trace in Fig. 12(b)shows the recovered serial data bit stream at the output of theIS-2B FSK demodulator at 200 kb/s.

C. Back Telemetry Link

The feasibility, design, and construction of a microwave rangehigh frequency back telemetry link have already been demon-strated by Gosalia et al. in [23] and [35]. The size of the im-plantable planar patch antenna, in Fig. 1 ( 6 mm), is wellwithin the size of the implant and coils discussed in earlier sec-tions and it can be added at the center or on the side of the powercoils, shown in Fig. 5. If the implantable device is packaged ina nonconductive material such as ceramic, can be either em-bedded in the package or designed within one metal layer of theimplant printed circuit board without resulting in a significantincrease in the implant size.

VI. CONCLUSION

A new approach for efficient power and wide-band data trans-mission through a wireless link to high performance IMDs, suchas cochlear implants and visual prostheses, has been presentedusing multiple carrier signals. Two pairs of coils and a pair ofpatch antennas utilize a low-frequency MHz high-amplitude V carrier for power transmission, amedium-frequency MHz medium-ampli-tude V carrier for forward data transmissionfrom the external sensor to the implant, and a high-frequency

MHz low-amplitude V carrier for backtelemetry from the implant to the external part of the system.Separation of these functions helps the designer to optimize var-ious aspects of the link for a specific application regardless ofthe other constraints.

Coupling coefficients between coils with various geometriesare modeled and design guidelines are deducted regarding how


to maximize or minimize them. Adoption of novel coil geome-tries and orientations have resulted in increasing the desired di-rect coupling within power or data coil pairs, while decreasingundesired cross-coupling between the pairs to reduce interfer-ence between power and data carriers. Simulation and modelingresults using coil analysis software as well as measurement re-sults using small handmade coils indicate the feasibility andfunctionality of this approach for simultaneous data and powertransmission using multiple carrier signals.

ACKNOWLEDGMENT

The authors would like to thank the members of the NCBionics Lab, Dr. G. Lazzi, and his group for their help inconducting some of the measurements.

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Maysam Ghovanloo (S’00–M’04) was born in1973. He received the B.S. degree in electrical engi-neering from the University of Tehran, Tehran, Iran,in 1994, and the M.S. (Hons.) degree in biomedicalengineering from the Amirkabir University ofTechnology, Tehran, Iran, in 1997; and the M.S.and Ph.D. degrees in electrical engineering from theUniversity of Michigan, Ann Arbor, in 2003 and2004, respectively.

His Ph.D. research related to developing a wirelessmicrosystem for micromachined neural stimulating

microprobes. In December 1998, he founded Sabz-Negar Rayaneh Co. Ltd.,Tehran, Iran, to manufacture physiology and pharmacology research laboratoryinstruments. In the summer of 2002, he was with the Advanced Bionics Cor-poration, Sylmar, CA, designing spinal-cord stimulators. From 2004 to 2007,he was an Assistant Professor in the Department of Electrical and ComputerEngineering,North Carolina State University, Raleigh, where he founded anddirected the NC Bionics Laboratory. In June 2007, he joined the faculty of theGeorgia Institute of Technology, Atlanta, GA, where he is an Assistant Professorin the Department of Electrical and Computer Engineering. He has more than40 conference and journal publications.

Dr. Ghovanloo has received awards in the operational category of the 40thand 41st DAC/ISSCC student design contest in 2003 and 2004, respectively. Hehas served as a technical reviewer for major IEEE and IoP journals in the areasof solid-state circuits, systems, and biomedical engineering. He is a member ofTau Beta Pi, Sigma Xi, and IEEE Solid-State Circuits, Circuits and Systems,and Engineering in Medicine and Biology societies.

Suresh Atluri (S’05–M’07) was born in India, in1983. He received the B.S. degree in electrical andelectronics engineering from BITS, Pilani, India, in2004 and the M.S. degree in electrical engineeringfrom the North Carolina State University, Raleigh, in2006 specializing in analog, digital, and microwavecircuit design.

His research was oriented towards developinginductive power and data transfer circuits forimplantable biomedical devices. He joined the Inte-grated Device Technologies Incorporated in Atlanta,

GA, in 2006, where he works as an Electronics Design Engineer.

Documents

A Wide-Band Power-Efficient Inductive Wireless Link for Implantable