Deliverable D5 - Terranova · D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation

D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests

TERRANOVA Project Page 1 of 92

This project has received funding from Horizon 2020, European Union’s

Framework Programme for Research and Innovation, under grant

agreement No. 761794

Deliverable D5.1 Report on preliminary THz RF-Frontend and

Antenna, Phased array beamforming,

baseband algorithms and optical RF-frontend

ready for implementation in off-line tests Work Package 5 – THz System Technology

TERRANOVA Project

Grant Agreement No. 761794

Call: H2020-ICT-2016-2

Topic: ICT-09-2017 - Networking research beyond 5G

Start date of the project: 1 July 2017

Duration of the project: 30 months



Disclaimer This document contains material, which is the copyright of certain TERRANOVA contractors,

and may not be reproduced or copied without permission. All TERRANOVA consortium

partners have agreed to the full publication of this document. The commercial use of any

information contained in this document may require a license from the proprietor of that

information. The reproduction of this document or of parts of it requires an agreement with

the proprietor of that information. The document must be referenced if used in a

publication.

The TERRANOVA consortium consists of the following partners.

No. Name Short Name Country 1

(Coordinator)

University of Piraeus Research Center UPRC Greece

2 Fraunhofer Gesellschaft (FhG-HHI & FhG-IAF) FhG Germany 3 Intracom Telecom ICOM Greece 4 University of Oulu UOULU Finland 5 JCP-Connect JCP-C France 6 Altice Labs ALB Portugal 7 PICAdvanced PIC Portugal



Document Information

Project short name and number TERRANOVA (761794) Work package WP5 Number D5.1 Title Report on preliminary THz RF-Frontend and

Antenna, Phased array beamforming,

baseband algorithms and optical RF-frontend

ready for implementation in off-line tests

Version V1.0 Responsible unit FhG Involved units FhG, PIC, ICOM, UOULU, UPRC, ALB Type1 R Dissemination level2 PU Contractual date of delivery 30.06.2018 Last update 30.06.2018

1 Types. R: Document, report (excluding the periodic and final reports); DEM: Demonstrator, pilot,

prototype, plan designs; DEC: Websites, patents filing, press & media actions, videos, etc.; OTHER: Software, technical diagram, etc. 2 Dissemination levels. PU: Public, fully open, e.g. web; CO: Confidential, restricted under conditions

set out in Model Grant Agreement; CI: Classified, information as referred to in Commission Decision 2001/844/EC.



Document History

Version Date Status Authors, Reviewers Description

v0.1 20.03.2018 Draft Thomas Merkle (FhG-IAF) Initial definition of a

document structure

v0.2 27.03.2018 Draft Thomas Merkle (FhG-IAF) First version of Section 1

and Executive Summary

v0.3 04.03.2018 Draft Thomas Merkle (FhG-IAF) Revision of structure

based on input of all

partners

v0.4 13.06.2018 Draft Robert Elschner (FhG-HHI)

Georgia Ntouni (ICOM)

Francisco Rodrigues (PIC)

Draft of Section 3 (PIC),

Section 5 (HHI), and

Section 6 (ICOM)

v0.5 19.06.2018 Draft Dimitrios Kritharidis (ICOM)

Alexandros Katsiotis (ICOM)

Francisco Rodrigues (PIC)

A.-A. A. Boulogeorgos (UPRC)

Janne Lehtomäki (UOULU

Input to Section 2

(ICOM), Section 3 (PIC),

Section 4 (UPRC) and

Section 6 (UOULU)

v0.6 22.06.2018 Draft Georgia Ntouni (ICOM) Input to Section 6

v0.7 26.06.2018 Draft Thomas Merkle (FhG-IAF)

Robert Elschner (FhG-HHI)

Input to Section 2, 3, 4

and 5

v0.8 28.06.2018 Draft A.-A. A. Boulogeorgos (UPRC)


Janne Lehtomäki (UOULU)

Review / proofreading of

Section 1, 6

v0.9 29.06.2018 Draft Dimitrios Kritharidis (ICOM)


Robert Elschner (FhG-HHI)

Thomas Merkle (FhG-IAF)

Review / proofreading of

all sections, final editing

v1.0 30.06.2018 Final Thomas Merkle (FhG-IAF)

Angeliki Alexiou (UPRC)

Revision of all sections



Acronyms and Abbreviations

Acronym/Abbreviation Description 5G Fifth Generation

A ACO Analog Coherent Optics ADC Analog-to-Digital Converter AFC Automatic Frequency Correction AFE Analogue FrontEnd AGC Automatic Gain Control AiP Antenna-in-Package AM Amplitude Modulation AMC Adaptive Modulation and Coding AP Access Point ASIC Application-Specific Integrated Circuit ATDE Adaptive Time Domain Equalizer AWG Arbitrary Waveform Generator AWGN Additive White Gaussian Noise AWV Antenna Weight Vector

B BB BaseBand BC Beam Combining BEOL Back End of Line BER Bit Error Rate BF BeamForming BPSK Binary Phase Shift Keying BS Base Station

C CAUI 100 gigabit Attachment Unit Interface CDR Clock and Data Recovery CFP C-Form Factor Pluggable CMOS Complementary Metal–Oxide–Semiconductor CoMP Coordination Multi-Point COTS Commercial Off-The-Shelf / Components Off-The-Shelf CPR Carrier Phase Recovery CRC Cyclic Redundancy Code CSMA/CA Carrier Sense Multiple Access with Collision Avoidance CW Continuous Wave DAC Digital to Analog Converter

D DC Direct Current DCH Data CHannel DDC Digital Down Conversion DEMUX DE-MUltipleXer DL DownLink DMT Discrete Multi-Tone DoA Direction of Arrival DoF Degree of Freedom DP-IQ Dual Polarization In-phase and Quadrature



DPD Digital PreDistortion DSB Double Sideband DSP Digital Signal Processing DUC Digital Up Conversion DWDM Dense Wavelength Division Multiplexing

E E2E End-to-End EC European Commission E/O Electrical-Optical ETSI European Telecommunications Standards Institute eWLB embedded Wafer Level Ball grid array

F FEC Forward Error Correction FD Full Duplex FDD Frequency Division Duplexing FDMA Frequency Division Multiple Access FIFO First In First Out FM Frequency Modulation FPGA Field-Programmable Gate Array FEOL Front End of Line FSO Free-Space Optics FSPL Free Space Path Loss FTTH Fiber To The Home FWA Fixed Wireless Access

G GaAs Gallium Arsenide

H HEMT High Electron Mobility Transistor

I I/Q In-phase and Quadrature I2C Inter-Integrated Circuit IEEE Institute of Electrical and Electronics Engineers IF Intermediate Frequency IM/DD Intensity Modulation/Direct Detection ISI InterSymbol Interference ISM Industrial Scientific and Medical band ITU International Telecommunication Union ITU-R Radiocommunication sector of the International

Telecommunication Union IQD Indoor Quasi Directional

K KPI Key Performance Indicator

L LO Local Oscillator LoS Line of Sight LNA Low Noise Amplifier

M MAC Medium Access Control mHEMT Metamorphic High Electron Mobility Transistor MIMO Multiple Input Multiple Output MMIC Monolithic Microwave Integrated Circuit mmWave Millimeter Wave MUE Mobile User Equipment



MUX MUltipleXer MZI Mach-Zehnder Interferometer

N NGPON2 Next-Generation Passive Optical Network 2 nLoS Non-Line Of Sight NR New Radio NRZ Non-Return to Zero

O OFDM Orthogonal Frequency Division Modulation OIF Optical Internetworking Forum OLT Optical Line Terminal ONUs Optical Network Units OOK On-Off Keying

P P2MP Point-to-Multi-Point P2P Point-to-Point PA Power Amplifier PAM Pulse Amplitude Modulation PCB Printed Circuit Board PDM Polarization-Division Multiplexing PDM-QAM Polarization Multiplexed Quadrature Amplitude

Modulation PER Packet Error Rate PHY PHYsical PIC Photonic Integrated Circuit PLL Phased Locked Loop PONs Passive Optical Networks PSP Pulse Shaping Filter PtMP Point-to-Multi-Point

Q QAM Quadrature Amplitude Modulation QoE Quality of Experience QoS Quality-of-Service QSFP Quad Small Form-Factor Pluggable

R RA Random Access RAT Radio Access Technology RAU Remote Antenna Unit RF Radio Frequency RoF Radio over Fiber RRM Radio Resource Management RSRP Reference Signal Received Power RSSI Received Signal Strength Indicator Rx Receiver

S SD-FEC Soft-Decision Forward-Error Correction SDM Space Division Multiplexing SDMA Space Division Multiple Access SDN Software Define Network SFF Small Form Factor SFP Small Form-Factor Pluggable SiGe Silicon-Germanium SISO Single Input Single Output



SLS Sector Level Sweep SM Spatial Multiplexing SMF Single Mode Fiber SNR Signal to Noise Ratio SOTA State Of The Art SPI Serial Parallel Interface SRC Sample Rate Conversion SSB Single-SideBand SSW Sector SWeep SSW-FBCK Sector SWeep FeedBaCK STM-1 Synchronous Transport Module, level 1 STS Symbol Timing Synchronization

T TDD Time Division Duplexing TDM Time Division Multiplexing TDMA Time Division Multiple Access TERRANOVA Terabit/s Wireless Connectivity by Terahertz innovative

technologies to deliver Optical Network Quality of

Experience in Systems beyond 5G

THz Terahertz TIA TransImpedance Amplifier TWDM Time and Wavelength Division Multiplexed Tx Transmitter

U UL Uplink UE User Equipment

V VCO Voltage Controlled Oscillator VGA Variable Gain Amplifier VLC Visible Light Communication

W WLAN Wireless Local Area Network WDM Wavelength Division Multiplexing WiFi Wireless Fidelity WLBGA Wafer Level Ball Grid Array

X XG-PON 10 Gbit/s Passive Optical Network XPIC Cross Polarization Interference Cancellation



Contents

1. Introduction ........................................................................................................................... 16

1.1 Scope ....................................................................................................................... 17

1.2 Structure .................................................................................................................. 18

1.3 References ............................................................................................................... 18

2. System Block Diagram REVIEW ............................................................................................. 19

2.1 Review of TERRANOVA Architecture Candidates .................................................... 19

2.2 Generic Architecture and Hardware Overview ....................................................... 19

2.3 Beamforming Demonstrator Block Diagram ........................................................... 20

2.4 References ............................................................................................................... 21

3. Optical Link and TERRANOVA Media Converter Design ........................................................ 22

3.1 Standardized Optical Transceivers .......................................................................... 22

3.1.1 IM/DD Transceivers ......................................................................................... 22

3.1.2 Coherent Transceivers ..................................................................................... 23

3.2 TERRANOVA Media Converter Design ..................................................................... 24

3.2.1 Proposal 1 – IM/DD using PICadvanced NG-PON2 technology ....................... 24

3.2.2 Proposal 2 – IM/DD using COTS 100G ............................................................. 26

3.2.3 Proposal 3 - IM/DD transceivers based on amplitude modulation ................. 26

3.2.4 Proposal 4 – Optical Coherent transmission ................................................... 28

3.2.5 Conclusion ....................................................................................................... 30

3.3 Concepts for Future Media Converter Integration ................................................. 30

3.4 Conclusions and Outlook ......................................................................................... 35

3.5 References ............................................................................................................... 35

4. RF Frontend and Antenna Prototypes ................................................................................... 36

4.1 THz Frontends for Point-to-Point Applications ....................................................... 36

4.1.1 Duplexing Techniques...................................................................................... 37

4.1.2 Transceiver Correction Schemes and Synchronization ................................... 38

4.1.3 Transceiver Architectures and Project Development Plan .............................. 40

4.2 New Process Technologies for III-V based MMICs .................................................. 41

4.2.1 Technology Overview ...................................................................................... 41

4.2.2 Motivation and Status of BEOL Development ................................................. 43

4.3 New MMIC Frontend Building Blocks ...................................................................... 46

4.3.1 Component Candidates for the TERRANOVA Media Converter...................... 47

4.3.2 Circuit Components for 220-320 GHz Transceivers ......................................... 49




4.5 References ............................................................................................................... 53

5. Baseband Digital Signal Processing for THz Systems ............................................................. 58

5.1 THz P2P transmission experiments ......................................................................... 58

5.1.1 Experimental setup .......................................................................................... 59

5.1.2 Used Digital Signal Processing ......................................................................... 60

5.2 First THz System Measurement Results .................................................................. 61


5.4 References ............................................................................................................... 67

6. Phased Array Beamforming ................................................................................................... 68

6.1 State-of-the-Art in Phased Array Beamforming techniques ................................... 68

6.2 Mathematical Model of Phased Array Architectures .............................................. 69

6.3 Comparison of Beamforming Techniques ............................................................... 70

6.3.1 Conventional Beamforming ............................................................................. 70

6.3.2 Tapered Beamforming ..................................................................................... 71

6.3.3 Null-Steering Beamforming ............................................................................. 74

6.3.4 Adaptive beamforming .................................................................................... 74

6.3.4.1 Minimum Variance Distortionless Response (MVDR) Beamformer ............ 76

6.3.4.2 Linearly Constrained Minimum Bariance (LCMV) Beamformer .................. 76

6.4 Beamforming Implementation Issues of Demonstrators ........................................ 80

6.4.1 Available Phase Shifter Resolution for Analogue Beamforming ..................... 80

6.4.1 Differential Phase Noise in THz Phased Array Systems ................................... 82

6.4.2 Four Element Horn Antenna Array .................................................................. 85

6.4.3 Beam Search and Alignment ........................................................................... 87

6.5 Calibration Techniques for Phased Array Antennas ................................................ 88


6.7 References ............................................................................................................... 90

7. Conclusions ............................................................................................................................ 92



List of Figures

Figure 1: WP5 project plan and connection with WP2 and WP6. ........................................... 17

Figure 2: General overview of the components under investigation within WP5. ................. 19

Figure 3: Generic view on the TERRANOVA media converter. ................................................ 20

Figure 4: LO generation option for the digital beamformer. .................................................. 21

Figure 5: Typical IM/DD optical transceiver pluggable in XFP form factor. ............................ 22

Figure 6: Basic block diagram of an XFP IM/DD transceiver module. ..................................... 23

Figure 7: Coherent receiver architecture. ............................................................................... 24

Figure 8: Possible application of the IM/DD transceivers using PICadvanced

NG-PON2 technology. ............................................................................................................. 24

Figure 9: IM/DD solution based on PICadvanced NG-PON2 transceivers. .............................. 25

Figure 10: IM/DD solution based on CFP transceivers. ........................................................... 26

Figure 11: IM/DD solution based on amplitude linear transceivers. ...................................... 27

Figure 12: Possible application of the coherent solution. ....................................................... 28

Figure 13: Coherent solution based on single polarization THz radio interface. .................... 29

Figure 14: Coherent solution based on the dual-polarization THz radio interface with

optional DSP at the radio front end. ....................................................................................... 30

Figure 15: Functional block diagram of the media converter for coherent optical

transmission. ........................................................................................................................... 31

Figure 16: Illustration of a typical ACO module and the cage system on the host board [3-9] .

................................................................................................................................................. 31

Figure 17: Insertion loss of the board-to-board connector on the host board, taken from [3-

8]. ............................................................................................................................................. 32

Figure 18: Manufactured baseband amplifier test module for risk and problem

identification. .......................................................................................................................... 32

Figure 19: Measured transmission line loss using different printed circuit board materials. 33

Figure 20: Calculated frequency limitations of the centred striplines as a function of the

waveguide thickness. .............................................................................................................. 34

Figure 21: State-of-the-art of hybrid integrated optical transponder modules. .................... 34

Figure 22: Channel allocation plan of IEEE 802.15.3d-2017 standard. ................................... 37

Figure 23: Overview of design activities of M1-M12. ............................................................. 41

Figure 24: Comparison of different transistor technology options for THz frontend design. 42

Figure 25: Comparison of various BEOL processes, for different transistor technologies ...... 43

Figure 26: BEOL development and tests in TERRANOVA, (1) base process, (2) development

run 1, (3), final solution for final transceiver integration. ....................................................... 44

Figure 27: Left: SEM picture of manufactured mixer IP core component for 220 to 300 GHz

with the 3LPP process (before TFMS end passivation), right: corresponding optical

photograph. ............................................................................................................................. 45

Figure 28: Left: Layout and photograph of manufactured transmission lines for process

testing and monitoring, right: measured attenuation of a microstrip line of 10 µm width. .. 45

Figure 29: Integration density of different BEOL for broadband RF applications (to scale), see

also [4-65] for the 65nm CMOS amplifier. .............................................................................. 46

Figure 30: Chip photograph of the fabricated Kukielka amplifier, schematic representation,

and summary of the measured key performance parameters. .............................................. 47



Figure 31: On-wafer measured performance of the manufactured Kukielka amplifier, (a) S-

parameters and noise figure, (b) output power compression characteristics at 10 GHz. ...... 48

Figure 32: Designed test package and photograph of the manufactured 4-channel Kukielka

amplifier chip. .......................................................................................................................... 49

Figure 33: Measured frequency response, left: magnitude in dB, right phase in deg for the 4-

channel Kukielka amplifier. ..................................................................................................... 49

Figure 34: First IP core library for the DL, MMIC for generating an LO signal that can be tuned

varied from 200 to 260 GHz. ................................................................................................... 50

Figure 35: On-wafer measured output power at 240 GHz for the LO chip ............................. 50

Figure 36: Chip photograph of manufacture 120 GHz driver amplifier and measured RF

performance (S-parameters and output power characteristics). ........................................... 51

Figure 37: Chip photograph of the fabricated broadband LO multiplier for selecting different

Rx/Tx channels from 220 to 290 GHz and measured output power characteristic at a fixed

input power level. .................................................................................................................... 51

Figure 38: Chip photograph of the fabricated 220 -260 GHz power amplifier for the DL

frequency band, left 3LPP design, right 4L design with MET4 layers (before MET4 processing)

. ................................................................................................................................................ 52

Figure 39: On-wafer measured RF performance of the 220 -260 GHz power amplifier, left S-

parameters, right output power at 240 GHz. .......................................................................... 52

Figure 40: Experimental setup for tests of THz P2P link ......................................................... 58

Figure 41: Photograph of first lab setup at Fraunhofer HHI.................................................... 58

Figure 42: Block diagram of the digital signal processing ....................................................... 61

Figure 43: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd using 16QAM

(without pre-emphasis, at maximum Tx output power, optimal Ry I/Q orientation, at a

carrier frequency of 306.36 GHz) ............................................................................................ 62

Figure 44: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd 16QAM (with pre-

emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier

frequency) ............................................................................................................................... 62

Figure 45: Received constellations under best case conditions. (a) 16 GBd 16QAM without

pre-emphasis @ BER = 1·10-3. (b) 16 GBd 16QAM with pre-emphasis @ BER = 6.5·10-4. ...... 63

Figure 46: BER vs. Carrier Frequency for 16 GBd 16QAM (no pre-emphasis, maximum Tx

output power, optimal Rx I/Q orientation, optimal digital DAC amplitude) ........................... 63

Figure 47: BER vs. Rx I/Q orientation for 16-GBd 16QAM (different carrier frequencies, no

pre-emphasis, maximum Tx output power, optimal digital DAC amplitudes) ........................ 64

Figure 48: Discussion of Rx I/Q orientation. (a) Schematic of Rx aligned QAM constellation

(orientation angle = 0°). (b) Schematic of Rx misaligned QAM constellation (orientation angle

= 45°). (c) Measured Rx constellation (only I component at Tx) at an orientation angle giving

the best case BER. (d) Measured Rx constellation (only I component at Tx) at an orientation

angle giving the worst case BER. ............................................................................................. 64

Figure 49: BER vs. THz attenuation for 16-GBd 16QAM (306.36 GHz carrier frequency, with

pre-emphasis, optimal digital DAC amplitudes, optimal Rx I/Q orientation) ......................... 65

Figure 50: BER vs. RX frequency offset for 16-GBd 16QAM (306.36 GHz carrier frequency,

with pre-emphasis, maximum TX output power, optimal digital DAC amplitudes) ............... 66

Figure 51: BER vs. Digital DAC Amplitude (I and Q component) for 32-GBd 16QAM (with pre-

emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier

frequency) ............................................................................................................................... 66

Figure 52: Received constellation under best case conditions for 32-GBd 16QAM with pre-

emphasis @ BER = 1.1·10-2. ..................................................................................................... 67



Figure 53: Plane wave impinging on a ULA ............................................................................. 70

Figure 54: Beampatterns for conventional beamforming ....................................................... 71

Figure 55: Beampatterns for tapered beamforming with binomial distribution and 4 AEs ... 72

Figure 56: Beampatterns for tapered beamforming with various windows. The ULA consists

of 16 elements and its steering direction is at in (d) is the number of nearly

constant-level sidelobes adjacent to the mainlobe ................................................................ 73

Figure 57: Beampatterns for Null-steering beamforming with 16 AEs ................................... 75

Figure 58: Beampatterns for Null-steering beamforming with 4 AEs and ............ 76

Figure 59: Magnitude of the beamformers’ output signals for a ULA with 4 AEs................... 78

Figure 60: Power pattern of beamformers for a ULA with 4 AEs ............................................ 78

Figure 61: Power pattern (rectangular) of beamformers for a ULA with 4 AEs ...................... 79

Figure 62: Magnitude of the beamformers’ output signals for a ULA with 16 AEs ................ 79

Figure 63: Power pattern of beamformers for a ULA with 16 AEs .......................................... 80

Figure 64: Beampatterns for conventional beamforming with and without quantization

assuming 4 AEs ........................................................................................................................ 81

Figure 65: Beampatterns for conventional beamforming with and without quantization

assuming 16 AEs ...................................................................................................................... 82

Figure 66: Functional block diagram of the 4 channel LO beamformer with DDS based phase

shifting. The synchronous DAC output is achieved by a master trigger (omitted in this

drawing for clarity), [6-14]. ..................................................................................................... 83

Figure 67: Phase noise of the individual channels of the LO beamformer, measured at the

output carrier frequency of 8.333 GHz. In comparison, the phase noise of a commercial

frequency synthesizer is shown. Carrier power Pc = -2 dBm in all cases, [6-14]. ................... 83

Figure 68: Measured cumulative constellation diagrams and EVM of a single receive channel

at 4 Gbaud for increasing QAM modulation depths. 100 constellation diagrams of 4096

symbols were accumulated in each plot, [6-14]. .................................................................... 84

Figure 69: Measured cumulative constellation diagrams and EVM of a single receive channel

at 4 Gbaud for a 16-QAM modulation format, after initial calibration by null-steering. 100

constellation diagrams of 4096 symbols were accumulated in each plot, [6-14]. .................. 84

Figure 70: Horn antenna element ........................................................................................... 85

Figure 71: Directivity pattern of the horn antenna element ................................................... 85

Figure 72: Directivity pattern of the four elements of the ULA .............................................. 86

Figure 73: Directivity radiation pattern of a ULA with 4 AEs pointing at ................. 86

Figure 74: Directivity radiation patterns of a ULA with 4 AEs pointing at ............. 87



List of Tables

Table 1: Main experimental parameters ................................................................................. 59

Table 2: Relation of clock and carrier frequencies used in the experiment ............................ 60

Table 3: Parameters for digital signal processing blocks ......................................................... 61

Table 4: SINR for various beamformers .................................................................................. 77

Table 5: Output noise power for various beamformers.......................................................... 77

Table 6: Interference power for various beamformers ........................................................... 77

Table 7: Horn antenna element dimensions ........................................................................... 85



Executive Summary

The deliverable “D5.1 Report on preliminary THz RF-Frontend and Antenna, Phased array

beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-

line tests” is the first of a series of three deliverables that accompany the research and

development work on the “THz System Technologies” of TERRANOVA. There are two overall

objectives in work package WP5. Firstly, research towards new software and hardware

components that address the peculiarities and challenges of hybrid optical-THz

communication systems. Secondly, practically implementing and providing components for

the integration in the two system demonstrators of WP6 (“THz Demonstrator

Implementation and Validation”). With the first demonstrator, beamforming algorithms and

concepts will be tested and validated. With the second demonstrator, the capacity bounds

of a hybrid optical-THz communication link will be experimentally explored. Different use-

case scenarios and suitable system architecture candidates for each scenario were identified

as part of deliverable D2.1 (“TERRANOVA System Requirements” and D2.2 (“TERRANOVA

System Architecture”). The system component design and implementation may reveal

challenges that were not anticipated during the early system concept phase of the project.

There are also open questions on what performance the THz system components can

achieve today and in future.

The design of preliminary THz system components means in the context of D5.1 that designs

“ready for implementation in off-line tests” are presented or in other words the technology

concept of the components is formulated. At this stage, the major hardware components of

interest are the hybrid optical-electrical interface, the THz Rx/Tx frontend at chip level and

the antenna. They define all together the optical-THz frontend for which also an integration

platform and future packaging strategies will be outlined in the course of the project. The

major software components are the baseband modem and beamforming algorithms. The

formulation of the hardware concepts is documented by a first set of prototype designs

“ready for implementation” which means ready for fabrication. Offline tests mean that the

algorithms can be tested in a simulation environment with artificial data or experimentally

recorded datasets. The successive deliverables D5.2 and D5.3 will provide proof of the

component concepts in a lab environment. This means that the functioning of the hardware

components will be experimentally tested and software components will be ready for the

implementation in a real-time environment. In WP6, the THz system components of WP5

will be integrated for demonstration of the TERRANOVA system concepts and the validation

of the technologies.

The main objectives of the deliverable are to progress in:

Development of a high level hybrid optical-THz wireless modem architecture, and

identification of the required key algorithms for point-to-point transmission;

Identification of concepts for Rx/Tx frontends and the design of integrated circuit

prototypes of the top candidates;

Methods for beamforming and calibration of THz Rx/Tx antenna arrays for future

use in line-of-sight beamforming schemes with pre-defined codebooks, and

Top-down design of the TERRANOVA media converter for the hybrid optical-THz link,

and identification of electro-optical THz frontend integration and packaging

solutions.



1. INTRODUCTION

One of the key enabling technologies of TERRANOVA is embedding terahertz (THz) wireless

links into fibre optic communication networks. Several use case scenarios and system

architectures were previously identified in the deliverables of D2.1 and D2.2 of this project

[1-1], [1-2]. They all promise to extend the quality of service and experience (QoS and QoE)

from optical systems to the THz wireless domain, which also implies date rates in the Tbit/s

regime. The currently available wireless baseband modem technologies do not, however,

meet the requirements of such systems. Instead of wireless baseband modems, the

TERRNAOVA solutions use COTS (commercial-off-the-shelf) optical transceivers. Provided

that the THz wireless system components can be designed to comply with the optical

transceivers and optical links, the enormous technical progress of the fibre optical industry

can be exploited, e.g. powerful ASICs (application-specific integrated circuits) for DSP (digital

signal processing) and specific ADC/DACs having large analogue bandwidths, which

nowadays are able to support data rates of 100/200 and 400 Gbit/s. Work package (WP) 5 of

TERRANOVA investigates and develops the concepts of THz modems, frontends and

antennas, and the interface to embed them into different fibre optic links.

The TERRANOVA use case scenarios consider P2P (point-to-point) and P2MP (point-to-multi-

point) link architectures, which can be part of a static, reconfigurable or mobile network.

This leads to quite a number of design variations and alternatives for the required THz

system components. In order to handle the complexity within this project, the

implementation of two different system demonstrators was planned, which are a high

capacity P2P hybrid fibre optic - THz wireless link and a beamforming THz wireless link. Each

demonstrator focuses on specific challenges and innovations only. The first demonstrator

explores the capacity bounds of wideband channels at frequencies from 220-325 GHz. The

second demonstrator investigates phased array frontend architectures and their calibration

and control for beamforming.

WP5 develops component prototypes for implementing the two different system

demonstrators. This WP also addresses research challenges and innovations within the two

TERRANOVA pillars, i.e., “Tbit/s Wireless Connectivity” and “Co-Design of Optical and THZ

Wireless Links”, where some of the important interdisciplinary topics are:

Spectral efficient THz transceivers above 200 GHz: There is a significant functional

gap between the available THz wireless frontends of today and what is already

considered state-of-the-art (SOTA) at millimetre-wave (mmWave) frequencies.

TERRANOVA focuses on the integration of broadband analogue baseband functions,

the revision of transceiver architectures and analogue/digital correction schemes to

operate with higher order modulated signals at THz frequencies.

End-to-end optimized hybrid optical – THz wireless links: The hybrid fibre optic –

THz wireless link raises the question how the fibre optic baseband DSP (optical DSP)

has to be modified to cope with the THz wireless link. TERRANOVA investigates de-

emphasis, frontend correction and impairment mitigation for such hybrid links,

targeting Tbit/s connectivity.

Signal conversion between optical and electrical carriers: There exists quite a

number of research works into the implementation of RoF (radio-over-fibre)

solutions at THz frequencies. In contrast, developing an electrical baseband interface



for mapping signals between the optical domain and the THz wireless domain, which

is consistent with standard COTS optical transceivers, has been little explored.

Coherency and synchronization: There is a need to accurately estimate the carrier

phase to guarantee coherency to the local oscillator. Digital schemes are quite

common for optical systems but there is little experience how those algorithms

perform for optical – THz wireless links.

THz antenna arrays and beamforming: There are no experimental studies available

beyond 200 GHz that explore the calibration and stability of phased array

communication systems and its applicability to different mobile use case scenarios.

TERRANOVA develops calibration schemes for the extraction of pre-defined

codebooks and models the application to different beamforming architectures.

All of the listed research and innovation areas are interdependent. The co-design of the THz

system components, hardware and digital signal processing, is inherently necessary to solve

the individual challenges. An overview of the high-level project plan of WP5 is provided in

Figure 1. The embedding of WP5 in between WP2 and WP6, and the interaction with WP3

and WP4 is vital for reaching the objectives.

Figure 1: WP5 project plan and connection with WP2 and WP6.

1.1 Scope

The objective of this deliverable is to formulate the concepts of the key THz system

components in detail, which means “ready for implementation”. In a first step, the system

block diagrams are broken down to the component level and suitable components and

concepts are identified. The main part of this deliverable identifies and develops first

prototype hardware components, suitable for fabrication, and algorithms suitable for off-

line testing. This requires setting up the design environment and the design flow towards

the implementation. The co-design of software and hardware components is considered and

highlighted where possible. Finally, based on simulation models, the relevant key

performance indicators are assessed for the investigated THz system components. Work



packages WP2 and WP3 will use those simulation results to refine the overall system models

and system simulations.

1.2 Structure

The structure of this document is as follows:

Section 2 (System Block Diagram) reviews the current system block diagrams to

break down the TERRANOVA candidate architectures into units for physical

implementation.

Section 3 (Optical Link and TERRANVOA Media Converter Design) identifies

transponder solutions for the main TERRANOVA architectures and down-selects

candidates for the demonstrators in WP6, defines the interfaces and baseband

development, and identifies future integration and packaging solutions.

Section 4 (RF Frontend and Antenna Prototypes) investigates analogue frontend

chipsets using a new back-end-of-line process for the identified THz frontend

architectures, the interfaces to the digital baseband, all accompanied by prototype

designs relevant for the demonstrators in WP6.

Section 5 (Baseband Digital Signal Processing for THz Systems) derives from the

optical PHY (Physical Layer) a modified DSP architecture for the TERRANOVA optical-

THz modem, sets up the development and test environment, and investigates the

capacity limits of the existing solution experimentally

Section 6 (Phased Array Beamforming) identifies and develops digital beamforming

algorithms for THz phased array antennas, sets up a simulation environment for

proof of concept and models the currently existing solution as a first step.

Section 7 (Conclusions) summarizes the main achievements in D5.1, and outlines the

next steps towards the component implementation and testing.

Sections 3-6 reflect the four tasks of WP5, but in different order. Since the THz system

components need to comply with the optical link and its interface, this aspect is presented in

Section 3 first, which covers work in WP5.4 mostly. The architectures of the THz frontend

(WP5.1) and modem (WP5.3) have to be co-designed and their presentation order is

somewhat arbitrary for that reason (Sections 4 and 5). For example, there are aspects of

frontend impairment calibration and carrier phase synchronization, where the capabilities of

the PHY DSP influence the THz frontend architecture. Vice-versa, the frontend performance,

functionality and implementation possibilities have influence on the PHY DSP. The

presentation of the technical work ends in Section 6 with the calibration and control of

phased array frontends, investigated in WP5.2. The components for the beamforming

demonstrator depend also on the phased array frontend architectures and implementation

limitations, which are part of work in WP5.1.

1.3 References

[1-1] “D2.1: TERRANOVA System Requirements”, EU Project TERRANOVA (grant

agreement 761794), public deliverable, v1.0, December 2017, online: https://ict-

terranova.eu/

[1-2] “D2.2: TERRANOVA System Architecture”, EU Project TERRANOVA (grant agreement

761794), public deliverable, v1.0, March 2018, online: https://ict-terranova.eu/

https://ict-terranova.eu/





2. SYSTEM BLOCK DIAGRAM REVIEW

The TERRANVOA system architecture candidates for the identified P2P and P2MP use-case

scenarios, which were proposed in the deliverables D2.1 and D2.2 of this project, [2-1], [2-2],

lead to quite a number of different hardware and algorithm components to be investigated.

This section briefly refines the system block diagrams further and breaks them down into

sub-assemblies, which are composed of the physical hardware components to be

investigated.

2.1 Review of TERRANOVA Architecture Candidates

A conceptual block diagram of the TERRANOVA hybrid fibre-optical – THz wireless link was

presented in D2.1, which depicts a unidirectional link for simplicity. Figure 2 depicts the

required software and hardware components and their assignment to the different tasks of

WP5. Deliverable D2.2 distinguished between incoherent optical link architectures compliant

to Ethernet and/or PON standards, and coherent optical link architectures compliant to

CFP2-ACO standards, depending on the location of the link in the optical network. This lead

to different THz frontend architectures and baseband specifications. A bidirectional solution

increases the implementation complexity of the THz frontend and its interface to the optical

link further. Finally, the P2MP use cases lead to the need of THz frontend array

architectures. In order to handle the complexity within this project, the implementation of

two different system demonstrators was planned, which are (1) a high capacity P2P hybrid

fibre-optic - THz wireless link and (2) a beamforming THz wireless link.

Figure 2: General overview of the components under investigation within WP5.

2.2 Generic Architecture and Hardware Overview

There are several functions that have to be provided for both of the demonstrators and

which are common to the different architectures considered so far. The IQ direct conversion

frontend (see [2-2], page 76) is the most natural match to the coherent optical link but

requires the support of two polarizations, thus two IQ channels. The incoherent optical link,



which was proposed in [2-2] lacks spectral efficiency. Incoherent wireless PAM4 modulators

may be a way out, consistent also with optical link COTS, but the improvement of the net

data rate is only moderate. In addition, this solution requires gearbox functions at the

electro-optical interface. Instead, an alternative incoherent optical link solution is discussed

in Section 3.2, which is compatible to the coherent IQ direct conversion THz wireless

frontend architecture. This scheme uses for each of the wireless I and Q components a

separate optical link, which is a highly scalable solution, for example when using optical

wavelength-division multiplexing over a single fibre. Thus, it can also support the 2-channel

IQ direct conversion wireless architecture by using 4 optical links for the two IQ pairs. In

conclusion, the 2-channel transceiver frontend is a very versatile component, which allows

the investigation of different optical link solutions within TERRANOVA. The corresponding

electro-optical interface must support four analog baseband signals for the 2-channel IQ

direct conversion transceiver. The overall generic block diagram is depicted in Figure 3. This

representation is a unidirectional representation. The bi-directional counterpart includes the

same scheme for the wireless receive path. Both, the receive and transmit path need to be

adjusted to the optical interface by the analog baseband converter, which requires in its

most simplistic form only signal conditioning functions but may also include analog equalizer

functions for example. The preferred interface solution is a DSP-less solution. Within WP5,

the components for the TERRANOVA media converter and the THz frontend will be

investigated. For signal conditioning, broadband baseband amplifiers are key components to

focus on at first. In addition, a low-loss packaging platform needs to be established, possibly

also consistent with MSA pluggable standards.

Figure 3: Generic view on the TERRANOVA media converter.

2.3 Beamforming Demonstrator Block Diagram

The beamforming demonstrator focuses at first on digital beamforming. This requires also a

solution supporting multiple IQ-channels, which can be composed of the discussed 2-

channel transceiver units. For the digital beamformer, the LO generation must maintain a

good phase synchronization between the individual channels which is heavily impaired by

frequency multipliers that amplify the relative phase fluctuations. For this reason, the LO

generation cannot be carried out individually for each channel. A multichannel LO signal

source is a flexible solution as depicted in Figure 4. The LO interface is selected at an

intermediate frequency that allows board level routing between the LO signal source and

the IQ direct conversion transceivers. The fourth sub-harmonic seems to be a good

compromise for this frequency, though a detailed analysis by sample designs is planned in

the next phase of the project.



The beamforming demonstrator requires an analog baseband interface to a customized E-

band modem. Since all beamforming functions are implemented in the modem in this

solution, there are no control signals required between the modem and the frontend, except

for the correction of frontend imperfections.

Figure 4: LO generation option for the digital beamformer.

2.4 References

[2-1] “D2.1: TERRANOVA System Requirements”, EU Project TERRANOVA (grant

agreement 761794), public deliverable, v1.0, December 2017, online: https://ict-

terranova.eu/








3. OPTICAL LINK AND TERRANOVA MEDIA CONVERTER

DESIGN

This section covers the progress on planning and designing the components for the optical-

THz wireless link. Different concepts were considered in the first project phase and four

different proposals will be presented in the first part of this section as a result. The second

part reports on first steps towards the actual physical implementation on component level.

3.1 Standardized Optical Transceivers

3.1.1 IM/DD Transceivers

Generally, two main detection schemes can be used to convert an optical signal to the

electrical domain: direct detection and coherent detection. The first one is simply based on

intensity modulation and direct detection (IM/DD), i.e. only the total power of the optical

field is converted into an electrical signal by a simple photodiode at the receiver side. In the

second approach, the received signal is mixed and boosted with an optical source signal

provided by a local oscillator (LO), and the full optical field information of the signal (i.e.

amplitude, phase and polarization) can be converted into the electrical domain [3-1].

Most applications in data-centres, metro and access networks employ IM/DD pluggable

transceivers as they are more cost-effective as compared to coherent solutions. They

perform sufficiently well for the relative short distances that needs to be bridged.

Most common form-factors of the IM/DD transceivers are the well-known SFP+, QSFP28,

XFP and CFP2, with bit rates varying from 1 Gbps to 100 Gbps, and distances from hundreds

of meters to 80 km and more.

XFP transceivers

XFP transceivers began in point-to-point applications, but soon spread to Passive Optical

Networks (PON) applications. Since they are a cost-effective solution, they can support the

recent optical network technologies such as XG-PON, XGS-PON and NG-PON2.

Figure 5: Typical IM/DD optical transceiver pluggable in XFP form factor.

PICadvanced is specialized in XFP form-factor development for the new access network

technology NG-PON2, which supports 10G/10G for up-/downstream and up to 64 clients for

each of the 4 wavelengths enabling 40 Gbps aggregated traffic in downstream.

XFP transceivers (10 Gigabit Small Form Factor Pluggable) follow the Multi-Source

Agreement (MSA). Despite of the different technologies such as 10G Ethernet, SONET/SDH,

ITU-T 10G PON, XFP transceivers all have the same basic recommendations such as electrical

interface, management interface and power dissipation limits.



Interface

As it is common to many other transceiver form factors, XFP transceivers use an I2C

management interface, in order to control and read internal diagnostics.

XFP transceivers use two lanes of 10 Gbps, i.e. transmitter (Tx) and receiver (Rx) data paths –

CML or LVPECL -, to provide 10 Gbps symmetrical service. When possible, they employ the

use of CDR and CTLE to improve even further the link budget and reach longer distances. A

basic block diagram of an XFP can be found in Figure 6.

Figure 6: Basic block diagram of an XFP IM/DD transceiver module.

3.1.2 Coherent Transceivers

Backbone applications need higher capacity, higher spectral efficiency and longer fibre reach

which the IM/DD technology cannot deliver. Coherent transceivers have superior optical

performance and can provide electronic equalization of fibre impairments, such as

chromatic dispersion or polarization-mode dispersion. Moreover, as the phase from the

signal is recovered, higher order QAM modulation schemes are allowed leading to the

support of higher bitrates with high spectral efficiency. However, coherent solutions are

more expensive when compared to common IM/DD transceivers as they comprise more

complex optical components and require DSP ASICs with high-speed DACs and ADCs. Current

transceivers in the market can already achieve net bitrates of up to 400 Gbps.

Pluggable transceivers for coherent applications can be divided in analog coherent optics

(ACOs) and digital coherent optics (DCOs) subcategories. Both contain all necessary optical

components for transmit and receive in a single package. ACO offer lower cost, and typically

smaller form factor, however DCOs already comprises the DSP ASIC with DAC and ADC inside

the package, while ACO needs DSP capabilities on a host board.

In the current/next generation, pluggable coherent optical transceivers use the CFP/CFP2

form factor and follow the Multi-source Agreement / OIF agreement for coherent optics and

all the inherent guidelines. CFP/CFP2 uses the MDIO management interface and can have

different types of data path interfaces, i.e. digital (DCO) or analog baseband (ACO). A typical

block diagram for a coherent receiver can be found in Figure 7.



Figure 7: Coherent receiver architecture.

This detector employs polarization-diversity, where two phase-diversity configurations by

using two 90° optical hybrid components are combined to detect the in-phase and

quadrature signals of each polarization of the light. The polarization beam splitter (PBS) is

used to split both signal polarizations, and after four balanced detectors, the in-phase and

quadrature signals from each polarization can be sampled using ADCs. Therefore, a great

advantage of these receivers is the ability to detect the full information of the optical field,

enabling the use of advanced modulation formats with all four dimensions in the optical

transmission. Finally, advanced DSP can be employed to manipulate the digitized

information towards the compensation of the optical link distortions.

3.2 TERRANOVA Media Converter Design

In this section, we present several possible solutions for the TERRANOVA Media Converter

design. The TERRANOVA Media Converter enables the possibility to transmit from 100 Gbps

up to 800 Gbps from TERRANOVA radio front-end through the fiber. In order to transmit

such high data rates, we have to consider state of the art optical IM/DD and optical coherent

transmission technologies.

3.2.1 Proposal 1 – IM/DD using PICadvanced NG-PON2 technology

The first solution presented comprises the use of a time-wavelength division multiplexing

(TWDM) transceiver (included in PICadvanced portfolio) that allows the transmission of

multiple 10 Gbps links with different wavelengths in a single fiber. It can be viewed as a

lower cost solution when compared to commercial IM/DD 100 Gbps solutions. A block

diagram of a possible application of this proposal is presented in the Figure 8.

Figure 8: Possible application of the IM/DD transceivers using PICadvanced NG-PON2 technology.

For this type of application, the Figure 9 depicts the proposed optical architecture with the

TERRANOVA Media Converter. Therefore, the idea behind the system is to multiplex a

defined number of 10 Gbps modulated optical carriers in order to achieve the desired

aggregated data rate over the optical link. For the XFP solution, the non-return-to-zero (NRZ)



modulation format must be used due to the integrated subcomponents, such as limiting

amplifiers and transimpedance amplifiers (TIAs), or the clock and data recovery (CDR)

subsystem.

DA

C /

AD

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I

Q

...

XFP

XFP

XFP

XFP

SSMF

WM

...

WM

10G N RZ

...

10G N RZ

10G N RZ

10G N RZ

XFP

XFP

XFP

XFP

Optical System

N x 10/25G N RZ

10G N RZ

...

10G N RZ

10G N RZ

10G N RZ

SerD

es

N x NRZ

DSP

SerD

es

Radio System

Figure 9: IM/DD solution based on PICadvanced NG-PON2 transceivers.

For the “central office-to-radio” direction, N parallel bit streams reach a Serializer /

Deserializer (SerDes) device, which converts the N parallel 10 Gbps / 25 Gbps data signals

into 10 Gbps signals. The de-serialized 10 Gbps NRZ signal lanes will then feed the different

XFP modules which will modulate different optical carriers.

Each XFP transmits on a different optical wavelength, resulting in multiple wavelengths at

10 Gbps each (WDM network). The wavelength multiplexer (WM) is used to combine all

wavelengths, and at the receiver side each one is then filtered in order to be directly

detected at the optical transceiver level. After photo-detection, 10 Gbps electrical NRZ

signals can then be connected to a DSP. Another option is to first convert to a different line

rate using a SerDes device in order to match the connections. At the DSP, no digital

equalization is required for the NRZ signals, therefore they can be directly linked to the DSP.

The DSP is only used to generate the in-phase and quadrature electrical signals to be

transmitted over the THz radio system.

For the “radio-to-central office” direction, the radio signal is affected with dispersion or

phase noise, and therefore, before being transmitted over the optical link, it must be

compensated in the DSP. The ADCs are used to digitize the in-phase and quadrature

electrical signals. After the DSP, the signal can then be sent to the optical link using multiple

10 Gbps electrical NRZ signals. The optical transceiver solution is also based on multiple

XFPs, each one transmitting on a separate optical wavelength, and the WM is used to

combine all wavelengths.

At the fibre, the “radio-to-central office” direction signal is composed by multiple 10 Gbps

NRZ wavelengths. Note that the “central office-to-radio” direction wavelengths must be

different from the “radio-to-central office” direction wavelengths in order to avoid non-

linear crosstalk. At the receiver side, each wavelength is firstly filtered using an optical WM

and then detected by the XFPs.

The advantages of this technology are the following:

Low cost solution. Since the idea is to parallelize a high electrical bandwidth signal

(100G/800G) into multiple 10 Gbps signals, the transceiver solution is cost-effective.

In addition, the transceiver is fully based on IM/DD components;

High flexible optical solution. The aggregated data rate can be easily extended by

increase the number of parallel XFPs.

And the disadvantages are:



Low spectral efficiency. Since the modulation format is NRZ, and a wavelength guard

band must be used to separate the different wavelengths (e.g. 100 GHz as the used

in the NG-PON2 technology), the optical spectral efficiency is low;

Fibre length links up to 20/40 km. In case the NG-PON2 standard is used as baseline,

maximum reach would be 40 km as the band of transmission exhibits chromatic

dispersion.

3.2.2 Proposal 2 – IM/DD using COTS 100G

Commercial off the shelf transceivers for 100 Gbps applications with IM/DD already exist in

the market - such as CFP to CFP4 form factors - and present a concurrent solution for

Proposal 1. Although this solution is available, it can be more expensive than solution for

Proposal 1, when applications for 10 km+ or 100 G+ are needed. A possible architecture of

this Proposal is depicted in Figure 10.

DA

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AD

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I

Q

I

QCFP

SSMF

100G N RZCFP

Optical System

N x 10/25G N RZ100G N RZ

SerD

es

N x NRZ

DSP

SerD

es

Radio System

Figure 10: IM/DD solution based on CFP transceivers.

Although quite simple, the solution does not offer high flexibility. A commercial off the shelf

transceiver is employed, and thus the solution is fixed at 100 Gbps, not allowing

parallelization since the wavelength is fixed. The purpose of the DSP is to generate and

demodulate the in-phase and quadrature signals from the radio system, since the optical

interface is based on NRZ signals. This solution can be also limited at <20 km due to the

chromatic dispersion.

3.2.3 Proposal 3 - IM/DD transceivers based on amplitude modulation

Proposal 3 comprises a similar scenario to Proposal 1, however in this case direct amplitude

modulation is applied from the IQ radio signals to the optical transceiver. The main

advantage for this application is that the IQ signals from the THz antenna can be directly

mapped into the optical transceivers, but this comes at the expense of fiber distance due to

dispersive distortions. The optical link would be reduced in comparison to the 20/40 km

presented for Proposal 1.

Figure 11 shows the proposed optical architecture. The idea is to use two optical

wavelengths per direction to transmit both in-phase and quadrature signals. Highly linear

IM/DD transceivers must be used to achieve full amplitude modulation.



Radio System

SSMF

WM

Optical System

I

Q

IM/DD

IM/DD

IM/DD

IM/DDWM

I

Q

DA

C /

AD

Cs I

Q

N x 10/25G N RZDSP

Figure 11: IM/DD solution based on amplitude linear transceivers.

For the “central office-to-radio” direction, the DSP is used to generate the in-phase and

quadrature electrical signals, and each signal is modulated in a different wavelength by using

for example a Mach-Zehnder Modulator (MZM). The WM is used to combine both

wavelengths, and at the receiver side both are direct detected. After optical detection, the

in-phase and quadrature signals are available to be transmitted to the radio system.

For the “radio-to-central office” direction, the concept is similar, and therefore another two

wavelengths are used to transmit the in-phase and quadrature radio signals. Since both

radio signals reach the optical system affected with dispersion or phase noise, at the

receiver side both signals must be sampled and compensated in DSP. The DSP could be

located directly after the radio system or after the optical system, in both cases it is required

to demodulate the in-phase and quadrature signals.

This technology is cost-effective since only four IM/DD optical transceivers are required and

has a great advantage, which is its easy scalability for higher symbol rates. Modulation

schemes of higher order such as 256QAM are in principle transparent for this technology, if

the linearity of the IM/DD transmission is sufficient. Furthermore, since both in-phase and

quadrature radio components are high bandwidth signals (>16 GBd), they are affected by

chromatic dispersion in the optical link thus the typical optical reach for such scenario would

be less than 10 km.

For the TERRANOVA project, and to the best of our knowledge, there is no commercial off

the shelf solution available for this approach. Thus, this proposal requires further

investigation, since most of the similar configurations in the literature are based on the radio

over fibre (RoF) applications using for instance orthogonal frequency-division multiplexing

(OFDM). In [3-2], experimental results of transmitting several digital 16QAM RoF signals with

high spectral efficiency were discussed, based on Sub-Carrier Multiplexing (SCM) techniques.

At the transmitter, a simple MZM is employed, and at the receiver, the optical signal is

converted to the electrical domain by a photo receiver. Due to the SCM, the IM/DD system

corresponds actually to a heterodyne detection, and the transmitted signal is available in

both amplitude and phase in the electrical domain at the receiver. Therefore, no DSP

subsystem is required for optical equalization and only filtering subsystems to improve the

signal-to-noise ratio (SNR) are used.

Since the idea of our proposed solution is based on transmitting two electrical baseband

signals with amplitude modulation, i.e. both I/Q radio signals carrying multi-level modulation

(which were previously slight distorted over the radio channel), an approach widely

investigated that can be compared to the proposed is based on transmitting pulse amplitude

modulation (PAM) signals using IM/DD transceivers. For instance, in [3-3] a single optical

carrier with a single polarization 56 GBd PAM8 signal combined with raised cosine shaping



and pre-emphasis is propagated over 80 km fibre using simple IM/DD transceivers. However,

to mitigate the chromatic dispersion induced over the 80 km link as well as bandwidth

limitations, a 3-tap feed-forward equalizer (FFE) is applied at the receiver. If the symbol rate

is decreased and lower distances were used, the FFE may be avoided. In addition, in [3-4]

and [3-5], Nyquist-shaped PAM signals are also employed using IM/DD transceivers to

achieve single data rate lines of >100 Gbps for shorter-reach applications. Also, in [3-6] 56

Gbps PAM signals are demonstrated for inter-data centre connection optical networks over

a 100 km optical link. All these configurations are using amplitude equalizers (e.g. FFE) at the

receiver side to improve the eye diagram performance. This could also be applied to this

proposal since the radio signal should be restored (and equalized) in the DSP located at the

central office.

3.2.4 Proposal 4 – Optical Coherent transmission

In applications where fibre cannot be employed and an “over the air” fibre extension is

needed (TERRANOVA “fibre extender” application), analog coherent transmission becomes

an interesting solution. This solution potentially reaches up to hundreds of kilometres. In a

best case scenario, it only requires the DSP+ADC/DACs to be located in the central office to

jointly recover the signals and mitigate wireless and optical impairments. While these kind of

transceivers and DSP for coherent optical links are usually more expensive than the

incoherent counterpart e.g. in Proposal 3, this disadvantage might be compensated by the

superior performance allowing to bridge higher distances and use higher bit rates and

spectral efficiencies. Furthermore, the relative DSP complexity required for the pure THz link

in comparison to the hybrid optical – THz link is still to be explored.

The Figure 12 shows a possible application of the coherent optical solution.

Figure 12: Possible application of the coherent solution.

In order to evaluate the optical system configuration, Figure 13 depicts details of the

technology for a single-polarization (single IQ) THz frontend. At the optical transmitter side,

the in-phase and quadrature signals are directly connected to one of the two IQ modulators

of the coherent ADO transceivers pluggable.



SSMF

Optical System

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Qx

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Figure 13: Coherent solution based on single polarization THz radio interface.

During the propagation over the optical link, the signal is affected by polarization rotations.

Therefore, from the “central office-to-radio” direction, the signal generally reaches the

optical coherent receiver as a dual-polarization signal, i.e. distributed over both

polarizations. Before being transmitted to the radio channel, the single-polarization signal

must be reconstructed in a coherent DSP. Traditionally, the coherent DSP includes the

following subsystems [3-7]:

Pre-processing subsystems with a matched or a low-pass filter in order to mitigate

noise interferences;

Amplitude normalization subsystem to improve the dynamic range of the DSP;

Clock recovery algorithm towards to compensate the ideal sampling instant of the

ADCs;

Static equalization to compensate the chromatic dispersion;

Adaptive equalization to compensate the rotations of the state-of-polarization (SOP)

of the optical signal;

Carrier frequency and phase recovery to compensate the frequency and phase noise

between the transmitted and the received local oscillators lasers, respectively.

For the “radio-to-central office” direction, the DSP located on the central office side is

expected to compensate both radio and optical interferences.

A second approach of the coherent system relying on a dual-polarization (double IQ) THz

frontend is shown in Figure 14. From the “central office-to-radio” direction, instead of

transmitting a single polarization signal, a dual-polarization signal from the DSP directly

feeds the two optical IQ modulators of the ACO pluggable, and both signals are propagated

over the optical link in separated polarizations. At the optical receive side, the two received

optical polarizations can then be in principle directly connected to the dual-polarization THz

front-end without a coherent DSP, while the joint impairments of the optical-THz link are

compensated at the central office DSP, including the polarization rotations in the optical link.

Optionally, a coherent DSP can be still used at the THz frontend.



SSMF

Optical System

Ix

Qx

Iy

Qy

Ix

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Iy

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Optional

Figure 14: Coherent solution based on the dual-polarization THz radio interface with optional DSP at the radio front end.

3.2.5 Conclusion

Each of these proposals have different application scenarios, so the choice of the application

will directly choose the media converter type for TERRANOVA. Proposal 3 is the simplest

solution, but it may be limited by the fibre reach and symbol rate operation, and due to the

lack of commercial off the shelf transceivers requires further investigation. Proposal 1 may

be the cheaper solution by using multiple XFP transceivers, but a DSP between the radio and

the optical system is required to enable an on-off keying (OOK) optical system. A similar

issue is observed with the Proposal 2, but it is further limited in flexibility providing a fixed

100 Gbps lane.

For high data rate throughput, the coherent solution may be the most attractive solution,

providing high spectral efficiency by using advanced modulation formats, highly extended

reach by using advanced DSP, and high sensitivity and wavelength selectivity provided by the

coherent detection, while this might come at the cost of a higher implementation

complexity, in particular with regard to the DSP with the associated ADC and DAC devices.

3.3 Concepts for Future Media Converter Integration

One of the objectives of WP5 is to investigate how to co-integrate a state-of-the-art high

bitrate optical transponder and a THz wireless frontend in a compact and cost-efficient unit.

In that context, it is also of interest to explore advanced electro-optical packaging solutions,

considering also the progress of PICs (Photonic Integrated Circuits). Integration aspects of

the coherent optical transmission (Proposal 4) are discussed in this section since it targets to

use optical components of the shelf with little modifications. The coherent hybrid wireless-

optical link promises the highest spectral efficiency by using two polarizations in the THz

wireless domain. However, the incoherent optical transmission of the IQ baseband signals

can be also expanded to four optical channels (corresponding to four wavelengths). In this

case, the coherent and incoherent optical transmission lead to the same analog THz

frontend architecture, though the specifications on the electrical analog interface deviate in

detail. The incoherent optical link and its interface is scientifically more interesting since it

allows to develop customized solutions which exploit the progress of PICs and new

packaging technologies, and may be in that sense more disruptive.

The block diagram in Figure 15 shows the assignment of the functions of the TERRANOVA

media converter to the host board and transponder module. The THz wireless frontend sits

on the host board and emulates some of the functions that a bridge would have. The host

board is not part of a switch but is actually part of an antenna feed. Physical requirements



on the size of this “THz frontend host board” will be investigated in the next phase of the

project, as part of the antenna design. The illustration of the CFP2-ACO module is taken from

[3-8], where more details on electrical and mechanical specifications can be found.

Figure 15: Functional block diagram of the media converter for coherent optical transmission.

Figure 16 shows the cage system of the host board as proposed in [3-9]. The connectors on

the host board are specified as part of the OIF Implementation Agreement for CFP2-ACO

modules. The pluggable is guided by the cage and guarantees repeatable and reliable

contact with the board-to-board connector.

Figure 16: Illustration of a typical ACO module and the cage system on the host board [3-9] .

The typical accepted performance of the connector is also specified in [3-9], derived from

measurements on test board. The reported small signal transmission (S21) of the de-

embedded connector is depicted Figure 17. From those specifications, the connector can be

used up to about 20 GHz, which suits the FDD frequency plan of TERRANOVA (see also

Section 4) with 40 GHz of RF bandwidth per channel. When targeting baseband bandwidths

>20G, plugs and host connectors need to be used for the 56 GBd standard with no cross

mating. Further experimental development work of this electro-optical integration approach

has started as part of work package WP6.



Figure 17: Insertion loss of the board-to-board connector on the host board, taken from [3-8].

Suitable board materials and transmission line technologies for the electro-optical

integration platform were considered. As soon as the interconnections extend over longer

distances of more than a few centimetres, the transmission line losses become important

and severely limit the system bandwidth. Figure 18 shows a test module, which was

designed and manufactured for characterizing the developed 4-channel baseband amplifier

chips of Section 4 using digital test signals. The module was used to get experimental

experience of the detailed problems. An insertion loss of 10 dB was measured at 30 GHz.

However, the 1-dB bandwidth should reach at least 20 GHz in TERRANOVA, the 3-dB

bandwidth should reach 40 GHz to be compliant with the CFP2 insertion loss requirements

and its future evolution. This sets the specifications for the new integration platform,

considering also the need to comply with the CFP2 board-to-board connectors (in terms of

thickness and trace metallization).

Figure 18: Manufactured baseband amplifier test module for risk and problem identification.



The module in Figure 18 was assembled using standard Ultralam material from Rogers

Corporation of 100 µm thickness (further referred to as “LCP100”). For the selection of

future board materials and the development of the integration platform, the transmission

line losses of different board materials are compared with on-chip transmission line losses in

Figure 19. Focusing on LCP50 and LCP100 material (microstrip lines on 50 and 100 µm thick

Ultralam single sided boards), which was previously used as antenna material up to 100 GHz,

it becomes obvious that the board thickness (or the ground-to-signal spacing) is the most

dominating parameter that determines the losses of a given transmission line technology.

This is also noticeable for on-chip transmission lines, e.g. coplanar transmission lines on

GaAs substrates, using a ground-to-ground spacing of 20 µm and 50 µm (GaAs20 and

GaAs50 in Figure 19).

Figure 19: Measured transmission line loss using different printed circuit board materials.

From this experimental study for risk mitigation, there are two routes that will be further

investigated in the second phase of WP5 for the integration platform of the TERRANOVA

media converter.

The first approach is using thick board materials compliant with the CFP2 standard. This

requires the use of different transmission line technologies, instead of microstrip lines,

dielectrically filled centred striplines will be investigated. For further reducing the losses of

filter sections, suspended striplines will be also considered. Both transmission line types

have frequency limitations, mainly set by the width (a) of the rectangular waveguide and the

associated cut-off frequency, which is a function of the height (b) for a>b. The height

determines the transmission line losses to first order, and a typical width-to-height ratio a/b

> 1.5 is targeted. Figure 20 compares the cut-off frequency of the centred striplines and

suspended striplines, when using LTCC (Low Temperature Co-Fired Ceramics) and Rogers

PCB materials (RO4350, 5880). All the materials have different dielectric constants. LTCC is

an attractive option due to its possibilities to realize 2.5D packages and the thermal and

mechanical stability. For future extension of the baseband signal bandwidth, the cut-off

frequency should be at least 60 GHz, limiting the minimum achievable losses.



Figure 20: Calculated frequency limitations of the centred striplines as a function of the waveguide thickness.

The second approach further investigated is to miniaturize and co-integrate all functions as

close as possible. Hybridly integrated transponder modules are typically surface mountable

on PCB boards, integrating PICs for wavelength multiplexing and laser modulator integration

with high speed driver electronics chips and passive filter components. [3-10] gives an

overview of the history and state-of-the art in packaging photonic components up to 2015.

The photograph of Figure 21 from [3-11] shows a typical image of a TOSA / ROSA module

(transmitter/receiver optical sub assembly) as a reference design, which is still widespread in

use. This approach is good start for prototyping and validating first system concepts. In a

next step, the test modules can be re-designed using more advanced integration concepts.

The next steps in WP5 will be to develop hybrid modules for testing in back-to-back

configuration the TERRANOVA media converter with the two outlined packaging

approaches.

Figure 21: State-of-the-art of hybrid integrated optical transponder modules, from [3-11].



3.4 Conclusions and Outlook

This section has addressed the WP5 objective of the proposal to explore different concepts

for the optical link between a baseband unit and a remote THz antenna. Four proposals

were considered in more detail. The coherent optical link and the transmission of the IQ

signals of the THz wireless link by an IM/DD analog optical link were identified as the most

promising candidates for the implementation in the next phase of the project. Another

related objective of the work package is the evaluation and testing of approaches for the co-

integration of optical transponders and THz wireless frontends. One approach is to work

with COTS transponder modules for which implementation details and specifications were

developed. The second approach is to use or develop customized optical-electrical

components. This approach suits the exploitation plan of the consortium partners very well.

In the next phase of the work package, proof-of-concept test modules for both ideas will be

designed.

3.5 References

[3-1] I. P. Kaminow, T. Li, and A. E. Willner, Optical Fiber Telecommunications. B: Systems

and Networks, 2008, vol. V.

[3-2] P. P. Monteiro et al., "Mobile fronthaul RoF transceivers for C-RAN applications,"

2015 17th International Conference on Transparent Optical Networks (ICTON),

Budapest, 2015, pp. 1-4.

[3-3] M. Chagnon et al., "Single Carrier 168 Gb/s PAM8 over 80 km Below HD-FEC Using

Simple Receiver Equalization for Data Centre Interconnects," 2017 European

Conference on Optical Communication (ECOC), Gothenburg, 2017, pp. 1-3.

[3-4] N. Kikuchi et al., "Intensity-modulated / direct-detection (IM/DD) Nyquist pulse-

amplitude modulation (PAM) signaling for 100-Gbit/s/λ optical short-reach

transmission," 2014 The European Conference on Optical Communication (ECOC),

Cannes, 2014, pp. 1-3.

[3-5] M. Xiang et al., "Single-Lane 145 Gbit/s IM/DD Transmission With Faster-Than-

Nyquist PAM4 Signaling," in IEEE Photonics Technology Letters, vol. 30, no. 13, pp.

1238-1241, July1, 1 2018.

[3-6] S. Yin et al., "100-km DWDM Transmission of 56-Gb/s PAM4 per λ via Tunable Laser

and 10-Gb/s InP MZM," in IEEE Photonics Technology Letters, vol. 27, no. 24, pp.

2531-2534, Dec.15, 15 2015.

[3-7] S. J. Savory, "Digital filters for coherent optical receivers," Optics Express, vol. 16, no.

2, pp. 804-817, January 2008.

[3-8] OIF Optical Internetworking Forum, “Implementation Agreement for CFP2-Analogue

Coherent Optics Module”, IA # OIF-CFP2-ACO-01.0, January 2016.

[3-9] “CFP2 Hardware Specification,” CFP Multi-Source Agreement (MSA), Rev. 1.0, July

2013.

[3-10] Z. Zhang, et. al.,“Packaging investigation of optoelectronic devices,“ Journal of

Semiconductors, vol. 36, no. 10, 2015.

[3-11] Kish F A, Nagarajan R, Joyner C H, et al. „100 Gb/s (10 _ l0 Gb/s) DWDM photonic

integrated circuit transmitters and receivers,“ Conference on Lasers & Electro-Optics

(CLEO), CMGG3, 2005.



4. RF FRONTEND AND ANTENNA PROTOTYPES

TERRANOVA’s research and development on RF frontend components is driven by the

objective to increase the spectral efficiency and the integration density of THz wireless

transceivers. Both aspects are crucial for their commercial success, considering the progress

of wireless technologies at lower frequencies, for example at the E-band (71-76, 81-86 GHz),

which promise longer distances and the use of higher order modulation schemes. For this

reason, operating wireless systems at THz frequencies aims to the exploitation of enormous

spectral resources available for LMS/FS (land-mobile / fixed services) at higher integration

densities as of today’s available frontend solutions.

A new generation of THz frontend MMICs (Monolithic Microwave Integrated Circuits) will be

developed in the course of the project for that reason. The circuit designs use the

InAlAs/InGaAs metamorphic HEMT (high electron mobility transistor) process of the

Fraunhofer IAF. The transistor gate length is 35 nm. Currently, this node is the most

advanced transistor node at the Fraunhofer IAF though its BEOL (Back End of Line)

interconnection possibilities are rather limited. In the first phase of WP5, from M1-M12, the

objective of increasing the integration density was addressed by the development of a new

4L BEOL process that replaces the traditional 2-layer/ airbridge technology found in standard

commercial III-V technologies at lower frequencies. Two wafer fabrication runs were

dedicated to that process development. For that purpose a wafer split was introduced,

meaning that wafers of the batch were taken out of the standard fabrication flow for

process development. Motivation and aspects of this work as well as first circuit results will

be presented.

The initial frequency plan of TERRANOVA considers FDD with DL and UL frequencies at 220 –

260 GHz and 260 – 300 GHz, respectively. The design of guard bands is subject to further

investigation. The spectrum allocation as part of the new IEEE 802.15.3d standard, plans for

multiple frequencies from 252 – 321 GHz [4-1], [4-2]. This standard was also put into ITU

WRC-2019 (World Radiocommunication Conference) for consideration. The system

architectures of D2.2 will be further investigated and detailed in WP2, while WP5 will

develop a flexible analogue IP core library based on the new BEOL process. The second

phase of this work package (WP5), M13-M24, will look into the application of this library to

specific transceiver designs based. The simulation and measurement results will also feed

into system simulations in WP2 and WP3. Another major objective of WP5 in that context is

the research on the electro-optical interface for the hybrid optical-wireless architecture. A

first study has focused on the design of analogue baseband amplifiers from DC to 50 GHz

and variable gain functions.

4.1 THz Frontends for Point-to-Point Applications

The first frequency plan of TERRANOVA, reported in deliverable D2.2, considers the

frequency bands 220 – 260 GHz for the DL and 260 – 300 GHz for the UL, while part of them

will be used for guard bands. More details on the frequency plan will be derived within WP2.

In comparison, the spectrum allocation as part of the new IEEE 802.15.3d-2017 is depicted in

Figure 22, allocating 69 channels between 252.72 and 321.84 GHz that may be used. There

are ongoing discussions, as part of the ITU WRC-2019 preparation, on how this frequency

plan can be introduced into the planned regulation of the frequency spectrum 275 –

450 GHz. Due to a conflict with RAS (radio astronomy services) and EESS (earth exploration-



satellite services), it is possible that only a sub-spectrum in 252.72 – 292 GHz will be

available, instead of the full spectrum up to 321 GHz.

Figure 22: Channel allocation plan of IEEE 802.15.3d-2017 standard.

The specific design options presented in Section 3 for the implementation of the hybrid

optical-wireless link lead to different THz frontend architectures and baseband interfaces.

The incoherent optical link with analogue IM/DD transceivers for transmitting the I/Q signals

of the THz link facilitates the exploitation of polarization for realizing duplex operation of the

wireless link. However, the minimization of transmitter (Tx) to receiver (Rx) interference by

means of antenna design and signal processing, is complex. Improved system models are

required before this is addressed in more detail in WP2. The optical link with coherent

transmission (Proposal 4) exploits the polarization of the wireless link by multiple-input

multiple-output (MIMO) signal processing to increase the capacity. This is a more natural

extension of the transmission scheme of the optical link to the THz wireless link. The

corresponding frontend architecture needs to support two I/Q channels while only one I/Q

channel is required for the incoherent case.

In summary, the THz frontend design needs to address different frequency band options and

different frontend architectures. The idea of this work is to develop a rather standardized

analogue IP core library of functional blocks suitable for the implementation of the various

transceiver options of TERRANOVA. In parallel, supporting the idea of co-design of the

wireless and optical link as part of WP2, the establishment of behavioural simulation models

of all functional blocks is also targeted. The following sub-sections review and analyse

transceiver correction schemes, to be considered for the final transceiver solutions, and

derives the block diagrams of the different transceiver architectures considered in the next

phase of WP5.

4.1.1 Duplexing Techniques

The separation between DL (Down Link) and UL (Uplink) is based on a duplexing scheme:

either FDD (Frequency Division Duplexing) or TDD (Time Division Duplexing) may be

employed. In FDD, a different frequency is used for each direction, whereas in TDD, both

directions use the same frequency and DL/UL separation is achieved in time, with the frame

being divided in DL and UL sections respectively. It is apparent that FDD offers lower latency,

albeit through higher cost due to the double transceivers used, compared to TDD.



TDD basically emulates a full duplex communication over a half-duplex communication link.

TDD also offers advantages in asymmetric communications, i.e. where

the uplink and downlink data rates differ. In such cases, when the uplink/downlink traffic

increases, capacity can be dynamically allocated, and as the traffic load decreases, capacity

can be taken away. Another important advantage of TDD is the fact that since both

directions share the same frequency, due to the principle of channel reciprocity, they face

similar channel characteristics. This is especially useful in techniques that require channel

state information at the transmitter.

On the other hand, FDD is more efficient in the case of symmetric traffic, as it avoids the

overhead of switching over from Tx to Rx that TDD would require. Another advantage of FDD

is that it simplifies radio planning. By transmitting and receiving in the same frequency, the

probability of interference is minimised. Respectively, a TDD system would have to maintain

guard times between neighbouring systems, unavoidably reducing spectral efficiency.

Synchronisation must also be imposed by guaranteeing common transmit/receive reference

times, something that increases complexity and cost.

Another duplexing technique that is recently gaining attention, is the FD (Full-duplex)

communication, with simultaneous transmission and reception at the same carrier

frequency. FD communication has for long time been seen as a very promising way to boost

spectrum efficiency, but its application was not practical due to the severity of self-

interference effects. A powerful transmitter is pushing its receiver to saturation, since they

both operate at the same frequency. Recent breakthroughs in advanced self-interference

cancellation (SIC) techniques, enable the implementation of FD communications,

demonstrating an almost doubled spectral efficiency.

4.1.2 Transceiver Correction Schemes and Synchronization

The fast evolution of wireless communication systems is driving the design and

implementation of modern flexible and software-configurable radio transceivers [4-1]-[4-5].

By definition, flexible radios are characterized by the ability to operate over multiple-

frequency bands, the support of different types of waveforms, and the compatibility with

current and future air interface technologies. The terms multi-mode and multi-band are

commonly used in this context. Furthermore, TERRANOVA is expected to support high data-

rate applications and services that require efficient and low-cost wideband radio designs for

the mobile terminal [4-6], [4-7].

In this context, the well-known direct-conversion architecture (DCA) has become

instrumental for realizing compact, low-power, and low-cost transceiver designs for

wideband radio [4-8]. In direct-conversion transceivers, quadrature mixing is used, which

theoretically provides infinite attenuation of the image band and removes the need for

analogue image-rejection filtering. However, in practice, the DCA is sensitive to

imperfections of the analogue radio frequency (RF) frontend sections of the transceiver, due

to fundamental physical limitations. Such imperfections are in-phase and quadrature

imbalance (IQI), phase noise (PHN) and amplifier non-linearities [4-5], [4-9]-[4-17]. Next, we

analyse these imperfections and we provide a state-of-the-art review of the compensation

techniques that can be employed.

IQI: stems from the unavoidable amplitude and phase differences between the physical

analogue in-phase (I) and quadrature (Q) signal paths. This problem arises mainly because of

https://en.wikipedia.org/wiki/Telecommunications_link

https://en.wikipedia.org/wiki/Downlink



the finite tolerances of the capacitors and resistors used in the implementation of the

analogue RF frontend components. Although a perfectly balanced quadrature down-

conversion corresponds to a pure frequency translation, IQI introduces a frequency

translation that results in a mixture of image and desired signals. In more detail, I/Q

mismatches decrease the theoretically infinite image rejection ratio (IRR) of the receiver

down to 20 − 40 dB, resulting in crosstalk or interference between mirror frequencies [4-6],

[4-18]. Consequently, IQI degrades the effective signal-to-interference-plus-noise ratio

(SINR) and causes performance degradation. The impact of IQI is more severe in systems

employing high-order modulations and high coding rates, such as Wireless Local Area

Networks (WLANs), Worldwide Interoperability for Microwave Access (WiMAX), Long-Term

Evolution (LTE), and Digital Video Broadcasting (DVB), among other standards [4-19]. Hence,

effective IQI compensation is essential for the design of high data-rate communication

systems employing the DCA.

Various approaches have been proposed so far to eliminate, compensate, and mitigate the

effects of IQI using baseband signal processing techniques, see [4-4], [4-6], [4-20]-[4-32], and

references therein. For example, in [4-28], the authors proposed a number of pilot designs

for channel estimation in orthogonal frequency domain modulation (OFDM) systems in the

presence of I/Q mismatches at both the transmitter and the receiver. Moreover, estimation-

based system-level algorithms, including least square equalization, adaptive equalization,

and post-fast Fourier transform least squares, were proposed in [4-32] to compensate the

distortions caused by IQI. Furthermore, blind (non-data-aided) digital signal processing-

based compensation of IQI for wideband multi-carrier systems was studied in [4-4], [4-6], [4-

20], [4-33], [4-25]. Specifically, in [4-20], a digital compensation method was proposed for

MIMO systems employing space-time block coding, which is based on the algebraic

properties of the received signal combined with a suitable pilot structure, while interference

cancellation-based and blind source separation-based compensation methods were

presented in [4-25].

All previously mentioned works deal with IQI as a source of impairment that should be

compensated. In contrast to this approach, IQI at the Tx may also be seen as a source of

diversity, due to the Tx-induced mirror-frequency interference. This diversity can be fully

exploited via joint maximum likelihood (ML) detection of the signals received in the mirror

subcarriers, or partially exploited by other sub-optimal nonlinear detection techniques such

as successive interference cancellation (SIC), as was demonstrated experimentally for OFDM

in [4-34], and later confirmed in [4-35]. Still, when weighed against the implementation

complexity of nonlinear Rxs, the small achievable signal-to-noise ratio (SNR) improvement

may prove to be too expensive [4-36]. Moreover, as pointed out in several prior works

including [4-34], Rx IQI is detrimental for the outage and error performance of wireless

communication systems, regardless of the utilized detector. The reason for this is that Rx IQI

affects both the received signal and the noise; hence, it is commonly believed that Rx IQI

should be compensated [4-4], [4-6], [4-20], [4-21], [4-11], [4-25], [4-31]. Finally, [4-37]

provides a low-complexity technique that achieves a diversity gain in the presence of Rx IQI.

PHN: Noise is of major concern in local oscillators (LOs), because introducing even small

noise into an LO leads to dramatic changes in its frequency spectrum and timing properties.

This phenomenon, peculiar to LOs, is known as phase noise or timing jitter, and was

identified as one of the major performance limiting factors of communication systems in

several studies, see for example [4-12], [4-38]-[4-53]. Generally, the disturbance of the



oscillator’s output amplitude is marginal. As a result, most influence of the oscillator

imperfection is noticeable in random deviation of the oscillator’s output frequency. These

frequency deviations are often modelled as a random excess phase, and therefore referred

to as phase noise. Phase noise will increasingly appear to be a performance limiting factor

especially in the case of multi-carrier communications, when DCA implementations or

systems with high carrier frequencies are considered, since, in those cases, it is harder to

produce an oscillator with sufficient stability.

Amplifier non-linearities: Feedback, feedforward, and predistortion are the most common

techniques for the compensation of the nonlinear transmitter [4-54], [4-55], [4-56]. The

principle of the feedback technique is to force the power amplifier (PA) output signal to

follow the input signal by feeding the output signal back to the input [4-57]. The

disadvantage of the feedback technique is its sensitivity to the delays introduced by the

amplifier and associated signal processing components [4-54]. For this reason, it is not

suitable for the compensation of modern wide bandwidth transmitters. However, wider

bandwidths can be handled using, for example, a Cartesian feedback technique [4-55].

Seidel has presented the first modern feedforward compensation scheme, which is based on

the early concept of Black from 1928. The principle behind this is to have an extra error

amplifier, whose output is subtracted from the main output signal. The error amplifier is

driven by the distortions produced in the main amplifier. The feedforward scheme is

applicable to wider bandwidth signals than the feedback technique. The power efficiency of

the entire Tx may be reduced, due to the use of the extra amplifier [4-58]. Likewise, time

mismatches between the parallel branches may reduce the performance. However, research

on feedforward compensation is still ongoing.

Macdonald has demonstrated the predistortion concept in audio communication as an

alternative to the feedback technique. The feedback technique was inconvenient for the

compensation of the nonlinear distortions of the loudspeaker. In the predistortion concept,

a nonlinear functional block, i.e., a predistorter, is inserted prior to the amplifier so that the

combined transfer function of these two components is almost linear. In other words, the

predistorter behaves as an inverse of the amplifier. In the same paper, Macdonald also

discusses postdistortion, i.e., the compensator is located after the amplifier. The

postdistortion approach was not suitable for the compensation of the loudspeaker either [4-

59].

Predistortion can be accomplished in the analogue domain at radio, intermediate, or

baseband frequencies, or in the digital domain at baseband [4-60], [4-56]. The analogue

predistortion has some advantages such as low cost and less complex DSP. For very high

bandwidth radio over fibre (RoF) applications such as fibre cable television systems,

analogue predistorters are also applicable, see e.g., [4-61] and references therein. However,

more flexibility is obtained using a digital predistorter handled by a digital signal processor

(DSP) and, in addition, an adaptive predistorter may be difficult to implement in the

analogue domain. For that reason, the digital predistortion approach has gained much

attention in the recent literature [4-62], [4-63].

4.1.3 Transceiver Architectures and Project Development Plan

Different basic transceiver architectures and reference designs were reviewed in deliverable

D2.2 ([4-64], Section 4). This leads to the development of common key building blocks for



the implementation of integrated frontend MMICs. The development of a standardized well-

understood analogue IP core library facilitates the implementation of the different

transceiver schemes, reduces risks and provides system designers with behavioural models

for their research. Those behavioural models are developed and used within WP2, WP3 and

WP4. The basic development strategy of the project is illustrated in Figure 23. The design

and investigation of new frontend components is based on the development of a new 4-

layer BEOL interconnection process. Apart from the process development, new transmission

line and interconnection test structures were investigated, characterized and validated up to

320 GHz. Test transceiver components were developed in the first period of this work

package for the DL frequency band from 220 – 260 GHz: The focus was on the DL for risk

mitigation reasons and easier experimental characterization.

Figure 23: Overview of design activities of M1-M12.

The 240 GHz frontend test core is comprised of mixer, amplifier and multiplier components,

which are typically most critical in the design process. The components were tested with

different BEOL variations. In the next step higher integrated transceiver MMICs will be

developed and tested for the DL. In the same iteration, the building blocks for the UL will be

developed.

4.2 New Process Technologies for III-V based MMICs

This section reports on the progress of the BEOL process development. The first part gives

an overview of available FEOL (Front End of Line) transistor technologies and typically

available BEOL options, as a reference of the state-of-the-art. The second part provides first

results.

4.2.1 Technology Overview

The technical motivation for using a III-V based HEMT technology for prototyping the system

components in TERRANOVA is summarized with the help of Figure 24. Current RF transistor

technologies for mmWave and THz applications (status September 2017), which are either

operated in a foundry mode or in a qualified process mode with a fixed FEOL process (both

are referred to as “quasi foundry mode”), were considered in this survey. In addition, some



emerging technology candidates in research are shown. For the devices in “foundry mode”,

measured data in their BEOL environment were considered, since relevant for the final RF

performance. The device size was chosen to approximate typical device sizes used in

mmWave amplifier designs operating above100 GHz.

Figure 24: Comparison of different transistor technology options for THz frontend design.

In the representation of Figure 24, the MSG (maximum stable gain) region and the transition

frequency (“k=1” stability point) to the MAG (maximum available gain) region, plotted on a

logarithmic frequency scale are used as a key performance indicator. The inset shows the

theoretical slope in the two regions, 10 dB/dec in the MSG region and 20 dB/dec in the MAG

region. This MSG/MAG curve is a practical upper boundary for the design of broadband

systems, if they are required to be immune to process variations. Please note, no measured

data of the “k=1” point were available for the silicon technologies, and for that reason they

represent merely the highest measured points (65 GHz or 110 GHz related to the

measurement system limits) but not the actual performance limit. However, the actual

“k=1” point must be located along the slope of 10 dB/dec containing the highest measured

MSG. In this transistor technology comparison, the III-V based transistor technologies stand

out (both HEMT and HBT). The HBTs have a higher MSG but a lower “k=1” stability point,

which can be improved by a common base configuration instead of a common emitter

configuration. The 35nm HEMT process can operate between 220 – 320 GHz in its MSG

region for typical devices sizes. This leads not only to superior noise figures but also to wider

system bandwidths for a given gain. In TERRANOVA, the focus is on wideband systems,

exploiting the frequency spectrum from 220 to 300 GHz with high spectral efficiency.

Broadband noise figure and output power are key performance metrics, and for this reason,

the 35 nm mHEMT transistor technology is a good candidate for the implementation of

those systems with superior performance.



On the downside, the standard BEOL of current III-V HEMT processes limits the integration

density for systems operating above 100 GHz. Figure 25 presents a survey of typical BEOL

processes developed for RF applications on silicon (SiGe BiCMOS, SOI CMOS), and III-Vs.

Figure 25: Comparison of various BEOL processes, for different transistor technologies

While the III-V HBT process has a rather flexible RF BEOL, it lacks the complexity that silicon

technologies offer for the implementation of additional digital functions. The standard BEOL

of III-V HEMTs uses traditionally the semi-insulating GaAs (or InP) substrate for the

implementation of transmission lines. For more complex routing, and the implementation of

coplanar transmission lines, an airbridge technology is available. All other BEOL processes

use thin film transmission lines on the front side. This leads to more routing flexibility but

also to higher losses in tendency. For isolated circuit functions, e.g. a stand-alone amplifier

or mixer MMICs, the III-V HEMT BEOL is inexpensive and offers very low losses, which

benefits power and low-noise applications. Silicon BEOLs use SiO dielectrics, while the III-V

processes shown in Figure 25 use BCB (benzocyclobutene), a polymer based dielectric that

can be structured either by dry chemical etching or photolithographic patterning. The

standard Fraunhofer IAF BEOL for the 35 nm mHEMT process employs two thin BCB layers

but still uses the airbridge technology. This complicates the design of higher integrated

MMICs, and has also compatibility issues with standard RF surface mount packaging

technologies used by silicon foundries nowadays. In contrast, the standard rectangular

waveguide packaging solution often used with the III-V HEMT BEOL is well established up to

more than 300 GHz for prototyping and offers excellent and reliable performance suiting lab,

outdoor and space applications.

4.2.2 Motivation and Status of BEOL Development

TERRANOVA considers new advanced electro-optical packaging approaches for the

integration of the media converter between the optical and wireless link. The development

of a new BEOL process is one important practical step towards increasing the integration

density of today’s existing THz solutions based on the Fraunhofer IAF 35 nm mHEMT

process. The requirements on packaging demand also the replacement of the airbridge

technology for mechanical reasons and an end passivation of the top metal interconnection

lines.

Within the reporting period, different BEOL options were investigated, which are

summarized in the schematic illustrations of Figure 26. The 3L mHEMT BEOL with the

traditional airbridge technology was replaced by the 4L mHEMT BEOL. In this first step, the



airbridge technology was replaced by a layer of BCB, filling and supporting the former

airbridge with dielectric material. This allows us to use the former airbridge metal layer

(TFMS) as a full additional routing layer for RF signals. In the next step, the deposition of a

thick BCB layer on top was investigated, a via process and the deposition of a thin top metal

(ANT) to be used as a layer for antenna integration were developed for that purpose. This 4L

mHEMT BEOL was compared to another variation, the 3LPP mHEMT BEOL. The 3LPP uses a

planarizing thin BCB layer on top of the TFMS metal layer instead of the thick BCB layer.

Figure 26: BEOL development and tests in TERRANOVA, (1) base process, (2) development run 1, (3), final solution for final transceiver integration.

First investigations of antenna prototypes showed that an even thicker dielectric layer than

in the 4L process would be necessary for broadband low-loss operation. This is addressed by

the idea of transferring a separate thick antenna substrate on top. The thickness of the

antenna substrate can be adjusted according to the optimal thickness required for optimal

antenna performance. In addition, the antenna substrate can be metallized on both sides

before the substrate transfer. In the end, it was found that the 3LPP mHEMT BEOL offers

more flexibility and reduces the complexity of the BEOL process (before antenna substrate

transfer). In addition, the 3LPP process is also compatible with the standard GaAs wafer

backside process, which comprises wafer thinning to 50 µm and backside via processing

through the GaAs wafer. The backside process keeps compatibility with standard waveguide

packaging technologies for prototyping and allows to address different markets.

As part of the process development, different test circuits were designed and manufactured,

though not fully optimized, since measurement-based passive component models were not

available at this stage. For that reason, it is expected that the presented circuit level results

of Section 0 will be further improved in the next three design and manufacturing cycles

scheduled as part of TERRANOVA. A SEM image of a manufactured 3LPP test circuit is shown

in Figure 27. Since BCB is nearly transparent optically, the corresponding chip photograph

does not allow us to distinguish the different metal layers well. As the BCB is non-conductive

and charges accumulate, some of the features appear distorted in the SEM picture.



Figure 27: Left: SEM picture of manufactured mixer IP core component for 220 to 300 GHz with the 3LPP process (before TFMS end passivation), right: corresponding optical

photograph.

Based on transmission line test structures, in parallel to the process development, passive component models were extracted. At first, those models were used for monitoring the repeatability of the process and for benchmarking. At a later stage, these test structures were used to develop models for circuit designs. As an example, the measured attenuation of some of the transmission line types are compared in Figure 28, normalized to the length in millimetres. As a reference for the state-of-the-art performance of the 3L process, coplanar transmission lines on GaAs having a ground-to-ground spacing of 20 µm and 50 µm (CPW20 and CPW50 GaAs) are also added to the graph. The attenuation of microstrip lines in the 3LPP BEOL using the TFMS signal layer are compared before (3LPP) and after final passivation (3LPP-Pass). The measured performance of the 3LPP microstrip lines is comparable to coplanar transmission line references of the 3L process, and can be ranked between a CPW50 and CPW20 transmission line. Although the final passivation slightly increases the losses, the effective dielectric constant is also increased, which reduces the effective electrical length. The thin film microstrip line of the 3L process is also presented, which has the highest attenuation of all reference lines in the graph.

Figure 28: Left: Layout and photograph of manufactured transmission lines for process testing and monitoring, right: measured attenuation of a microstrip line of 10 µm width.

Summarizing the status of the BEOL development, while the initial target was the 4L BEOL,

the 3LPP BEOL turned out to be more flexible. The first results indicated that more

experience should be collected to optimize all process design rules, which has to be acquired



by more manufacturing cycles and circuit designs with increasing complexity. The chip size

can be reduced by about a factor of 4, as demonstrated on the example of three different

driver amplifiers, which operate at similar frequency. The layouts are depicted to scale in

Figure 29 with the dimensions referring to the photograph outline (including pads).

Figure 29: Integration density of different BEOL for broadband RF applications (to scale), see also [4-65] for the 65nm CMOS amplifier.

The first example at 90-110 GHz uses a 100 nm mHEMT with the 3L BEOL. The second

example at 120-150 GHz uses a 65 nm CMOS with its typical RF BEOL. These are compared

to the result of this work, at 100-145 GHz, which uses the 35 nm HEMT technology with the

developed 3LPP BEOL. The results highlight that the integration density with the 3LPP

process reaches the one of silicon CMOS designs. The 35 nm mHEMT leads to superior

linearity and higher gain per stage, which also translates into the superior noise

performance of this technology. Looking at the output power, however, due to similar

breakdown voltage Vbreak and maximum drain currents Id,max, the 65nm CMOS design

achieves an even slightly higher saturated output power than the 35nm mHEMT design. The

gain per stage of the 65 nm CMOS design corresponds reasonably well to the technology

comparison in Figure 26. For this reason, with the 35nm mHEMT, more gain can be traded

for bandwidth while maintaining the two key performance parameters, linearity and noise

figure.

4.3 New MMIC Frontend Building Blocks

The following measurement results comprise the first circuit designs for the DL, from 200-

260 GHz using the 3LPP BEOL. The building blocks tested were used to assist the

development of the 3LPP BEOL process and for setting up a design library. This design library

of functional building blocks will be the basis for the integration of the different transceivers

for the high capacity demonstrator and the beamforming demonstrator. In the first

subsection, initial test designs of baseband amplifiers for the TERRANOVA media converter

are presented. A test package was also designed for the fabricated circuits, which is reported

in more detail in Section 3.3 in the context of the co-integration of the THz wireless frontend

with the optical transponder solutions. The second subsection presents some of the



measurement results of the first designs that highlight the progress towards the WP5

objectives.

4.3.1 Component Candidates for the TERRANOVA Media Converter

For the implementation of the TERRANOVA media converter, some of the critical analogue

interface circuits were investigated, along with the possibility to integrate them using the 35

nm mHEMT process. In the current 35 nm HEMT process version, only normally-on

transistors are available. As a result of the negative threshold voltage, the gate-source

voltage required for class-A operation of a common-source amplifier is only about 100 mV.

The optimum operating drain-source voltage of a single stage is at least 800 mV. Hence, the

resulting DC potential difference between two cascaded common-source amplifier stages, is

greater than 500 mV (Fig. 1). This voltage has to be either compensated by a level-shifter

circuit or reduced by choosing specific types of circuit topologies. As a result of the low gate-

source voltage of only 100 mV, an efficient level-shifting using a multistage source follower

is not feasible. Therefore, resistively coupled amplifiers, a diode-level-shifter and a Kukielka

amplifier were investigated. In the course of TERRANOVA, the Kukielka solution was finally

selected as the solution to be further investigated and integrated. The manufactured

Kukielka test amplifier and its schematic circuit representation are shown in Figure 30.

Figure 30: Chip photograph of the fabricated Kukielka amplifier, schematic representation, and summary of the measured key performance parameters.

The Kukielka concept is a two-stage broadband Cherry-Hooper amplifier with multiple

feedback loops. The design equations based on the Kukielka configuration have been

studied in [4-65]. In contrast to the standard topology, the local series feedback loop at the

common-source input stage was omitted. This simplifies the input matching of the used

mHEMT and reduces the DC potential at the drain of transistor M1. Hence, the direct

coupling of the input stage and the output stage in the Darlington configuration is achieved

without shifting the DC level. Instead, the DC potential at the Darlington stage is increased

by the series feedback resistor Rs2.

The on-wafer measured S-parameters and noise figure of the fabricated test circuits are

presented in Figure 31. An important advantage of the Kukielka amplifier is the low noise

figure of less than 4 dB between 1 and 50 GHz, the achieved gain of 21 dB and the P-1dB

compression point of 2.5 dBm, which requires a very small chip area of 185 x 270 µm (0.05

mm²), in comparison to the other investigated concepts. The 3-dB bandwidth is 50 GHz.

State-of-the art amplifiers realized in CMOS, SiGe and InP HBT technologies achieve gain

levels above 30 dB since higher numbers of gain stages are used [4-67], [4-68]. The required

chip area of [4-68], [4-69] is larger than 0.2 mm2, since inductive peaking techniques were



used, which in contrast were omitted in the Kukielka solution. One of the critical aspects is

the DC power consumption of 200 mW per amplifier, which becomes a severe problem in

arrays using digital phase shifting, since it scales with the number of channels. For this

reason, the overall power consumption of one channel of the THz transceiver is significantly

increased by the baseband amplifiers.

(a)

(b)

Figure 31: On-wafer measured performance of the manufactured Kukielka amplifier, (a) S-parameters and noise figure, (b) output power compression characteristics at 10 GHz.

In a further step, the possibility to use the Kukielka concept in the 2-channel receiver of the

coherent optical link (with two IQ receive signals), was investigated. As part of a risk

mitigation approach in the project, a separate test chip was designed and fabricated for that

purpose. The test chip targets to get information at an early phase of the project on the

homogeneity and unexpected parasitic effects that may occur in this multi-channel

configuration. In addition, test packages were developed (see Section 3.3 for test results). A

photograph of the fabricated test chip and a CAD drawing of the test package are shown in

Figure 32. The pitch of the 4-channel amplifier was chosen to be consistent with the pitch of

the planned 2-channel receiver. The DC bias supply in the current test chip emulates the DC

supply situation in the receiver chip, where multiple-channels will share one common bias

supply. The measured small-signal S-parameters in Figure 33 demonstrate that the phase of

the frequency response of all four channels was nearly identical (less than 2 deg variation),

and the corresponding magnitude showed variations of less than 1 dB, using a single bias

supply for all 4 amplifiers.



Figure 32: Designed test package and photograph of the manufactured 4-channel Kukielka amplifier chip.

Figure 33: Measured frequency response, left: magnitude in dB, right phase in deg for the 4-channel Kukielka amplifier.

4.3.2 Circuit Components for 220-320 GHz Transceivers

The following subsection presents some of the first components that were designed,

manufactured and tested in the context of the 3LPP BEOL development. The components

comprise different frontend functions that need to be accurately implemented and fine-

tuned for a successful transceiver design in the next step. All functions were measured on-

wafer, and can be used to develop behavioural models for system simulations. In addition,

the measured results can be used to improve the passive and active device models for the

next circuit design phase.

The motivation for the design of a library containing all major frontend functional blocks in a

standardized way, consistent with the objective to develop phased array transceivers for

220-320 GHz, is depicted in Figure 34. The floorplan of the multi-channel transceiver suits a

minimum antenna array pitch of 500 µm, which is half a free-space wavelength at 300 GHz.

For this reason, all functional blocks operating at 220-320 GHz and the first 1st subharmonic

of 110-160 GHz were designed to have a height of 225 µm. A corridor of 25 µm is reserved



for the bias supply of the functional blocks. Functions operating at the 2nd subharmonic (55-

80 GHz) are designed to fit into a height of 450 µm respectively 900 µm since they will drive

multiple channels in array architectures and can consume more space for that reason. This

leads to a grid plan that simplifies the implementation of different transceiver architectures.

Figure 34: First IP core library for the DL, MMIC for generating an LO signal that can be tuned varied from 200 to 260 GHz.

The on-wafer measured performance of the MMIC for generating a local oscillator signal

between 200 and 260 GHz is depicted in Figure 35. The maximum output power at 240 GHz

was about 8 dBm at a DC drain bias voltage of 1.2 V.

Figure 35: On-wafer measured output power at 240 GHz for the LO chip

The measured performance of the 120 GHz driver amplifier and the frequency doubler from

120 to 240 GHz are shown in Figure 36 and Figure 37. In a similar way, all other functional

blocks were characterized as additional breakout circuits.

The measured S-parameter and output power characteristics of the 3-stage 120 GHz driver

amplifier are shown in Figure 36. The amplifier provides 28 dB of small signal (SS) gain from

105 - 140 GHz. The input / output matching was better than 10 dB over the defined

frequency band. The gain flatness deviates slightly between scalar measurements and vector

network analyser measurements due to different source power levels and the on-wafer

calibration limitations by parasitic mode excitation and lack of probe isolation. Transistors in

common source configuration were employed. The output stage had a total gate width of

200 µm for the targeted operation resulting in a saturated output power of 12 dBm and a P-

1dB output compression power of >10 dBm, at a DC drain bias of 1.2 V.



The on-wafer measured performance of the driver amplifier, followed by a standardized

frequency doubler function, is shown in Figure 38 and Figure 39. The combination of both

functions was verified to operate in 220-290 GHz, allowing to select arbitrary channels in the

IEEE 802.15.3 channel plan, as well as the TERRANOVA FDD frequency plan.

Figure 36: Chip photograph of manufacture 120 GHz driver amplifier and measured RF performance (S-parameters and output power characteristics).

Figure 37: Chip photograph of the fabricated broadband LO multiplier for selecting different Rx/Tx channels from 220 to 290 GHz and measured output power characteristic at a fixed

input power level.

The 3-stage topology of the 120 GHz driver amplifier was also selected for the

implementation of a 240 GHz output amplifier for a Tx chip. The amplifier consumes an area

of 500 x 225 µm. A photograph of the dedicated on-wafer test structure is shown in Figure



38. The calibration reference plane is at the probe tips, both for the S parameter

measurements and the scalar power measurements with an Erickson PM4 calorimeter. The

total gate width of the output stage was reduced to 120 µm. The SS performance of the 240

GHz power amplifier exhibited a gain of 20 dB with a flatness of better than 1 dB from 210 -

250 GHz (Figure 39). A 3 dB bandwidth of nearly 60 GHz with an input/ output matching of

better than 10 dB was achieved. The compression characteristic of the output power versus

the input power is provided also in Figure 39 for a frequency of 240 GHz. The impact of

different DC drain bias conditions on the output power and P-1dB output referred

compression point, are compared in this graph. The saturated output power at the probe

tip, can be increased by 1 dB to 10 dBm when increasing the DC drain voltage from 1.0 V to

1.2 V. The loss due to the access lines from the probe tip to the amplifier cell was estimated

to be 1.0 dB. However, this access line for testing is replaced by a similar transmission line in

future integrated transceiver MMICs to connect to antennas or waveguide transitions. For

this reason, the measured output power equals the power that is also expected in a

transceiver circuit. The P-1dB output referred compression point was better than 4 dBm.

Figure 38: Chip photograph of the fabricated 220 -260 GHz power amplifier for the DL frequency band, left 3LPP design, right 4L design with MET4 layers (before MET4 processing)

.

Figure 39: On-wafer measured RF performance of the 220 -260 GHz power amplifier, left S-parameters, right output power at 240 GHz.


The presented work in progress has covered three objectives of WP5 towards new THz

frontend technologies. The first objective is the development a new BEOL process that

addresses the requirement to increase the functional integration density of current THz

frontends. A first version of this BEOL process was established but needs further testing at

circuit design level to eliminate all potential bugs. Instead of the initially proposed 4L BEOL,

the 3LPP BEOL was established, after first experiments with the 4L BEOL. The 3LPP BEOL has

actually several advantages in comparison to the initial proposal. The main advantage is its

compatibility to various packaging processes including split-block waveguide packages. The

second objective was the development of first building blocks for the implementation of



new transceiver architectures. Here, accompanying the BOEL process development,

functions for the DL were developed in a first step. In a next step, their modelling and the

design of a first library of RF building blocks for the UL will be designed and tested. The final

step will be the implementation of the transceiver solutions for the different proposed

demonstrators of WP6, including the beamforming demonstrator. From a timeline, this

seems to fit quite well, since the system proposals are gaining more detailed specifications,

while at the same time the THz IP core library is growing by all components required. The

third objective is the investigation of the analogue baseband interface for the TERRANOVA

media convert. First circuit candidates were designed, manufactured and compared. The

most promising candidate was the Kukielka amplifier concept, which was also tested as a 4-

channel version.

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estimation and decoding for MIMO systems in the presence of phase noise and

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with phase noise over fading channels," IEEE Trans. Commun., vol. 56, no. 7, p.

1119–1126, 2008.

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systems with phase noise," IEEE Trans. Broadcast., vol. 50, no. 2, pp. 97-106, 2004.

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Magazine, vol. 2, no. 4, pp. 37-49, 2001.

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Sevic and N. O. Sokal, " Power amplifiers and transmitters for RF and microwave.,"

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2014.



5. BASEBAND DIGITAL SIGNAL PROCESSING FOR THZ

SYSTEMS

In this Section 5, we describe the work in progress and the first experimental results for Task

5.3 entitled “Baseband signal and code design for THz systems”. In particular, the main goals

of the first investigations were: (1) the basic characterization of the THz frontend from a

system point of view, (2) the validation of the prototype single-carrier digital signal

processing algorithms for the THz link, and (3) the assessment of the maximum achievable

bitrate over the THz P2P link with the current module generation.

5.1 THz P2P transmission experiments

As a first step, experiments at Fraunhofer HHI were performed for the simplest case, i.e. for

the pure THz P2P link without additional optical links. In this investigation, the currently

available THz module generation from Fraunhofer IAF was used. In the following, the

experimental setup, the DSP used for evaluating the system, and the experimental results

will be presented.

Figure 40: Experimental setup for tests of THz P2P link

Figure 41: Photograph of first lab setup at Fraunhofer HHI

DAC Att.

Att.

Tx module x12 x3

f = 8.11 – 8.71 GHz

f = 2.625 GHz

φ d link

Rx module

Scope

x12 x3

Offline DSP

10 MHz Reference

Att

.



5.1.1 Experimental setup

The construction of the first THz setup is illustrated in the block diagram in Figure 40 and the

photograph in Figure 41. The main experimental parameters are summarized in Table 1.

Table 1: Main experimental parameters

Symbol rate 16 GBd 32 GBd

Modulation format 16QAM

Overhead 28% (FEC + Framing)

Forward Error Correction SD-FEC with threshold BER of 2·10-2

Pulse shape RRC 0.35

Spectral width 21.6 GHz 43.2 GHz

Gross bit rate 64 Gb/s 128 Gb/s

Net bit rate 50 Gb/s 100 Gb/s

The baseband transmitter comprised a digital-to-analog converter (DAC) operating at 84

GS/s (8 bit, 3-dB bandwidth of about 25 GHz) which generated two electrical signals

representing the in-phase (I) and quadrature components (Q) of a single-polarization

quadrature amplitude modulation (QAM) signal. Throughout the investigations, 16QAM

modulation was used due to its higher spectral efficiency as compared to 4QAM (4 bits per

symbol for 16QAM vs. 2 bits per symbol for 4QAM) and the higher vulnerability to

component nonlinearities. The symbol rates under test were chosen to be 16 and 32 GBd,

which correspond to 64 Gb/s and 128 Gb/s raw bit rate, respectively. Assuming the use of a

soft-decision forward-error correction (SD-FEC) scheme with a threshold BER of 2·10-2, the

net bit rates correspond to 50 and 100 Gb/s, respectively. To limit the spectral extent of the

modulated signal without imposing a large peak-to-average power ratio (PAPR), a root-

raised cosine (RRC) pulse shape was used with a moderate roll-off value of 0.35. Digital pre-

emphasis could be applied to compensate for transmitter bandwidth limitations.

The DAC output signals were then applied to the I and Q inputs of the TX module comprising

a direct-conversion I/Q mixer. The output swing was in the order of 500mVpp. In order to

operate the mixer in the linear regime, fixed 3 dB attenuators were placed at the output of

the DAC. For characterization purposes, the drive signal swing could be further attenuated

by digitally reducing the DAC output amplitude at the cost of DAC resolution.

For the generation of the local oscillators, a tunable clock signal was upconverted to

approximately 300 GHz using a frequency multiplier-by-12 followed by frequency multiplier-

by-3. Initially, the clock signal was split and fed both into the Tx and Rx path to have a stable

clock relation. This allowed, by means of a manually-set phase delay φ, to control the

mapping of the transmitter I/Q components to the receiver I/Q components in most

experiments. In later experiments, the setup was changed to use two independent clock

generators in order to also test the performance with free-running carrier frequencies (not

shown in Figure 40).



Table 2: Relation of clock and carrier frequencies used in the experiment

Clock signal Generated carrier frequency

8.11 GHz 291.96 GHz

8.15 GHz 293.4 GHz

8.21 GHz 295.56 GHz

8.31 GHz 299.16 GHz

8.41 GHz 302.76 GHz

8.51 GHz 306.36 GHz

8.61 GHz 309.96 GHz

8.71 GHz 313.56 GHz

The THz P2P transmission link comprised two horn antennas with 23 dBi gain each and a 58

cm-long free-space link. In order to emulate longer link distances, a manually tunable

attenuator could be used. In the Rx module, the received signal was converted back to

baseband using a direct-conversion I/Q mixer. The I and Q outputs were captured by a digital

storage oscilloscope with 80 GS/s (8 bit, 32-GHz bandwidth), which was synchronized with

the DAC using a 10-MHz reference clock.

Finally, digital signal processing was performed offline as described in the next section.

5.1.2 Used Digital Signal Processing

The block diagram of the utilized digital signal processing is shown in Figure 42. The

transmitter DSP comprised blocks for the generation of a random bit sequence, the mapping

to the 16QAM constellation, the insertion of training sequences [5-1] at the beginning of

each frame, pulse shaping (i.e. RRC with 0.35 roll-off) and pre-emphasis. The receiver DSP

comprises blocks for receiver-side front-end I/Q corrections [5-2], a training-aided block [5-

1] including: frame synchronization, carrier-frequency offset compensation and T/2-spaced

equalization (i.e. feed-forward, 51 taps), carrier-phase estimation (i.e. blind-phase search [5-

3] with 32 angles and 128 symbols per block), T-spaced equalization of Tx I/Q impairments

[5-4] (i.e. LMS adaption, 201 taps), demapping, decision and, finally, BER counting. The main

parameters for the DSP are also summarized in Table 3.



Figure 42: Block diagram of the digital signal processing

Table 3: Parameters for digital signal processing blocks

Parameter Value

Sequence length per frame 215 symbols

Length of training sequence 448 symbols

Tap number of T/2 spaced equalizer 51 taps

Block length for carrier-phase estimation 128 symbols

Number of test angles for carrier-phase estimation 32 angles

Tap number of T-spaced I/Q equalizer 201 taps

5.2 First THz System Measurement Results

The following subsection presents the first measurement results. Throughout the

experiments, the antennas were optimally aligned. In the first step, a symbol rate of 16 GBd

was chosen, corresponding to a net bit rate of 50 Gb/s. Figure 43 shows the measured BER

as a function of the digital DAC amplitudes of I and Q (127 was the maximum value for a

resolution of 8 bit). The curves were extracted without utilizing pre-emphasis and at a carrier

frequency of 306.36 GHz. The Tx output was maximized (no additional THz attenuation) and

the phase between the Tx and Rx local oscillator was optimized (optimal Rx I/Q orientation).

The curves indicate that the optimum DAC amplitudes are 75 and 70 for the I and Q

components, respectively, indicating a small Tx I/Q imbalance. In this case, a BER of 1·10-3

was achieved which is well below the threshold BER of the SD-FEC, i.e. error-free

transmission with 50 Gb/s net rate can be expected after FEC decoding. For higher DAC

amplitudes, the performance degrades, which is attributed to nonlinear distortion due to

saturation of the direct-conversion mixer. For lower DAC amplitudes, the degradation is

attributed to a lower SNR.

Map

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Pre

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asis

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ann

el

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C

DA

C

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orr

ecti

on

s

Fram

e S

ynch

ron

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ion

Car

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-off

set

com

pen

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on

Trai

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ided

T/2

-sp

aced

Eq

ual

izer

Car

rier

-Ph

ase

Esti

mat

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Car

rier

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ase

Esti

mat

ion

T-sp

aced

I/Q

Eq

ual

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Dem

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ecis

ion

BER

co

un

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Tx DSP Rx DSP

Ran

do

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ion



Figure 43: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd using 16QAM (without pre-emphasis, at maximum Tx output power, optimal Ry I/Q orientation, at a

carrier frequency of 306.36 GHz)

Figure 44: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd 16QAM (with pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier

frequency)

In Figure 44, results from similar measurements are plotted, but now a digital pre-emphasis was applied in order to flatten the Tx transfer function. In this case, the optimal digital DAC amplitudes are shifted towards higher values, and the BER is improved to 6.5·10-4, i.e. by almost a factor of 2. The shift of the optimal DAC amplitudes is attributed to the smaller average power of the digital waveform, as the digital pre-emphasis is realized by attenuating the lower frequency components in order to raise the level of the higher frequency components.

1,00E-04

1,00E-03

1,00E-02

55 65 75 85 95 105

BER

Digital DAC Amplitude (Quadrature Component)

60

65

70

75

80

85

90

95

100

105

Digital DAC Amplitude(Inphase Component)

1,00E-04

1,00E-03

1,00E-02

55 65 75 85 95 105

BER


60

65

70

75

80

85

90

95

100

105




16 GBd 16 QAM without pre-emphasis 16 GBd 16 QAM with pre-emphasis

Figure 45: Received constellations under best case conditions. (a) 16 GBd 16QAM without pre-emphasis @ BER = 1·10-3. (b) 16 GBd 16QAM with pre-emphasis @ BER = 6.5·10-4.

Figure 46: BER vs. Carrier Frequency for 16 GBd 16QAM (no pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, optimal digital DAC amplitude)

Figure 46 shows the dependence of the optimal BER on the carrier frequency, without pre-

emphasis. There are two distinct minima at 295.56 GHz and 306.36 GHz, while the BER is

slightly increased within this frequency range. Outside of this range, the BER is increasing

monotonically. The operation window with a BER below 1·10-2 is about 20 GHz, i.e. from

293.4 GHz to 313.56 GHz.

1,00E-03

1,00E-02

1,00E-01

290,00 295,00 300,00 305,00 310,00 315,00

BER

Carrier frequency [GHz]



Figure 47: BER vs. Rx I/Q orientation for 16-GBd 16QAM (different carrier frequencies, no pre-emphasis, maximum Tx output power, optimal digital DAC amplitudes)

Figure 47 shows the dependence of the BER on the Rx I/Q orientation angle, without pre-

emphasis, as controlled by the manual phase shifter in the Tx local oscillator path. It can be

observed that the BER is strongly depends on the orientation angle while the angles with

best and worst performance are independent of the carrier frequency. The reason for this is

the Rx mixer saturation as the peak amplitudes of the I and Q components at the Rx depend

on the I/Q orientation. This is schematically depicted in subfigures (a) and (b) of Figure 48,

where it is shown how the received I and Q peak amplitudes are increased by a factor of

in case of a Tx-Rx misalignment with an orientation angle of 45°, i.e. the peak power

increases by a factor of 2. In Figure 48 (c) and (d), measured RX constellations are shown at

an orientation angle giving the best case BER (c) and the worst case BER in (d). To illustrate

the effect, only the I component was transmitted in both cases. As expected from the

discussion above, the measured orientation angle corresponds to 0° when a best case BER is

achieved, while for the worst case BER, the orientation angle equals 45°.

Figure 48: Discussion of Rx I/Q orientation. (a) Schematic of Rx aligned QAM constellation (orientation angle = 0°). (b) Schematic of Rx misaligned QAM constellation (orientation angle = 45°). (c) Measured Rx constellation (only I component at Tx) at an orientation angle giving the best case BER. (d) Measured Rx constellation (only I component at Tx) at an orientation

angle giving the worst case BER.

1,00E-03

1,00E-02

1,00E-01

0 0,2 0,4 0,6 0,8 1 1,2 1,4 1,6 1,8 2

BER

RX I/Q Orientation Angle [rad/(pi/4)]

291,96

293,4

295,56

299,16

302,76

306,36

309,96

313,56

Carrier Frequency[GHz]

A I

Q

A A

I

Q

A

(a) (b) (c) (d)



Figure 49: BER vs. THz attenuation for 16-GBd 16QAM (306.36 GHz carrier frequency, with pre-emphasis, optimal digital DAC amplitudes, optimal Rx I/Q orientation)

Figure 49 shows the dependence of the BER on the additional attenuation in the THz free-

space link for different digital DAC amplitudes (each optimized in terms of Tx I/Q imbalance)

and with digital pre-emphasis. It can be observed that, due to the high margin towards the

threshold BER of the SD-FEC, significant additional attenuation can be tolerated, indicating

the potential for an increased free-space distance.

To further investigate the performance of the THz system, the setup was reconfigured in

such a way that two independent synthesizers were connected to the transmitter and

receiver modules, respectively. Since the frequency of the synthesizers’ clock signals drifts

slightly over time, random phase noise is introduced on the received constellation resulting

in a time-dependent RX I/Q orientation angle. Its compensation requires a carrier phase

estimation algorithm. Furthermore, considering that both devices are not synchronized,

there usually exists a small difference in the frequency both transmitter and receiver

modules operate on, which is denoted as frequency offset. It translates into a fast rotation

of the received constellation over time. In order to be able to receive the transmitted signal

correctly, this offset has to be compensated by means of a frequency-offset compensation

algorithm. The performance the applied DSP compensation methods for both phase noise

and carrier frequency offset is depicted in Figure 50. It can be observed that the DSP is able

to correctly compensate up to a discrepancy of almost 4 GHz between the operating

frequencies of the transmitter and receiver modules, resulting in a slight BER degradation

when compared to the previous experiments. All DSP parameters remained the same except

for two: the block length for carrier phase estimation was set to 2048 symbols and the

number of test angles for carrier-phase estimation was changed to 64. This was done to

better compensate the increased phase noise experienced by the data samples in this case

and improve the overall performance of the system.

1,00E-04

1,00E-03

1,00E-02

1,00E-01

27,532,537,542,547,5

BER

Attenuator setting

70/60

75/65

85/75

85/75

85/80

90/85

95/90

100/95

105/100

105/105

Digital DAC

Amplitudes (I/Q)



Figure 50: BER vs. RX frequency offset for 16-GBd 16QAM (306.36 GHz carrier frequency, with pre-emphasis, maximum TX output power, optimal digital DAC amplitudes)

Finally, Figure 51 shows measurement results for the transmission of a 32-GBd 16QAM

signal with a net data rate of 100 Gb/s. For this experiment, however, the setup was rolled

back to the one-synthesizer configuration, where a single device generates the clock signal,

which is then fed both to the transmitter and receiver modules. Pre-emphasis was applied in

order to optimize the transmit spectrum. Compared to the results for 16 GBd shown in

Figure 44, the optimal digital DAC amplitude is shifted towards higher values with optimal

performance at the maximum available amplitude. Thus, the performance might be further

optimized by removing the fixed 3-dB attenuators at the DAC output. This shift in required

DAC amplitude is attributed to the enhanced pre-emphasis as compared to the case for 16-

GBd. At the optimal DAC amplitudes of 125/125, a BER of 1.1·10-2 was measured. The

corresponding constellation is shown in Figure 52. This means that an error-free

transmission with a net data rate of 100 Gb/s was shown in this experiment.

Figure 51: BER vs. Digital DAC Amplitude (I and Q component) for 32-GBd 16QAM (with pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier

frequency)

1,00E-02

1,50E-02

2,00E-02

2,50E-02

3,00E-02

75 80 85 90 95 100 105 110 115 120 125 130

BER


80

85

90

95

100

105

110

115

120

125




Figure 52: Received constellation under best case conditions for 32-GBd 16QAM with pre-emphasis @ BER = 1.1·10-2.


In this section, we documented the first TERRANOVA experiments on single-carrier high

symbol rate transmission with high spectral efficiency over a THz free-space link. As a main

result, the applicability of the prototype single-carrier DSP to the THz wireless transmission

was demonstrated under realistic experimental conditions. In particular, the DSP algorithms

were shown to efficiently combat bandwidth limitations, I/Q impairments, frequency offset

and phase noise. Along with the basic characterization results, several issues with current

THz module generation were found, e.g. the dependence of the BER on the carrier frequency

as well as on the Rx I/Q orientation angle, which will be addressed in the next generation

chips as well as through further calibration measurements and procedures. Nevertheless, an

error-free 100 Gb/s THz transmission using a 32-GBd 16QAM signal on a 300 GHz carrier was

demonstrated with the current module generation, showing the high potential of this

technology to reach the envisioned project goals.

5.4 References

[5-1] R. Elschner, F. Frey, C. Meuer, J. K. Fischer, S. Alreesh, C. Schmidt-langhorst, L. Molle,

T. Tanimura, and C. Schubert, “Experimental demonstration of a format-flexible

single-carrier coherent receiver using data-aided digital signal processing,” Opt. Exp.,

vol. 20, no. 27, pp. 28786–28791, Dec. 2012.

[5-2] I. Fatadin, S. Savory, and D. Ives, “Compensation of quadrature imbalance in an

optical QPSK coherent receiver,” IEEE Photon. Technol. Lett. 20(20), 1733–1735

(2008).

[5-3] T. Pfau, S. Hoffmann, and R. Noé, “Hardware-efficient coherent digital receiver

concept with feedforward carrier recovery for m-QAM constellations,” J. Lightwave

Technol. 27(8), 989–999 (2009).

[5-4] C. R. S. Fludger and T. Kupfer, “Transmitter Impairment Mitigation and Monitoring

for High Baud-Rate, High Order Modulation Systems”, Proc. ECOC 2016, p.Tu.2.A.2.



6. PHASED ARRAY BEAMFORMING

6.1 State-of-the-Art in Phased Array Beamforming techniques

Beamforming emphasizes signals in the desired direction while suppresses signals in other

directions. An array of antenna elements (AEs) is required in order to implement

beamforming and concentrate the transmitted/received power towards a particular

direction and, thus, to increase the antenna gain. The ideal baseband digital beamforming

requires one distinct radio frequency (RF) chain per antenna element. However, this

requirement is prohibitive in terms of complexity, because mmWave as well as THz

communications potentially utilize large antenna arrays to compensate the severe path

losses with highly directional transmissions [6-1]. To this end, hybrid beamforming has

gained the interest of the scientific community with the ideal requirement of using as many

RF chains as the desired data streams. Hybrid beamforming combines digital with analogue

beamforming and can achieve a performance close to the ideal fully digital beamforming

with the benefit of reduced hardware complexity and power consumption [6-2], [6-3], [6-4].

In the analogue domain, the phase shifters more commonly found in the literature are

digital, due to their immunity to noise present in voltage control lines. Interestingly, Chen et

al. in [6-5] stated that graphene transmission lines have low insertion losses in the THz

frequencies. There is ongoing research on the phase shifters to be utilized at high carrier

frequencies [6-6], while the key challenges include the insertion losses, the cost and the

complexity of their design [6-7]. Analogue beamforming is cheaper, simpler and less power

consuming than digital beamforming, but cannot be applied at wideband systems due to the

frequency-dependent nature of the phase shifters. Digital beamforming can be performed

over wide bandwidths, both in the time domain with tapped delay line filters and in the

frequency domain by using a fast Fourier transform. The research interest in digital

beamforming concentrates on algorithms to improve the computation time and the signal

tracking.

In [6-8], it has been proven that the optimal architecture for hybrid beamforming should

meet a trade-off between complexity and performance, while it highly depends on the

application and the channel conditions. Interestingly, determining the optimal number of RF

chains is a complex, but very challenging optimization problem [6-3]. In the literature, there

exist two main hybrid beamforming architectures: a) the full-connected, where each RF

chain is connected to all the AEs and b) the sub-array, where each RF chain is connected to a

group of AEs [6-3], [6-1]. Another possible hybrid beamforming architecture referred to as

virtual sectorization structure can be found in [6-8]. This structure creates multiple “virtual

sectors” in the digital domain, so that each set of RF chains is processed separately in

baseband in order to reduce overhead and complexity. Interestingly, 3GPP Release 13 for

LTE-Advanced Pro [6-9] is related to the sub-array hybrid beamforming architecture, while

Release 14 [6-10] includes developments that are closer to the 3D hybrid beamforming, i.e.

higher resolution.

The THz beamforming demonstrator will most likely consist of four horn antenna elements.

Since hybrid beamforming is a meaningful compromise in terms of complexity when the

number of antenna elements is large, digital or analogue beamforming are more suitable in

our case. Besides, motivated by the fact that using four RF chains is not prohibitive, digital

beamforming in the baseband seems to be a promising technique for the demonstrator. As



already mentioned, digital beamforming results to better performance, offers more degrees

of freedom and is superior in terms of reliability compared to analogue beamforming.

6.2 Mathematical Model of Phased Array Architectures

The THz beamforming demonstrator consists of an RF-frontend that operates in the

frequency band above 275 GHz. The relationship of the signal bandwidth with the carrier

frequency determines whether a system is considered as wideband or narrowband. When

the bandwidth of the signal is adequately small compared to the carrier frequency, the

system is considered narrowband. This assumption is not well defined and many variations

exist in the literature. In general, it holds true when the signal bandwidth is less than one

percent of the carrier frequency. In our case, the signal bandwidth, B, of the mmWave

modem is inherently small compared to the carrier frequency, fc (which is around 300 GHz),

and thus the system is considered narrowband. Interestingly, this assumption allows us to

approximate the propagation delays of the impinging signal between the antenna elements

with phase shifts.

For the beamforming demonstrator, the study is concentrated on uniform linear arrays

(ULAs), where the AEs are located along a line and placed in equal distance one from the

other. This inter-element spacing, d, is considered the spatial equivalent of the time interval

and, similar to the Nyquist–Shannon sampling theorem, it should hold that d ≤ λ/2, in order

to avoid the formulation of grating lobes [6-11], [6-12]. Grating lobes are lobes that have the

same radiated power as the main lobe but in a different direction. However, lowering the

element spacing conflicts with the desire to have as large aperture as possible for a fixed

number of elements. To this end, we generally set d = λ/2, as in the case of a ULA the

aperture equals the distance between the first and the last element of the array.

The angle depicted in Figure 53, is the direction from which the signal is received, while

the propagation speed equals the light speed c. Since the antenna elements are equally

spaced, the propagation paths between two adjacent elements differ by d sin that results

in a time delay equal to . For a narrowband system, this delay corresponds to a

phase shift equal to 2 π d sin / λ.

In order to ease the analysis, the centre point of the phased antenna array has been utilized

as the reference point. Therefore, for a ULA with M AEs, the array response vector or

steering vector [6-12], if M is even, equals

where T stands for the transpose.



Figure 53: Plane wave impinging on a ULA

6.3 Comparison of Beamforming Techniques

The following beamforming techniques equally concern transmit (Tx) and receive (Rx)

beamforming. It should be mentioned that adaptive beamforming is more commonly

applied at the receiver, while it can also be applied at the transmitter in the presence of

feedback. For uniformity reasons, we selected to focus on Rx beamforming in the presented

simulations. The desired signal corresponding to a specific direction, i.e. the angle , is

extracted by implementing a certain weighting on the received signals before adding them.

In the following, beamforming techniques candidates for implementation in the

demonstrator are presented and compared.

6.3.1 Conventional Beamforming

Conventional beamforming utilizes fixed and predefined phase shifts that are chosen

independent of the signal received by the antenna array [6-11]. The beamforming weight

vector is the conjugate transpose of the steering vector at divided by , i.e.,

where H stands for the conjugate or Hermitian transpose. The division with is introduced

to satisfy the following equation . It should be noted that in the case of a ULA,

the beamformer is a spatial filter having a direct analogy to an FIR frequency-selective filter

at the time domain. In order to evaluate the performance of a beamformer, we utilize the

beampattern which is equal to , where . In Figure 54, the

beampatterns for different and values are depicted. The larger lobe of the

beampattern is steered at and we refer to it as mainlobe, while the lower lobes are

known as sidelobes. Due to selection of antenna spacing, grating lobes are not present.

Conventional beamforming would be perfectly sufficient, if the only signal present at the

receiver aside from the additive thermal noise was the signal of interest from . However,

in many cases, there exist signals propagating in the same carrier frequency and impinging

on the array from different angles. We refer to these signals as interference. In many cases,

we utilize adaptive methods in order to identify and overcome these interferers. However,

conventional beamforming can be further optimized in order to mitigate the effect of

interference to some extent.



Figure 54: Beampatterns for conventional beamforming

6.3.2 Tapered Beamforming

The phases of the weight vector steer the mainlobe to the desired direction and, thus,

emphasize impinging waves at this angle. It has been proven that the sidelobe levels of the

beampattern can be further reduced by tapering the weights of the conventional

beamformer vector. More specifically, a vector t with non-negative real values is utilized and

the tapered beamformer vector equals

,

where represents the Hadamard product, i.e., the element-by-element multiplication of

the two vectors. The vector t is determined by the selection of a proper window from the

existing ones in the literature, where the window length is set equal to the number of

antenna elements M. Besides, the taper vector values may be distributed according to the

binomial distribution. Utilizing the binomial distribution has the advantage of completely

eliminating the sidelobes with the drawback of significantly increasing the mainlobe width

(see Figure 55).

Fixed windows, such as the Hanning and the Hamming windows, result in a fixed main

sidelobe level, which does not depend on the number of elements. However, there also exist

windows, such as the Taylor, the Kaiser and the Dolph-Chebyshev windows, where the main

sidelobe level is a design parameter and can be selected. In Figure 56, various windows

are implemented and the corresponding beampatterns are extracted assuming a ULA with

16 AEs and a desired beamsteering angle . Interestingly, a tradeoff exists between

the main sidelobe level and the width of the mainlobe. More specifically, as the main

sidelobe level decreases, the mainlobe width increases. This interplay is one of the

important issues during the selection of the appropriate window, which is mainly driven by

the actual application.



Figure 55: Beampatterns for tapered beamforming with binomial distribution and 4 AEs



Figure 56: Beampatterns for tapered beamforming with various windows. The ULA consists of 16 elements and its steering direction is at in (d) is the number of nearly

constant-level sidelobes adjacent to the mainlobe

Below, there are some characteristics of the most useful windows.

The Hanning window corresponds to a constant -32 dB main sidelobe level, while it

reduces the number of the sidelobes.



The Hamming window corresponds to a -43 dB main sidelobe level, but has more

sidelobes than the Hanning window.

The Blackman-Harris window almost eliminates the sidelobes, but results in a wide

mainlobe.

The Taylor window offers the ability to select the number of nearly constant-level

sidelobes adjacent to the mainlobe expressed by the symbol in Figure 56.

However, the adjacent sidelobes have lower level than the more distant ones.

The Dolph-Chebychev window has an “equiripple” behavior, which means that it

results in sidelobes that all have the same desired level.

The Kaiser window decreases the level of all the sidelobes without changing their

number. Moreover, it results in a lower mainlobe width increase when compared to

the Taylor and the Dolph-Chebyshev windows using the same desired main sidelobe

level.

Consequently, applying a window, results in a wider mainlobe with respect to the

conventional beamforming. Hence, the sidelobe level reduction can be achieved only at the

expense of the resolution. Furthermore, utilizing windows seems to be more efficient in the

case where many sidelobes exist and the mainlobe width is narrow. This holds true

particularly when an antenna array with an adequate number of antenna elements is

utilized.

6.3.3 Null-Steering Beamforming

When the direction of interference is known, should be selected so that nulls are

introduced in the direction of the interfering signals. If is the desired signal direction and

are the directions of the L interfering signals, should be selected to satisfy

These constraints can be rewritten

as , where and are defined as follows

By using the method of Lagrange multipliers, can be obtained from the following equation

In Figure 57, the derived beampattern of the null-steering beamforming with 16 AEs is

depicted. It can be easily observed that a null has been introduced at , both in the

case where the beamsteering direction is and . Interestingly, null-steering

beamforming may not always be a good design. More specifically, if the null direction is

close to the desired mainlobe direction and/or the number of antenna elements is small, the

steering capability of the ULA is tremendously reduced. This can be easily verified from

Figure 58, where introducing perfect nulls at the direction of interference has a detrimental

effect on the mainlobe, which does not point exactly at the desired direction.

6.3.4 Adaptive beamforming

In adaptive beamforming, the weight vector is derived by maximizing the theoretical signal-

to- interference-plus-noise ratio (SINR) and is implemented in the digital domain (i.e, digital



beamforming). This design is considered more efficient in the sense that it balances between

the minimization of noise and interference from signals impinging on the array from

directions different than the desired one [6-12]. Maximizing the SINR is identical to

minimizing the interference-plus-noise power, which assuming a weight vector , equals

, where stands for the interference-plus-noise correlation matrix. In

practice, a priori knowledge of the second order statistics of the array data is not feasible. To

this end, techniques to estimate the unknown statistics from the received array signals are

required. An estimate of the interference-plus-noise correlation matrix requires that

no desired signal is present. To this end, the desired signal transmission is muted and

samples of the array received vector are collected. The estimate of the correlation matrix is

given by the equation below

,

where selecting a higher value for results in a better estimate of the correlation matrix.

Figure 57: Beampatterns for Null-steering beamforming with 16 AEs



Figure 58: Beampatterns for Null-steering beamforming with 4 AEs and

6.3.4.1 Minimum Variance Distortionless Response (MVDR) Beamformer

The maximization of the SINR is achieved by solving the following optimization problem

where the constraint preserves the desired signal. The solution of this constrained

optimization problem is found by using Lagrange multipliers and is given by

6.3.4.2 Linearly Constrained Minimum Bariance (LCMV) Beamformer

Some applications may require additional conditions on the beamformer, which in most

cases concern the rejection of interference signals received from specific angles. If is the

desired signal direction and are the directions of the L interfering signals,

should satisfy the following equations

Thus, the LCMV beamforming minimizes the undesired signal output power of the

antenna array, while preserving or nulling the power in selected directions. The set of

constraints can be formulated in a proper equation as where is the constraint

matrix and the constraint response vector, which in our case are defined as follows

Hence, the constrained optimization problem can be expressed as

while the LCMV beamforming vector is given by



In the following, the conventional beamformer is compared to the adaptive MVDR and

LCMV beamformers, when ULAs with 4 and 16 AEs are utilized, respectively. We assume that

the desired direction is at , while there exist two interference signals at

and with mean power 0.1 and 0.2, respectively. Finally, it has been assumed that

the existence of interference at is known, while the one at is not. To

this end, for the LCMV beamformer, it holds that

From Figure 59−Figure 63, it becomes evident that the adaptive beamformers outperform

the conventional one in the presence of interference. The SINRs of the two configurations

for the three different beamformers are listed in Table 4. It should be noted that the LCMV

beamformer completely eliminates the incoming interference at . Besides, when

compared to the MVDR beamformer, LCMV beamformer completely eliminates the

interference at but does not combat the interference at , as efficiently

as the MVDR beamformer (see Figure 61).

Table 4: SINR for various beamformers

SINR [dB] Conventional MVDR LCMV

4 AEs 4.58 4.99 5.16

16 AEs 12.05 12.35 12.35

Table 5: Output noise power for various beamformers

Output noise power Conventional MVDR LCMV

4 AEs 0.1149 0.1476 0.1314

16 AEs 0.0207 0.0221 0.0221

Table 6: Interference power for various beamformers

Interference power Conventional MVDR LCMV

4 AEs 0.1571 0.0267 0.0492

16 AEs 0.0058 0.0001 0.00002



Figure 59: Magnitude of the beamformers’ output signals for a ULA with 4 AEs

Comparing Figure 59 and Figure 60 with Figure 62 and Figure 63, it is obvious that the

efficiency of the LCMV beamformer is higher compared to that of the MVDR beamformer,

when the number of utilized antenna elements is low. More specifically, it has been

observed that when 16 AEs are utilized, the MVDR and LCMV beamformers perform almost

identically. It should be emphasized that with adaptive beamforming, a balance between the

rejection of interference and the output thermal noise is achieved. Thus, the resulting

output noise does not cause a reduction in the SINR (see Table 4, Table 5 and Table 6).

Figure 60: Power pattern of beamformers for a ULA with 4 AEs



Figure 61: Power pattern (rectangular) of beamformers for a ULA with 4 AEs

Figure 62: Magnitude of the beamformers’ output signals for a ULA with 16 AEs



Figure 63: Power pattern of beamformers for a ULA with 16 AEs

6.4 Beamforming Implementation Issues of Demonstrators

6.4.1 Available Phase Shifter Resolution for Analogue Beamforming

Assuming that a discrete codebook has been designed to cover a large range for the desired

angle direction, each codeword of the codebook corresponds to a set of beamforming

phases, which should be applied by phase shifters [6-13]. When analogue beamforming is

implemented, the utilized phase shifters are of finite limited resolution. This indicates that

the applied phase shifts are not always equal to the ideal ones; instead the closer possible

value that is supported by the phase shifter is applied. As expected, this causes a variation in

the performance of the beamformer often referred to as quantization error, whose

importance is related to the resolution of the phase shifters. Hereby, it should be

emphasized than an -bit phase shifter has a resolution or step equal to .

From the beampatterns of Figure 64 and Figure 65, it is obvious that the phase shifter

resolution does not affect all the desired directions to the same extent, i.e., for the

beampattern is not affected at all because the optimal phase shifts are supported by the

phase shifters, i.e., they are integer values of . Furthermore, when the mainlobe width is

large, as in the case of 4 AEs, the difference in the performance can be considered negligible.



Figure 64: Beampatterns for conventional beamforming with and without quantization assuming 4 AEs



Figure 65: Beampatterns for conventional beamforming with and without quantization assuming 16 AEs

6.4.1 Differential Phase Noise in THz Phased Array Systems

The role of the local oscillator phase noise in phased array systems was experimentally

studied using a 4-channel testbed for the frequency range from 275 GHz to 325 GHz at the

initialization of the project [6-14]. The testbed itself was developed in the national funded

project TERAPAN. In this work, local oscillator phase shifting was initially identified as an

interesting solution for broadband phase shifting at THz frequencies. Although this leads to a

similar frontend architecture as required for digital phase shifting of baseband signals, the

main difference is that it would require only a single data stream and a single ADC/DAC. This

becomes attractive for THz high-data rate applications.



Since low-phase noise fundamental oscillators at THz frequencies are difficult to realize, the

more promising approach is the frequency multiplication of a low frequency LO reference

signal. The generation of a local oscillator signal of set phase relation to a stable reference

crystal oscillator can be realized with the help of direct digital synthesis (DDS) circuits. By

synchronizing DDS circuits, a multi-channel local oscillator signal can be generated at very

low frequencies with set phase difference. Those LO signals can be multiplied in frequency

with integrated frequency multipliers to 300 GHz. The block diagram of this scheme is shown

in Figure 63 for the used 4-channel phased array testbed. The output frequency of the DDS

circuits was 2.083 GHz.

Figure 66: Functional block diagram of the 4 channel LO beamformer with DDS based phase shifting. The synchronous DAC output is achieved by a master trigger (omitted in this

drawing for clarity).

The measured phase noise of the individual channels at 8.333 GHz after frequency

multiplication-by-4 is shown in Figure 67.

Figure 67: Phase noise of the individual channels of the LO beamformer, measured at the output carrier frequency of 8.333 GHz. In comparison, the phase noise of a commercial

frequency synthesizer is shown. Carrier power Pc = -2 dBm in all cases.

The phase noise of the individual channels was nearly identical. While the single channel

phase noise was comparable to the phase noise of a commercial frequency synthesizer, the

differential phase noise between different channels, which becomes amplified by the

frequency multiplication, is very critical for the stability of phased array systems. Although

splitting the LO at very low frequencies is a rather worst case scenario, interconnect losses at



THz frequencies do not allow the splitting and distribution of LO signals over longer

distances. The question on what differential phase noise can be tolerated, and what the

lowest frequency is to best split the LO signal to the individual channels is currently

unresolved and subject to the next step of this work. This question is for LO phase shifting

and digital phase shifting of the baseband signals the same.

The differential phase noise degrades the beamforming efficiency by modifying the shape of

the mainlobe. Eventually, the effect of phase noise is obvious in the reduction of the

achievable SINR and consequently, in the decrease of the achievable data rate. For the

current four channel receiver at 300 GHz, data transmission experiments with 1, 2, 3, and 4

receive channels were conducted at a symbol rate of 4 Gbaud. The performance of a single

channel is shown in Figure 68. For the transmission experiments with multiple channels, the

transmit power was reduced by a factor of 4. Figure 69 shows that the beamforming

efficiency is reduced and phase noise effects start to evolve with increasing the number of

receive channels though the EVM is numerically the same as in the single channel case

(about 13%rms).

Figure 68: Measured cumulative constellation diagrams and EVM of a single receive channel at 4 Gbaud for increasing QAM modulation depths. 100 constellation diagrams of 4096

symbols were accumulated in each plot.

Figure 69: Measured cumulative constellation diagrams and EVM of a single receive channel at 4 Gbaud for a 16-QAM modulation format, after initial calibration by null-steering. 100

constellation diagrams of 4096 symbols were accumulated in each plot.



6.4.2 Four Element Horn Antenna Array

In the following, the horn antenna array utilized by the TERRANOVA demonstrator was

simulated in MATLAB. It should be emphasized that the carrier frequency is set equal to 200

GHz, which is the maximum permitted in MATLAB, as the actual value (i.e., 300 GHz) is not

supported. Therefore, these results are only indicative of the way the demonstrator antenna

array is expected to work. The dimensions of the antenna elements are summarized in Table

7, while the inter-element distance is set equal to C+0.25 mm.

Table 7: Horn antenna element dimensions

Figure variables Dimension Value

B Horn aperture width 3 mm

C Horn aperture height 1 mm

F Horn flare length 3.577 mm

D WR3 wave guide width 0.8640 mm

E WR3 wave guide height 0.4320 mm

d Inter-element spacing C+ 0.25 mm = 1.25 mm

Figure 70: Horn antenna element

Figure 71: Directivity pattern of the horn antenna element

In Figure 71, the directivity pattern of a horn antenna element is depicted, when its horn

aperture width is placed along the z axis. Placing the horn antennas as depicted in the left

bottom part of Figure 73 and steering the array at 0⁰, the directivity pattern of every single

element is depicted in Figure 72, while that of the array is depicted in Figure 73. Obviously,

the directivity patterns of the middle and the edge elements are symmetric to each other,



respectively. Moreover, beamsteering is implemented in the elevation plane as depicted in

Figure 74. However, given that λ=1mm at 300GHz, the fact that the inter-element distance is

higher than λ/2 degrades the efficiency of the beamforming as grating lobes are created.

Through these indicative simulations, it became evident that the steering range of the

demonstrator’s array is not expected to be higher than [-45⁰, +45⁰].

Figure 72: Directivity pattern of the four elements of the ULA

Figure 73: Directivity radiation pattern of a ULA with 4 AEs pointing at



(a) Coordinate system: polar

(b) Coordinate system: rectangular

Figure 74: Directivity radiation patterns of a ULA with 4 AEs pointing at

6.4.3 Beam Search and Alignment

In order to implement beamforming, it is required that the transmitter establishes a link

with the receiver, i.e. their beams are steered pointing at each other. Communication links

established at THz frequencies are directional, which results in the need for a different

procedure that the ones commonly used in the microwave systems where synchronization

signals are broadcasted omnidirectionally. As the demonstrator consists of 4 AEs in a ULA,

the beamsteering will take place either in the azimuth or the elevation direction. Moreover,



the scanning range of the array to significantly less than 180⁰ due to the inter-element

spacing and the utilization of directional horn antennas as AEs.

Designing a codebook, which contains multiple beams in order to cover the feasible angular

space (i.e., the scanning range of the array), makes a directional search possible. To this end,

an exhaustive search will be implemented by the demonstrator [6-15]. More precisely,

utilizing all the codewords one-by-one, the demonstrator will scan the whole space and

determine the codeword, and thus the direction, that results in the highest received SINR

value. Although this method is regarded as time consuming, it is efficient to be utilized by

the demonstrator mainly due to the small codebook size. Interestingly, as the half power

beamwidth of the four element horn array is expected to be close to 15⁰, the number of

codewords will probably be less than 15.

Hence, the beam search procedure significantly depends on the beamforming algorithm that

will be implemented at the demonstrator.

6.5 Calibration Techniques for Phased Array Antennas

In this section, a survey on calibration techniques for phased antenna arrays operating at

mmWave and lower frequencies is presented. It should be noted that techniques for the

calibration of systems operating at THz frequencies have not been presented in the

literature yet. During the calibration, the amplitude and phase degradations of each array

element are measured and aligned in order for the array to match the requirements in

accuracy. Thus, the general goal is to ensure that the amplitude and phases applied to each

element are corrected in order to derive the desired radiation pattern.

The calibration technique described in [6-16] is called mutual coupling method (MCM) and

can be implemented in arrays with uniformly spaced elements that have symmetric

radiation patterns. MCM measures the mutual coupling between all adjacent elements and

aligns the insertion attenuation and phase of the array. The rotating element electric field

vector (REV) method presented in [6-17] is a practical self-calibration method for a phased

array, based on measurements of the array’s power variation while each element phase is

shifted. A faster, more steady and accurate method has been proposed in [6-18], where

calibration is achieved by sequentially measuring the near-field mutual coupling signal

between each active antenna element and the passive element used for calibration. The

measured data, i.e. the ones calculated in orbit, is compared with the stored data, i.e. the

reference values supplied at the manufacturing, to make the required compensations.

Two novel on-site calibration algorithms that compensate the mutual coupling effect as well

as gain, phase and location errors of active antenna arrays in satellite communications have

been presented in [6-19]. The experimental demonstration has been conducted at the L-

band (around 1.6 GHz). Besides, the calibration technique in [6-20] is based on digital signal

processing of the received signal and is intended for a satellite acquisition system. The array

composed of eight radiating elements operating in the S-band (i.e., 2.2–2.3 GHz). The

realization of a specific radiation pattern is achieved by properly adjusting the elements’

phase shifters, attenuators, and the division ratios of the power dividers to provide the

required excitations in phase and amplitude. A genetic algorithm (GA) with an antenna

measurement facility has been implemented in [6-21] for the optimization of the radiation

patterns by using the actual phase and amplitude states of the RF components.



A cost-efficient calibration concept based on semi-passive backscatter transponders and an

internal power detection circuitry has been proposed in [6-22]. Using external feedback

through BTs, a fully flexible array design is achieved that scales well for an increasing

number of antenna elements. Thus, this approach is suited for online calibration of massive

MIMO systems. In [6-23], an amplitude-only measurement method for phased array

calibration has been proposed, measuring the signals (i.e., complex electric field) of

individual antenna elements. This method has the advantage of minimum measuring time,

while the demonstrated system is operating at 2 GHz. In [6-24], all the antenna element

signals are simultaneously measured by an auxiliary antenna as the antenna element phase

settings change. The calibration method is based on calculating the antenna elements’

excitations by solving linear equations.

Low-cost manufacturing tests and calibration procedures of phased arrays are critical to

enable applications at mmWave and higher frequency bands. A built-in self-test technique

for phased arrays has been proposed in [6-25]. Orthogonal code modulation is applied to

each antenna element and parallel measurements are conducted. The test signals are then

down-converted to a baseband interference signal consisting of code-modulated complex

cross correlations between all element signals. Using orthogonal code products, each cross

correlation is extracted from the interference signal and used to obtain amplitude and phase

data of each element. The demonstrated system of [6-25] is operating at 57 to 67 GHz. A 60

GHz phased array with four elements employing LO phase shifting has been studied in [6-

26]. Beamforming calibration has been achieved by performing gain equalization, I/Q

calibration and successive –approximation phase tuning.

Digital beamforming provides enhanced calibration capabilities, resulting in ultralow

sidelobes and wideband equalization [6-27]. A wideband closed-loop adaptive calibration

method for digital beamformers has been presented in [6-28] for operation from L- to X-

band. The channels are equalized relative to a selected channel before the implementation

of beamforming, using an internally distributed and injected additive white Gaussian noise

source. In [6-29], hybrid beamforming is implemented at the receiver side with a calibration

technique referred to as image injection (IMI). In IMI, the nonlinear behavior of the mixer is

utilized and the inherent LO distribution network is multiplexed as the injection path of the

pilot signal. The IMI calibration is implemented in real time at the IF stage and targets the

electrical performance variation of the RF circuit, thus, no dedicated transmitter is required.


Digital and analogue phase shifting techniques have been discussed and evaluated through

simulations. Although the most prominent technique in the recent literature is the hybrid

beamforming, this technique is not appropriate the THz beamforming demonstrator based

on the existing antenna array. Besides, analogue phase shifters for the frequency band

between 275 and 300 GHz are not fully studied in the literature and are prone to severe

phase noise. This is expected to degrade the system performance in terms of both the

beamforming efficiency and the achievable data rate. To this end, the effect of phase noise

and other impairments introduced by the analogue components will be further examined in

the study to be conducted in the forthcoming months. Moreover, the existing system setup

and the conducted simulations reveal that digital beamforming is expected to perform

better for the THz beamforming demonstrator. This approach proposes utilizing element-

level processing with a dedicated RF chain at each antenna element. Based on the



conducted evaluation and the derived results, digital beamforming algorithms will be further

investigated. It should be emphasized that the restrictions introduced by the digital and

analogue components will be considered in order to implement the most efficient

beamforming technique for the demonstrator.

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7. CONCLUSIONS

The first deliverable of work package WP5 has reported on the advances in the design of

preliminary system components to be handed over to WP6 for implementation. The main

Sections 3-6 address the major objectives and individual tasks, which were proposed initially

at the beginning of the project.

In Section 3, the initial ideas for the hybrid optical THz wireless link, as developed in D2.2

were refined and further detailed. Part of the challenge was to identify among many

different options the most promising candidates and start investigating and implementing

first functions of the TERRANOVA media converter. In that context, most promising

integration approaches for the co-integration of state-of-the-art optical transponder

modules or functions with the THz wireless frontend were also identified. This work was

supported by some experimental implementation of critical components for risk mitigation.

The work in progress towards new THz frontend integrated circuit technologies was

presented in Section 4. Different options for a BEOL addressing the requirements of THz

applications were experimentally explored. Limitations of the 4L BEOL process were

identified and a compromise, the 3LPP BEOL process, was proposed and developed.

Although the 3LPP BEOL needs further experimental experience at circuit design level for

establishing the final design rules, first promising test circuits for the TERRANOVA downlink

(220-260 GHz) were already manufactured and tested using this BEOL.

Section 5 has experimentally investigated an existing THz link demonstrator of the first

generation. The detailed characterization of this link allowed the validation of existing signal

processing algorithms and the identification of the required improvements. The maximum

data rate for the current transceiver modules was determined, and for the first time an error

free 100 Gbit/s data transmission could be demonstrated using 16-QAM signals at a symbol

rate of 32 Gbd. The next steps will include modelling and design of an improved generation

of algorithms and developing new frontend hardware specifications.

Digital and analog phase shifting were evaluated in Section 6 by simulations. Since complex

hybrid analog-digital beamforming architectures cannot be implemented currently, the

focus is on small linear arrays with 4-8 elements. This work helped to narrow-down the

demonstrator hardware options and to identify the algorithms that will be explored in the

next phase of the project in more detail.

In the first phase of the project, the ongoing work covered all major objectives of WP5 as

planned. Among many options the most promising components and architectures were

identified for further implementation in the next project phase, which was actually the main

goal of this first deliverable.

Documents

Deliverable D5 - Terranova · D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation