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Digital control of chopper-fedDC motor drive

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• A Doctoral Thesis. Submitted in partial ful�llment of the requirementsfor the award of Doctor of Philosophy of Loughborough University.

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;

DIGITAL CONTROL OF CHOPPER-FED

DC MOTOR DRIVE

by

THAMIR FARAJ MURAD, BSc

A Doctoral thesis submitted in partial fulfilment of the requirements for the award of the degree of

Doctor of Philosophy of Loughborough University of Technology

-"."-

October 1985

Supervisor: Professor I R Smith BSc, PhD, DSc, CEng, FlEE

.Department of Electronic and Electrical Engineering

I!: by Thamir Faraj Murad 1985

.... ~

LOUgr:b

i

.. ACKNOWLEDGEMENTS

I am greatly indebted to the Iraqi leadership for the Scholarship opportunity and the invaluable support in many aspects which really

made this work· possible.

I would also like to take this opportunity to express my deepest grat itude to both Professor I R Smith and Mr Gordon KettleborOligh for their invaluable guidance, advice, encouragement and patience . throughout the research and the preparation of this thesis.

I am particularly grateful to Mr K Gregory with whom I often held useful discussions, and to all the good friends and colleagues in the Electronic and Electrical Engineering Department.

I would also like to thank Mrs Janet Smith who competently typed this thesis, and Mrs Janet Redman who patiently prepared the figures and the diagrams.

Finally, I wish to acknowledge the great understanding, endurance and moral support of my wife and family which helped a lot in com~

pleting this work.

i i

DIGITAL CONTROL OF CHOPPER-FED DC MOTOR DRIVE

Summary

This thesis is concerned with a variable-speed DC drive system, based on the armature control of a separately-excited DC motor driven by a four-quadrant chopper. The chopper circuit employs gate-turn-off thyristors, whose firing sequence is determined so as to provide the

• mode of operation required in a particular drive application.

Closed-loop control of the motor is implemented in a completely digi- . tal form. The control strategy is such that, when the speed error is greater than a certain value, the maximum acceleration and torque are both determined by a current limit function, restricting the armature current to a given predetermined value. When the speed error is redu-ced to below the limit fixed by this function, a control algorithm comes into operation so as to ensure that the required speed is attai-ned with .an .. op,l:imal step response from the control system.

Closed-loop control is achieved by means of only a single speed feed-back loop without the use of any AID or DIA converters, and the need for a separate current loop is obviated by the real-time control algo-rithms employed.

Acknowledgements Summary

Contents

List of Principal Symbols

CHAPTER 1: INTRODUCTION

DC DRIVES

i i i

CONTENTS

CHAPTER 2: 2.1 DC Motor Speed Control ...•

2.1.1 Field Current Control

....

2.2

2.1.2 Armature Resistance Control 2.1.3 Armature Voltage Control Electrical Braking

2.3 2.2.1 Regenerative Braking Solid-State Drives 2.3.1 AC to DC Converters

Page No

i i i

i i i vii

1

5

5

6

7

7

8

8

8

9

2.3.1.1 Phase control 10 2.3.1.2 Integral cycle control 10-

__ 2.3.2 DC to DC Converters 11

2.3.2.1 Inverter/rectifier scheme... 11 2.3.2.2 DC choppers 11

CHAPTER 3: DC CHOPPERS

3.1 Principle of Chopper Operation 3.2 Chopper Configurations 3.3· Control Schemes for 4-Quadrant Choppers 3.4 Switching Devices

CHAPTER 4: GATE-TURN-OFF THYRISTORS 4.1 GTO Structure

4.2 GTO Characteristics 4.3 GTO Applications

16

16

18

19

21

28

28

31 32

-------,

CHAPTER 5:

5.1

iv

POWER CIRCUIT DESIGN Device Rating

5.2 Protection

Page No

35 35 36 36 37 40 40 40

~ 5.2.1 Over-Current Protection '- 5.2.2 Snubber Circuit Design

5.3 GTO Gate Drive

~5.4

5.3.1 Gate Drive Requirements 5.3.2 Gate Drive Circuits

5.3.2.1 Un-isolated drive circuit .. 41 5.3.2.2 Isolated drive circuit

Power Supplies 5.4.1 High Voltage Power Supply

41 42 43

5.4.2 Main Control Circujt Power Supply... 43 5.4.3 Gate Drive Multiple-Output Power

Supply 43 5.5 Open-Loop Trigger Pulse Generating Circuit.. 44

. CHAPTER 6: DIGITAL CLOSED LOOP CONTROL SYSTEM 58

6.1 DC (~otor Transfer Function and Block Diagram 58 6.2 The PlO Controller 60

6.2.1 The Digital PID Controller 61 6.3 Current-Limit Control Algorithm 64 6. 4 [nc~em;n~~ Contro 1 A 1 gori thm 66

CHAPTER 7: SPEED MONITORING AND MEASUREMENT SYSTEM 74 7.1 Shaft Encoder Interfacing Circuit 75 7.2 Direction of Rotation Detection Circuit 77 7.3 Clock Circuit and Gating Signals Generation 77 7.4 Binary Counter Circuit 78 7.5 BCD Counter Circuit Display 79 7.6 Demand Speed Interfacing Circuits 79

CHAPTER 8: EXPERIMENTAL CLOSED-LOOP SYSTEM 92 8.1 Bi nary Arithmeti c 93

8.1.1 12-bit Adder/Subtractor Circuit 95

v

Page No

8.2 Speed Comparison Circuit 96 8.3 Control Algorithm-l (Current·Umit Controller

Circuit) .•.. 97 8.4 Control Algorithm-2: l~on-Linear Controller

Circuit ••.. 100

8.5 8.6

8.7

CHAPTER 9: 9.1

9.2

8.4.1 Additional Integrator in the' Incremental Controller ' 103

8.4.2 Starting with Control Algorithm-2 ... 104 Control Algorithm Decision Closed-Loop GTO Gate Drive Circuit

Circuit Pul se Generation

Timi ng and Synchroni sation Ci rClli t

MATHEMATICAL MODEL OF THE DC DRIVE Open-Loop Operation 9.1.1 Power Application 9.1.2 Free-Wheel i ng' .... 9.1.3 The' Regenerative Mode

105

105 106

122 123 124 124 125

9.1.4' Torque Balance Equation 125 9.1. 5 Open-Loop Computer' Impl ementat,ion ... 126

9.1.6 Open-Loop Simulation Closed-Loop Operation 9.2.1 Algorithm-l 9.2.2 Algorithm-2

9.2.2.1 PI controller 9.2.2.2 PID controller 9.2.2.3 Incremental controller 9.2.2.4 Incrementa 1 controller with

128 129 130 131 131 131 132

'integrator 132 9.2.3 Closed-Loop Simulation 133

9.2.3.1 PI controller as algorithm-2 133 9.2.3.2 PID controller as a1gorithm-2 134 9.2.3.3 ~ In~~~~e-;;t;'- controller as

, °a 1 gorJtnm-,2 . . . . 136 9.2.3.41 Incremental ,controller with

, integrator as a1gorithm-2 136

CHAPTER 10: 10.1

10.2

CHAPTER 11:

REFERENCES

APPENDICES:

vi

EXPERIMENTAL STUDIES .... Open-Loop Tests and Results 10.1.1 Open-Loop Circuit Waveforms 10.1.2 Drive System Response with Open-Loop

Control Closed-Loop Tests and Results 10.2.1 Closed-Loop Circuit Waveforms 10.2.2 Drive System Response with C1osed-

Page No

166

166 167

169 171

171

Loop Control 172

COMMENTS AND CONCLUSIONS 217

221

233

Appendix 1: GTO Thyristor Specification 233 Appendix 2: Heat Sink Design for a BTW5B GTO 244 Appen9.i x 3: Snubber-Ci rcui t Des ign 246 Appendix 4: Component Specifi cations 247 Appendix 5: Shaft Encoder Specification 249 Appendix 6: 4-Digit Counter Driver Specifications 251 Appendix 7: 8-Bit by 8-Bit Multiplier Specification 263 Appendix 8: DC Motor Drive System Parameters 274 Appendix 9: List of Symbols Used in the Simulation Program 275 Appendix 10: Runge-Kutta Method in Computer Simu1arion Form 277 Appendix 11: Listing of Subroutine 'COMPRS' Used for Arma-

ture Current Envelope 278 Appendix 12: Published Work 279

A

-- Cl +0

4

0 n

0 s

--. ~o e-a

eb ef E

E 0

f

fl+f 2

F

G-,H

Gl ,G2 i

0

i a

if

lA

I amax

vii

LIST-OF PRINCIPAL-SYMBOLS

Incrementa 1 speed encoder output constant-

Closed-loop control system constants

External timing capacitor for monostables

Snubber circuit capacitor

Free-wheeling diodes

Number of chopper period intervals representing the chopper output pulse width at sample n

Snubber circuit diode

Chopper output instantaneous voltage

Armature applied voltage

Armature generated back emf

Motor field applied voltage

DC power supply voltage

Chopper output mean voltage

Incremental encoder output frequency

Over-current protection fuses

Viscous friction coefficient of the motor and the load

Control system ga ins

Counter gating signals

Chopper output current

Armature current

Fi e 1 d current

GTO anode current

Maximum allowable armature current

viii

IG GTO gate current

IGT GTO gate triggeri ng current

IL GTO latching current

IC Integrated circuit component

INVa +INVf Shmttt-trigger inverters

J

K

K a

Kb

KD

KI

K P

Kch

Ll

La

Lf

LL

Ls

Ml+MS

Ma

Moment of inertia factor for the motor and the load

Constant

Torque constant of the motor

Back emf constant of the motor

Derivative controller constant

Integral controller constant

Proportional controller constant

Chopper constant

Current-limiting reactor self-inductance

Armature self-inductance

Field self-inductance

Load circuit self-inductance

Power supply self-inductance

Monostable multivibrators

Acceleratij'on torque

Electromagnetic torque

Load torque

Loss torque

n Number of sampling intervals

N Encode r rota t i ona 1 speed

Q Logic circuit output

s

S

sw t

T1+T4

T c

TR1+TR4

Ul+U

8

Vamax VT

X

Y

Cl

T a

III

ix

Armature circuit resistance

Field circuit resistance

GTO gate series resistance

Load res i stance

Snubber circuit resistor

External timing resistor for monostables

Lap1ace transform

Ideal semiconductor switch

Mechanical switch

Time

--ON timE!" -interval of the chopper

OFF time interval of-the chopper

Chopper period

Transformers

Chopper period interval

Transistors

Timing and synchronisation pulses designation

Maximum allowable armature applied voltage

Reference voltage

Motor rotational speed

Speed error limit in the closed-loop control system

Chopper duty cycle

Motor air gap fl ux

Electrical time constant of the motor and the load

Mechanical time constant of the motor and the load

Angular velocity

x

Wm Mechanical angular velocity

wr Reference angular velocity

I1V Appl ied voltage increment

I1w Speed error 1 imit

All other symbols are defined as they appear •

. . -

.:";

1

CHAPTER 1

INTRODUCTION

Variable-speed drives are required in many industrial applications, with DC rather than AC motors often being adopted because of their high efficiency over a wide range of conditions. A further advantage is that their speed may be changed using relatively simple techniques, such as armature voltage control, based on either AC/DC or DC/DC power converters.

Historically, the motor/generator set was used for power conversion, but the many disadvantages associated with its use resulted in a very restricted range of applications. The development of efficient, rugged and compact power thyristors, with short switchi ng times, has greatly increased the available ratings of power converters,as well as reducin~ their maintenance costs and increasing their reliability(l ,2).

A new device, termed the gate-tu rn-off thyristor (GTO) , has a major advantage over the conventional thyristor, in that it can be turned-off by the application of a negative signal to the gate, thereby eli-minating the need for a commutating circuit. This feature is particu-larly useful in DC/DC converters or chopper circuits fed from a DC supp ly.

Recent advances in both digital electronics and microelectronic tech-nology have led to their adoption in almost every field of power electrical engineering, and the digital control of variable-speed drives appears to be a particularly interesting application. One immediate advantage of using digital methods for DC motor speed con-trol is that the system accuracy is increased, because of the elimina-tion of the drawbacks inherent in analogue systems, such as component ageing and temperature drift. The signal may also be transmitted over a long distance without distortion, if remote control is required(3-5).

The role of microprocessors in control applications has usually been

that of a programmable logic controller. However, they then function

2

mainly as a computational element, performing calculations as defined by a control algorithm. In practice, microprocessors take a much longer time to execute calculations than do the discrete integrated circuit controllers, and in some applications, where· a very fast response is required, microprocessors are unsatisfactory. In addition, a relatively high level of both engineering training and technical understandi ng is requi red by the mai ntenance staff, and thi s .i s not always. available. It is primarily, for these reasons that the digital

, ,

speed control of a DC drive using disc~ete ,integrated circuit components is investigated in this thesis.

Ana 1 ogue contro T systems for 'DC dri ves are we 11 known, and many concepts found in these systems have to a,considerable extent been carried over into some of the digita'l control sch~mes reported in the literature(6) In general, a control scheme contains both an outer speed feedback loop and an inner current feedback loop .. :The speed information may be obtai-ned from a digital tachometer and applied directly to the digital con-troller, but the armature current information requires an analogue-to-digi.tal converter in the current loop, which increases the hardware complexity. Although several papers dealing with this problem have been published, these are concerned mainly with low-power servo applica-tions and do not consider certain technical requirements of industrial drives, such as a maximum current limit, controlled acceleration and deceleration, fast speed reversal, etc ..

This thesis describes a four-quadrant digitally-controlled DC-drive system, based on the armature voltage control of a separately-excited DC motor. The power circuit is a four-quadrant H-bridge chopper using GTO power switching devices. Motor speed control is achieved entirely by digital techniques speed feedback loop.

and is novel, since it uses only a.single outer The scheme described eliminates the need for the

analogue-to-digital conversion necessary in an inner current feedback loop.

3

Initially, the theoretical and practical open-loop behaviour of the drive is investigated, and the general problems inherent in the system are identified. One such problem is the large inrush of current when the motor is started from rest. Closed-loop operation is also investi-gated and different control strategies are used, including a recently developed non-linear control algorithm. The drive investigated is expected to be valuable in a wide range of industrial applications. For instance, in the printing and paper industries, precise motor speed control is necessary when a number of motors are either mechanically interlinked or distributed in a defined sequence along the paper web path and required to run at precisely the same speed. Another important area arises in military applications, where the accurate control of both speed and position is a prime requirement of the weapon systems in many fighting vehicles.

The equivalent circuit of the separately excited DC motor is presented in Chapter 2 of the thesis, with the corresponding performance equations for the motor being derived. It is established that armature voltage control provides the best technique by which the purposes of the pre-sent project can be illustrated. The chapter concludes with a discus-sion of the principles of operation of various types of solid-state power converters and considers the areas of application in DC motor -control to which they are more suited.

Chapter 3 details briefly the operating principles of various types of DC chopper circuits. A 4-quadrant chopper was chosen as the experimen-tal circuit and various control schemes for this particular configuration are discussed.

The structure, characteristics and applications of GTO thyristors are also outlined in Chapter 4. Since both the turn-on and the turn-off processes of a GTO differ from those of a conventional thyristor, these differences are discussed and analysed so as to lead to a better under-standing of the design of the gate drive circuit.

In Chapter 5, a detailed account is presented of the-power circuit design, in particular the chopper bridge and its auxiliary circuitry,

4

such as the snubber circuits needed for protection purposes and the power supply circuits which provide the gate drives to the GTOs. Several gate drive configurations are explained, with the discussion finally concentrating on an isolating arrangement as,the most suitable for use with the H-bridge chopper configuration.

In Chapter 6 several control algorithms are discussed, and the mathema-tical equations which represent their operation are derived. A detailed discussion concentrates on the digital version of the conventional PlO controller(7), the current limit algorithm, and th~ incremental control algorithm(8). The last two are both combined and i'mplemented in the experimental closed-loop control system.

Chapter 7 describes circuits for the measurement of motor speed and explains the interfacing arrangements required between the speed demand as set by the system operator and the control system.

Chapter 8 discusses the practical implementation of the control algo-• rithms using discrete integrated circuit components, with the circuit

interfacing the control system to the chopper drive circuits also being discussed. Some attention is devoted to the important functions of timing and synchronisation in the control system operation.

In Chapter 9, mathematical models are derived for both open-loop and closed-loop operation of the system, and these are validated by exten-sive comparisons between the theoretical and the'experimental perfor-mance recorded in Chapter 10. A wide range of simulations was perfor-med using the PlO control algorithm,since this has not been implemented practically in the ,project, and the results are compared with the non-

" control algorithm whi ch was implemented practi ca lly.

Finally, Chapter 10 investigates the transient response of the controlled motor under different load conditions, and demonstrates the effect on the response of changing various significant system control constants.

5

CHAPTER 2

DC DRIVES

Although DC motors have high capital and maintenance costs when com-pared with induction motors, their use in industrial applications has increased as a result of the ease by which their speed may be adjusted and of their superior speed-torque characteristics. Speed variation may be achieved by resistance inserted in series with the armature or the field, or by varying the DC voltage applied to the armature. In the latter case, the variable supply may be provided by a DC generator (Ward-Leonard set), a conventional rectifier circuit, a power amplifier, or a thyristor converter system.

With the advent of thyristors, AC/DC power conversion units have dras-tically decreased in size. They are now less costly than motor-genera-tor sets of the same rating(2,lO), less time and cost. is .required for their maintenance and they are more reliable. The extensive use of integra-ted circuits in control systems (1 1 ,12) has further enhanced these advantages.

2.1 DC Motor Speed Control

Variable-sp£ed drives and servo-systems often involve separately-excited DC machines, due to their ease of speed adjustment, and the analysis in this section therefore concentrates onthis situation.

Figure 2.1 shows the equivalent circuit of a separately-excited DC motor, for which there are six·terminal variables; the field voltage and current (ef and if)' the armature voltage and current (ea and i a), and the mechanical angular velocity and torque at the shaft (wm and ML) .. The equations relating these variables are:

Rf if + di f (2.1 ) ef = Lf (It

K dia

ia (2.2) ea = "'m + La dt + Ra

6

and (2.3)

If any of the quantities if' ia' or wm remains constant, the correspon-ding term in equations (2.1)to (2.3) containing its time derivative becomes zero. The generated armature voltage eb is directly proportio-nal to the angular velocity wm' or

During steady-state operation, eb is less by an amount equal to the voltage drop in From equation 2.2, the steady-state speed

(2.4)

than the terminal voltage ea the armature resistance i R

a a of the motor is therefore:

(2.5)

The relationships contained in equation 2.5 show that the steady-state motor speed may be varied by adjustment of either:

a) Flux (using field current control) b) Armature circuit resistance, or c) Armature terminal voltage.

2.1.1 Field Current Control

With field current control, the lowest motor speed obtainable corres-ponds to the maximum allowable field current, with the 'highest speed being restricted by armature reaction effects which eventually cause either the speed to become unstable or the commutation to become poor. This type of control is referred to as a constant power drive, since

7

the maximum output power available remains constant over the entire speed range, whereas the torque varies directly with the motor flux.

2.1.2 Armature Resistance Control

Armature resistance control is achieved by inserting external resistance in series with the armature. However, for a fixed value of resistance, the speed varies widely with load, while the power loss in the resistor is large' especially when the speed is low. Armature resistance control provides a constant torque drive, because both the flux and the maximum armature current remain constant as the speed changes.

2.1.3 Armature Voltage Control

Armature voltage control utilizes the fact that a change in armature voltage is accompanied by an almost proportionate change in both the generated emf and the motor speed. Traditionally, a motor/generator set has"been used to provide a controlled armature voltage, but with the development of high-power solid-state devices the range of possibi-lities for the precise control of motor speed using these devices has increased considerably.

Control by reduction of ' the armature voltage provides speeds below the rated or base value, while for speeds above the base value field control must be adopted. The direction of rotation of the motor depends on the polarity of the main magnetic field and the di~ection of the armature

/ current, so that a reversal of rotation is/achieved when the polarity of either of these quantities is _changed.

8

2.2 Electrical Braking

Electrical braking based on either dynamic or regenerative techniques is provided in many controlled DC drives, to enable the kinetic energy of the rotating system to be converted to electrical energy. In systems using dynamic braking the energy is dissipated in a resistor stack, whereas in systems using regenerative braking it is returned to the supply thereby improving the drive efficiency.

2.2.1 Regenerative Braking

Regenerative braking is conceptually simple and, in the case of a sepa-rately-excited DC motor, it may be achieved with no circuit changes. Any increase in speed due to load rejection causes the back emf to

. exceed the 1 i ne voltage, whi ch results in a reversal of the armature current and power being returned to the supply. Also, if for any rea-son the supply voltage becomes less than the armature emf, regeneration will again take place. However, the scheme is not normally self-stopping, and to bring the drive to rest requires either plugging or the use of a mechanical brake. At high speed, the braking effect produced by regen-eration is far less than that provided by dynamic braking, but it is widely used because of its efficiency and simplicity, especially in transit vehicles and battery-operated electric cars.

2.3 Solid-State Drives

Historically, variable armature voltage control was achieved using a Ward-Leonard arrangement(6), but.with the advent of modern electronic components an increased reliability and a better peformance have both been achieved using electronic drives. The advantages of a power electronic solid-state drive are that:

a) It eliminates the field and armature time constants inherent in the Ward-Leonard set

9

b) It is simple.and reliable . . c) Minimal maintenance is required d) The efficiency is high e) The small size and weight together result in reduced space

requirements and lower cost.

~ The disadvantages are:

a) The ripple content of the converter output is high b) No power factor improvement is achieved c) It has a comparatively low overload capability

d) The AC supply voltage may be distorted and telephone interference produced.

The large extent to which solid-state converters are now used shows that these disadvantages are not generally too serious or they can be readily overcome. '>

Three basic methods are available for the provision·of·a variable DC voltage from a fixed AC or DC supply voltage, these being phase control, integral cycle control, and chopper control. The first two methods achieve conversion from AC-to-DC by rectification, whereas the final method involves DC-to-DC conversion.

2.3.1 AC to DC Converters

In both phase and integral cycle control schemes, the·AC supply voltage

turns off the semiconductor switch in the converter. No commutation circuits are therefore necessary and the schemes are simple and inex-pensive. Phase control is widely used, since it can control the output voltage smoothly over a wide range, but it has the disadvantage that the supply power factor decreases .at.lower output voltages. Integral cycle control is satisfactory if the supply frequency is high, otherwise the motor may oscillate about a mean speed.

)

10

2.3.1.1 Phase control

Phase control connects the motor to the converter for only a portion of each half-cycle of the AC supply, to provide the armature voltage waveform shown in Figure 2.2(a). Commutation occurs naturally, since the incoming thyristor reverse-biases the outgoing thyristor and turns it off. No additional commutation circuitry is therefore required.

Phase-controlled converters may be broadly classified as single-phase converters (Figure 2.3) or 3-phase converters (Figure 2.4). Semi-converters are one-quadrant arrangements in that there is only one polarity of voltage and current possible at the DC terminals. Full converters, on the other hand, are two-quadrant arrangements, in that although the voltage polarity· can reverse the current remains always unidirectionaL Dua lor inverse-parallel converters can operate in all four quadrants.

In many single-phase converters the motor armature current may become discontinuous, causing a deterioration in the drive performance. In 3-phase converters the motor current is usually continuous, and the higher ripple frequency at the motor terminal results in less onerous filtering requirements.

The 3-phase half-wave converter is impractical for most purposes, because the supply current contains a DC component. Semi-converters and full-converters are most commonly used in practice, with dual converters being employed for reversible drives having power ratings of up to several megawatts.

2.3.1.2 Integral cycle control

With integral cycle control the converter connects the motor to the supply for a discrete number of complete half cycles, to provide an armature voltage waveform as shown in Figure 2.2(b).

11

2.3.2 DC to DC Converters

The two main methods of DC to DC conversion using solid-state swit-ching devices are by an inverter/rectifier combination or by a DC chopper.

2.3.2.1 Inverter/rectifier scheme

In this technique, the DC is first converted to AC, which is then stepped up or down by a transformer before being rectified back to DC. Because the conversion isin two stages, the scheme is costly, bulky and less efficient than the DC chopper considered below, but the iso-lation provided by the transformer between the load and the supply is a useful and widely required feature.

2.3.2.2 DC choppers

As illustrated in Figure 2.2(c) and (d), this form of control converts directly from DC to DC. It may be visualised as the DC equivalent of a variable-ratio transformer. The chopper can replace the resistor sometimes used in series with the DC motor armature to provide speed control so that it can be used in battery-operated vehicles where energy saving is a prime consideration.

Since choppers can provide regenerative braking, they are commonly used in transportation systems where frequent stops.occur. Choppers· are also used in other applications, such as trolley cars, marine hoists, fork-lift trucks and mine-haulers.

In the experimental work of this thesis, the DC chopper was adopted as the power converter for the DC motor drive, and it is therefore discu-sed in more detail in the following chapter. The power switching devi-ces used in the practical circuit were the recently developed gate-turn-

off thyristors (GTOs) and an in-depth discussion of these is given in Chapter 4.

12

MECHANICAL LOAD

Fig 2.1 EQUIVALENT CIRCUIT OF A SEPARATELY -EXCITED DC MOTOR

INPUT VOLTAGE

13

PHASE (ONTROL

MA tU \7'7]

alPhase control L..-___ --'

OUTPUT VOL T AGE

Ac 1\ 1\ 1\ 1\ 1 INTEGRAL 1 V \TV v. ~6~~~OL r, -----'-----bllntegral- cycle control L...-___ ....J A---::A----1..Eo

I

Dc :t::E:I (HOPPER (ONTROL

to' t",

-Cl D( (hopper IT ~ d"1- ---LEo . t

M f\ t1 :~~TlFlER 11-1-1 D_C ------l VI FILTER _ CHOPPER I t"'ttJffE, dl Rectifier-chopper arrangement

Fig 2,2 (ONTROL TE(HNIQUES FOR OBTAINING A VARIABLE D(-VOL T AGE

14

+ a) Half-wave

b) Semi-converter +

. -

cl Full-converter

+

d) DUill-converter

Fig 2.3 SINGLE-PHASE PHASE-CONTROLLED CONVERTER CIRCUITS

A B C

N

A B C

A_ .... B --4--'" C

A --4 B C

15

Fig 2.4 THREE PHASE

la

e' a a) Half-wave

la

+

ea b) Semi-converter

ea c) Full-converter

d) Dual-converter

~~~_C ~-+--B

A

PHASE-CONTROLLED CONVERTER CIRCUITS

16

CHAPTER 3

DC CHOPPERS

DC chopper circuits were developed for controlling DC motors supplied from fixed voltage sources, such as the battery of an electrical vehicle or the third rail or overhead catenary wire of a rapid tran-sit car. Chopper circuits eliminate the need for added resistance in the armature circuit, and have the advantage of higher efficiency, continuous control and regenerative operation without any additional circuit components.

In this chapter, the principles of chopper operation are explained, and circuit configurations for different-applications are outlined·. The discussion concentrates mainly on 4-quadrant choppers and their associated control strategy, as'implemented in the circuit used in the experimental investigation.

3.1. Principle of Chopper Operation

A basic chopper circuit is shown in Figure 3.l(a), where the switch S repeatedly connects and disconnects the load from the DC supply to produce the chopped load voltage shown in Figure 3.1(b).

During one cycle of operation, the supply is connected to the load for a time ton and disconnected for a time toff' During the latter time, current flows around the loop formed by the load and the free-wheeling diode D, to give the decay of load current waveform evident in Figure 3.l(c). The time constant of the exponential current build-up is dependent on the combined load.and supply inductance, whilst that of the exponential decay is due to the load inductance only. The mean value of the DC load voltage is

17

or

t E on

I

where T = ton + toff is the chopping period

and t on . a = I 1S the duty cycle.

(3.1 )

Clearly, changing the chopper duty cycle changes the load voltage and this may be achieved in one of two ways.

a) The chopping frequency f = t, and if this is .kept constant while the on-time.ton"is increased, the mean output voltage is also increased, as shown in Figure 3.2(a). This technique is known as constant frequency operation or pulse width modulation

b) The chopping period is varied while either the on-time or the off-time is kept constant. This technique is known as variable frequency operation or frequency modulation.

Figures 3.2(b) and (c) shovl how either approach results in control of the mean output voltage. The former method is generally preferred, since variation of the off-time has the following significant disad-

vantages:

i) To provide a wide range of output .vo1tage requires the frequency to be varied over a wide range

ii) Filter design for variable frequency operation is difficult

iii) Interference with signalling and telephone lines is more pron-ounced

18

iv) The large off-time for low output voltages may cause discontinuous current, especially in small DC motor 'loads.

An efficient and economic fast-response chopper drive system obviously requires a high operating frequency with minimum chopper losses, a low harmonic content in the output voltage and the input current, and small size input and output filters.

3.2 Chopper Configurations

Various chopper configurations are shown in Figures 3.3, 3.4 and 3.5. Figure 3.3(a) shows a first-quadrant chopper, in which the armature

terminal voltage eo = E when the switch 51 is conducting and eo = 0 when the switch is non-conducting, due to the presence of the free-wheeling diode D. With positive load voltage Eo and current 10 defined as in Figure 3.3(a), power flows from the SOUI"Ce to the load such that the machine is motoring.

Figure 3.3(b) shows a second~quadrant or regenerative chopper, in which

eo = 0 when switch 52 is conducting and eo = E when it is non-conducting. The direction of the supply current is reversed from that of the first-quadrant chopper, and the machine is regeneratively braking by feeding power back to the supply.

The previous two circuits may be combined as in Figure 3.4(a), to give a 2-quadrant type A chopper, where eo = 0 if either switch, 52 or diode D2 is conducting, whereas eo = E when either 51 or Dl conducts. How-ever, since the armature current 10 is positive if 51 conducts but negative when 52 conducts, whereas Eo is always in the same direction, the power flow is reversible. This chopper configuration may there-fore be used for both motoring and regenerative braking of the machine.

Figure 3.4(b) shows a type B, 2-quadrant chopper, in which eo = +E if both 51 and 52 are conducting and eo = -E if both 51 and 52 are

19

non-conducting but 01 and O2 are conducting. The armature voltage Eo

is either positive or negative, depending on whether the conducting time of the thyristors is greater than or less than their non-conduc-

ting time. Since 10 is always positive, whereas Eo is reversible,

power flow is also reversible and this configuration may also be used for both motoring and regenerative braking of the machine. Figure 3.5

shows a 4-quadrant chopper, with the facility for reversing both Eo

and 10

. With both 51 and 54 conducting and 52 and 53 non-conducting, both Eo and 10 are positive, giving first-quadrant operation. When

52 and 53 are conducting and 51 and 54 are non-conducting, both Eo and 10 are negative, giving third-quadrant operation. When 01 and

04 are conducting Eo is positive and 10 is negative, giving second quadrant operation, and when O2 and 03 are conducting Eo is negative

and 10 is positive, resulting in fourth-quadrant-operation. Thus, the 4-quadrant chopper provides a reversible, regenerative drive and for this reason it was used for the present project. Extended details of chopper circuits analysi s may be found in the 1 iterature(20-24).

3.3 Control Schemes for 4-Quadrant Choppers

The various control strategies for a 4-quadrant chopper may be explai-ned by reference to the arrangement shown_ in Figure 3.5 which possesses the following idealised properties:

a) - Switches 51-54 and diodes 01-04 have zero impedance when conduc-ting and infinite impedance when non-conducting

b) Only two of the four circuit switches may be closed at any instant. The instantaneous output voltage thus depends on the state of the switches and is independent of the load current

c) The load inductance is sufficiently large to produce a smooth

load current.

20

In the simplest or bi-polar method of chopper control (26), switches

51 and 54 are closed for a time ton during which the output is +E, and switches 52 and 53 are closed for the remainder of the cycle during which the output voltage is -E. With bipolar control, change-over of the output voltage from positive to negative or vice versa is smooth. However the input current and the output voltage ripples are large, especially at low output voltage, and four commutations occur per chopper cycle.

Unipolar control is an alternative technique which involves only two commutations per cycle. For positive. output voltage 53 is permanently open and 54 is permanently closed.

51 is closed for a time ton to apply a voltage +E to the load, whereas for the rest of the· cycle 51 is opened and 52 is closed. For a posi-tive average output voltage, the instantaneous terminal voltage is therefore:

e = E o

e = 0 o

o < t < ton (51 and 54 closed)

ton < t < T

For a negative output. voltage 53 is permanently closed and 54 is per-manently open, with 51 and 52 being switched alternately in a manner similar to that described above. With this technique both the output voltage ripple and the input current ripple are low, especially at low output voltage. The unipolar control strategy may be modified in accordance with the application, or to enable microprocessor imp le-mentation(27) .

An important modification is adopted for the present work. A positive output voltage is obtained using the conventional unipolar strategy described above, but for a negative output voltage 52 is permanently

closed with 51 permanently open, while 53 and 54 switch in a comple-

21

mentary manner. Taking the positive voltage situation as an example, it is thus evident that the state of the switches on one CHOPPER SIDE (53 and 54) remains unchanged, whi.le those in the other CHOPPER SIDE (51 and 52) are continuously changing state in a complementary manner. For a negative output voltage, the .two CHOPPER SIDES interchange their switching schemes, with the pair on the right-hand side (53 and 54) switching in a complementary manner, and the pair on the left-hand

side (51 and 52) remaining always in the same state.

This modification was developed to achieve a simple hardware implemen-tation, which could be interfaced in an easy and reliable way to meet the control system circuit requirements as will become evident in Chapter 5.

3.4 Switching Devices

The three main types of switching device currently available for high power applications are power transistors, conventional thyristors and GTOs. Transistor power ratings are presently quite limited, and although lOOOV, lOOOA units are available, these are fragile and in-capable of withstanding voltage and current overloads. In addition, the number required in a series/parallel connection to satisfy high voltage or high current requirements both increases the complexity and reduces the reliability of the ·equipment. Although thyristors are fast switching devices and have excellent power handling capabi-lities, the accompanying commutation circuitry introduces undesirable re 1 i abil i ty problems. The GTO is a pn pn devi ce whi ch can be tuned either on or off by an appropriate control signal applied to the gate-cathode junction(29), and this feature is important, since it elimi-nates the additional and complex commutation circuits necessary when conventional .thyristors are employed in applications such as choppers. Taking all factors· into account, the GTO now appears to be the most attractive switching device for many high power applications and GTOs were therefore employed in the practical power circuit described in

22

this thesis. Chapter 4 contains a more detailed discussion of this device and its use in many applications.

23

T~ LL 10 . I XliX) 1 ~v

E D .., ~ ,., LOAD

-- I a) Basic chopper circuit

-E o r-~~L-L-____ L-L-____ ~~ __ ~t

T

b) Output chopped voltage

F-------~------~L-----~~-----I~n

r-------------_______________ t

c) Load current

Fig 3. 1 CHOPPER CIRCUIT CONFIGURATION AND OPERATION

24

.. ~ ~ ~ ~ ~ E -to 1X=25%

"1 T_

I I I I I I I I E 0

1X=75% - t al Constant-frequency system

"1- ,",_ I I I ~ 1X=25%

"[''''- - t I I I I I I1 Eo I 1X=75% .. t bl Constant tON variable -frequency system

e01

t-=t=OF=F=-jDtj======fD=t====jDtjt====jD==f===.: ~o 1X=25% l! .. d I I 1X=75% .. t

cl Constant tOFF' variable-frequency system

Fig 3. 2 OUTPUT VOLTAGE WAVEFORMS FOR BASIC CHOPPER

25

+ Sl 10 EO 0 11""""':

01 10 E eo Eo

a) First-quadrant chopper

+

E

b) Second-quadrant or regenerative chopper

Fig 3. 3 ONE-QUADRANT [HOPPER [ONFIGURA TIONS

26

+

5, 0, Eo

E

I eo 5 10 O2 a) Two-quadrant type" A" chopper

+

Eo O2

E 10

0, eo 52

b) Two-quadrant type "B" chopper

Fig 3.4 TWO-QUADRANT (HOPPER (ONFIGURA TlON5

27

\

Fig 3. 5 FOUR-QUADRANT CHOPPER [ONFIGURA nON

28

CHAPTER 4

GATE-TU RN-OFF THYRISTORS

The GTO combines the advantages of the thyristor with those of the high voltage switching transistor. It is a fast 4-layer, 3-terminal device having a construction similar to that of the conven-tional thyristor. Like the thyristor, the GTO can be turned on by positive gate drives, but unlike the thyristor it can also be turned off by negative gate-drives. In addition, it incorporates the high blocking voltage and high over-current capabilities of the thyristor, and the fast switching and gate drive simplicity associated with transistors. The GTO also has the capability to withstand high rates-of-rise of anode current and voltage, and operates with low gate currents (30) .

The duration of the negative gate current withdrawn from a GTO needs to be no longer than a few microseconds. Since, in a conventional thyristor, commutation current flows for several tens of microseconds, the energy required to turn off a GTO is clearly much less than that required for a thyristor.

This chapter describes the structure and characteristics of the GIO, and outlines some of its important application areas.

4.1 GTO Structure

A conventional thyristor comprises the four layers shown in Figure 4.l(a) and indicated by PE' NB' PB and NE· Its operation is often described using the 2-transistor analogy of Figure 4.l(b), where the base terminal of a PNP transistor is connected to the collector of an npn transistor, and vice versa, and positive feedback between the

two transistors keeps the ~n-state current flowing. The self-hold function of the thyristor depends on this technique, and once a

29

thyristor is turned on, it cannot turn off by itself because of the high internal loop gain of the transistor pair. If the regenerative loop gain can be carefully controlled and a sufficiently l?w internal gate series resistance can be obtained, it becomes possible to con-struct a 4-layer device which can be turned off from the gate (i.e. GTO). This method of operation can be achieved as a result of recent advan-ces in semiconductor device-fabrication technology, including!~38)'

a) Careful control of the N- and P-base widths and resistivities, together with the use of new doping techniques such as ion-impl ccntation, which ensures that the internal gains are well-defined

b) A fine-geometry interdigitated finger structure for the gate-cathode junction, so that the whole of the active area of the device can be turned off simultaneously. This arrangement is shown in Figure 4.2(a)

c) The reduction in gain of the PNP transistor structure, by care-fully placed short-circuiting spots or bars between the P-layer emi tter and the N-l ayer base

d) Precise control of carrier life time by killing with gold or

other metals.

The equivalent circuit of the resulting structure is 'shown in Figure 4.2(c), where the short-circuiting spots have a very small resistance R. Resistor R represents the residual gate series resistance. The a g

shorted portions are located on the P-layer emitter opposite the area adjacent to the central portion of the N-layer emitter. This construction effectively suppresses the injection of holes from the P-layer emitter, and reduces the PNP transistor gain. In the anode shorted-emitter construction, since the P-layer emitter junction is

short-circuited, the device loses its reverse voltage blocking capa-bility. However, it achieves a lower on-state voltage, and a reduced

30

leakage current, and with its behaviour becoming almost temperature .- independent, this form of construction is effective for higher voltage

devices because of its lower on-state voltage(37). Turn-off of a GTO

is achieved by withdrawing sufficient current from the gate connection (via R ) to reduce the internal gain of the transistor pair to less than u~ity(38)

Consideration of the consequences of the internal operation and struc-ture of the device on circuit design, enables the following conclusions

'(38) to be drawn:. ,_

a) The interdigitated gate/cathode structure results in a very good turn-on performance. The whole of the active area is turned on simultaneously, which results in a rapid rate-of-change of anode current

b) The low internal regenerative gain results in an on-state voltage drop higher than in a conventional thyristor ;.. ..

c) There is a maximum current which can be drawn from the gate during turn-off, determined by the gate/cathode reverse breakdown voltage and the effective internal gate resistance

d) The anode current path through the device is squeezed ·down to a thin filament under each cathode finger during turn-off, until only one such filament is left conducting. External circuit precautions must be taken to prevent the rising anode/cathode voltage from causing excessive dissipation in this filament

e) When the flow of cathode current has ceased, current flow still continues for a short time between anode and gate, very much like the reverse-recovery transient current in a junction diode.

The charge which flows can be controlled by the level of heavy metal (gold) diffusion, but this is at the expense of the on-state voltage drop

31

f) Due to internal regeneration within the device, it can with-

stand high surge current pu1 ses (usually measured in terms of I2 t), which means that it can conveniently be protected by a fast semiconductor fuse.

4.2 GTO Characteristics

The forward characteristics of a GTO depend mainly on two parameters; the gate current required to achieve triggering IGT and the latching current IL.

When the anode current is less than IL the GTO behaves like a high voltage transistor, with an amplification factor IA/IG. If the gate current is less than IGT the device is in the off-state, with only a low leakage current flowing from anode to cathode. If the gate current becomes equal to or exceeds potential difference then

IGT the GTO turns on, with only a small present between the anode and cathode.

The GTO will revert to the off-state if the gate current falls below IGT , whilst the anode current remains below the latching level IL. If, however, the anode current exceeds IL, the GTO behaves like a thyristor and will remain latched in the on-state, even when -gate current ceases to flow. Unlike the thyristor, the GTO can be turned-off again by reversing the polarity of the gate drive vo1tage(39). For successful turn off, the required level of the negative gate

current must be reached before the expiry of the GTO storage time. This time delay can be as short as 500 ns, whereas the required gate current can be a few amperes, as it has to be about 1/5 to 1/3 of the anode current.

The GTO is incapable of blocking a high reverse voltage and, although this does not present problems for DC switching, a diode must be

connected in series with the GTO if reverse voltage blocking is requi-red for AC switching. During fast s~litching it is essential to minimise

32

the turn-on time, so as to achieve minimum device loss. With increased forward gate current, both the delay time and the rise time of the turn-on period can be reduced.

4.3 GTO Applications

GTO applications can be classified into the two basic categories of low frequency and high frequency circuits.

Low frequency switching circuits, up to frequencies of several kHz, include applications such as DC motor speed control by chopping a DC supply voltage, and AC motor speed control using pulse-width-modulated inverters.

High-frequency circuits, with switching frequencies of tens of kHz, include switched-mode power supplies, TV line detection and other newer power supply techniques such as those found in series-resonant circuit switched-mode power supplies~(50)

33

Anode

PE

ga !e NB

PB

NE

4) cath ode

(a) structure representation

Anode

PNP transistor

gate NPN transistor

cathode

(b) equivalent circuit

Fig 4.1 CONVENTIONAL THYRISTOR

34

gate elements cathode fingers

anode -'--__ N PNPNPN

(a) structure (layers)

Anode

P E PE

Ne Ga te

Pe

NE

Cath ode

(b) structure (representation)

PNP transistor

R

Anode R a

gate o--{::::r-~---:--f--1

(c) equivalent circuit

Fig 4.2 GATE-TURN-OFF THYRISTOR

NPN transistor

cathode

35

CHAPTER 5

POWER CIRCUIT DESIGN

This chapter discusses various design aspects of the experimental chopper drive. Figure 5.1 shows a block diagram of the drive, with the 4-quadrant bridge containing GTOs as the power switching devi-ces. Four isolated gate drive circuits are required to provide suitable firing pulses to turn-on and turn-off the GTOs, in accor-dance with the sequence defined by the modified unipolar control scheme discussed in Section 3;3.

Protection techniques for GTO devices, including both over-c.urrent and snubber circuit designs are also covered in this chapter, due to their importance both· in the overall system design and in the protection of the switching devices.

Figure 5.2 shows the arrangement of the experimental chopper bridge and its relevant auxiliary circuits, including the GTO gate drive circuits, overcurrent protection and snubber circuits. The necessary isolation of the gate drive circuits is achieved using a carefully designed isolated power supply. Multiple supplies are required for the gate drive circuits adopted, and these power supply circuits are explained in Section 5.4.

Open-loop operation of the system is also ·considered in this chapter, with the design of the open-loop trigger pulse generation being given in Section 5.5.

5.1 Device Rating

The rating of the GTOs depends on both the power supply voltage and the expected motor armature current. With the power supply of 240V,

36

and the motor armature rated at 240V, 2A DC, the GTOs must be capable of handling 240V, 2A with a reasonable overload capability. It was decided that the Mu11ard GTO's type - BTW58-1000R devices specified in Appendix 1 would be suitable, since these are rated at 6A anode current and 1000V anode/cathode voltage. The free-wheeling diodes D1-D4 must be similarly rated, and Mu11ard BYW19 fast reco-very diodes were found to be suitable.

The heat sink design for the GTOs is also an important part of the power circuit design, and is included in Appendix 2. In general, the power circuit employed can provide up to 6 kW load, and this can easily be extended to 10 kW if BTV58-1000R devices are used, with their capability of handling up to lOA anode current.

5.2 Protection

In common with other semiconductor devices, GTOs are susceptible to damage by excessive voltage or current, and in'genera1 the protection

.. . requirements of conventional thyristors apply also to GTOs. The gate-turn-off capability offers however a different method of overcurrent protection, although at the same time presenting another possible source of failure as discussed below.

5.2.1 Overcurrent Protection

There are two basic methods of overcurrent protection for a GTO. The first of these comprises a series current-limiting fuse, together with a parallel crowbar thyristor (42). When an overcurrent is detected,

the control device turn-off signal is inhibited and the crowbar thyristor is fired. The fault is subsequently cleared by the fuse within the non-repetitive surge current rating of the device and the crowbar thyristor. The device is clearly required to have a high over-

current rating and the protection cannot be reset.

37

The second method of protection comprises a current-limiting reactor to slow down the rate-of-rise of fault current. When an overcurrent is detected, device turn-off is activated with minimum delay~

Overcurrent turn-off gate pulses may be generated either by the normal gate drive circuit, or by a separate stand-by high current auxiliary circuit with a high rate-of-rise of current to decrease the turn-off time(43). The fast-fuse technique is commonly used,

but it suffers from the disadvantage that the fuses have to be replaced. Furthermore, it is becoming increasingly difficult to obtain fuses able to operate at the speeds required to protect the new and increa-singly fast devices. The crowbar provides overvoltage protection, but it is slow both in operation and in resetting.

The current-limiting reactor self-protect system allows resetting following a fault, and is particularly appropriate when very short transient conditions are likely to be experienced. For the experi-mental work described in this thesis, a current-limiting reactor was used, with an additional pair of fast semiconductor fuses, fl

and f 2' connected. in series with the chopper bri dge_ as shown in Figure 5.2. The current-limiting reactor l.l is by-passed by the diode DL during regenerative operation.

5.2.2· Snubber Circuit· Design

It is generally necessary to connect a snubber circuit across a semi-conductor switch, both to absorb the energy associated with the recovery current in the device and to limit the magnitude of the resulting voltage spike across the device. Although many studies have been undertaken on snubber circuits for conventional thyristors(44,45)

it is difficult to apply these directly to the GTO situation because:

38

a) The voltage polarities across the device during the turn-off

interval are inverted, because of the different turn-off mechanism

b) A GTO snubber circuit also provides protection from failure, due to current crowding, during the GTO conducting interval (46).

In addition, a snubber circuit improves the device turn-off swit-ching capability, whilst at the same time reducing the switching losses. However, its use produces higher voltage and/or current stresses in the device. Snubber circuits for GTOs have been descri-bed and analysed by Steigerwald (47). He discussed in detail the voltage spike and.~~e rate-of-rise of anode voltage during the conduc-tion interval, and related the energy dissipation in the snubber cir-cuit to the rate-of-change of anode current and the voltage rating of the device. Device characteristics and circuit parameter conside-rations were included in a subsequent model proposed by Ohashi (48), who concluded that connecting a snubber circuit across a GTO is a matter of considerable importanc.e. To illustrate this, Figure 5.3 shows a snubber circuit consisting of a diode, a capacitor and a resistor. Resistor Rs limits the peak discharge current of capaci-tor Cs when the GTO is turned on. The effect of diode Ds is to provide a low resistance path for the anode current during the turn-off time, during which capacitor C is charged to a voltage equal to s . -that across the GTO. The rate-of-decrease of anode current during turn-off may be sufficiently high to produce a large voltage spike across the device, due to the stray inductance of the snubber cir-cuit. For this reason, to be effective, the snubber circuit must be physically as close as possible to the GTO. This voltage spike increases the turn-off loss and may possibly result in a breakdown of the GTO, although the turn-off loss can be minimised by using a fast-turn-on diode with a low forward recovery voltage, a low induc-

tance capacitor and short lead lengths. The anode voltage of the GTO rises at a rate equal to IT/Cs' where IT is the anode current

39

being turned-off. The energy stored in C , when the GTO turns off, " s is di ssi pa ted in Rs during the next on-peri od of the GTO.

The capacitance of Cs should therefore generally not significantly exceed the minimum required, since if this is so the energy loss in Rs may be excessive. A further point to note is that Cs may not fully discharge during the on-time, so that when the GTO is turned-

off the anode voltage will rise rapidly to whatever voltage remains on Cs. This may cause excessive power dissipation in the GTO, and possibly lead to re-triggering of the device.

The snubber circuit shown in Figure 5.3 is suitable only for single-ended switching circuits (i.e. those using only one power switch). It is ineffective with bridge or push/pull circuits, as each switch

receives a large turn-on current pulse from the sudden increase in charge of the capacitor associated with the other switch (49), as is

the case in the exper.imental chopper circuit of Figure 5.2. The snubber circuit consists therefore of only a capacitor, connected

between the anode and cathode of each GTO, as shown in Figure 5.4. The GTO can easily withstand the anode current pulse caused by the capacitor, due to its high surge current and dI/dt ratings.

A limit to the value of snubber capacitance is defined by the peak

discharge current, which must not exceed the maximum controllable anode current rating of the "GTO. If a larger capacitance is employed, it becomes necessary to include also an inductance in the circuit, so

as to limit any current surge.

In Appendix 3, various methods of calculating the snubber circuit component values is included for the single-ended, as well as the

bridge switching circuits. An experimental snubber capacitor of 1 nF

was accordingly chosen to be connected across each GTO in the chopper

,power ci rcuit.

/ I

\

..------

40

5.3 GTO Gate Drive

A GTO will perform only as well as its gate drive permits. The

development of a suitable gate drive circuit is therefore extremely important and a careful study of the gate requirements will result in the attainment of the best device performance.

5.3.1 Gate Drive Requirements

The gate current required to turn-on a -GTO is determined by both

the turn-on time and the rate-of-rise of anode current, and the higher the gate current the shorter is the turn-on delay time.

The leading edge of the applied gate current should rise sharply to a relatively high level, which must be maintained sufficiently

_ long. for the anode voltage to fall to its steady-state level. There-after the gate current should be maintained at a lower level through-out the conduction period, so as to decrease the on-state losses.

To turn-off a GTO, a gate current of about 1/5 to 1/3 of the anode - ~-~- -

current has to be withdrawn very rapidly from the gate, in fact --- -... ---- -

with~n the device storage time. -This is achieved using a negative .-. -.. - ,-

gate drive voltage less than the gate/cathode reverse breakdown voltage, and it is usually switched from a small series capacitor charged during the on-time.

Gate turn-off should not be attempted too soon after turn on, since

sufficient time must be allowed for the snubber circuit to discharge. Failure to do this may produce a voltage step at turn-off sufficiently

large to destroy the device.

5.3.2 Gate Drive Circuits

This section describes two different GTO drive techniques which may

be used over a wide range of applications. The appropriate circuit for a given application is largely dependent upon whether or not gate

41

isolation'is needed and on the required range of duty cycle and switching frequency.

Drive circuits are currently the subject of rapid development and improvement, as the range of applications for the devices extends.

5.3.2.1 Un-isolated drive circuit

The simple un-isolated gate drive circuit shown in Figure 5.5 com-bines simplicity with low cost, but it has a limited range of appli-cations confined to areas such as television, switched-mode power supplies, horizontal deflection circuits and simple series-resonant power supplies(50).,

5.3.2.2 Isolated drive circuit

The power circuit of the GTO chopper bridge shown in Figure 5.2 requires isolation of both the control signals and the gate drive for each GTO. The lower two devices have common cathode connections to the negative rail of the DC power supply and therefore present few problems for isolated drives. The upper devices have however independent cathodes, switching at the high voltage levels of the output waveforms. The devices require therefore gate drive isolation circuits that can function correctly under the high stresses likely to be imposed. The gate drive requirements defined-in_Secfion 5.3.1 are achieved using the circuit shown in Figure 5.6.~(50) ,

Isolation of the control circuit from the GTO- is achieved by a pulse transformer with a low leakage inductance and minimum inter-winding capacitance, energised by the switching transistor TRl in Fig.5.6in its primary circuit. The secondary voltage of this transformer is a differentiated version of the primary square,wave, and this is res tored to its ori gi na 1 shape usi ng the two LOC~10S inverters INVa and INVb, which act as a combined Schmitt-trigger and memory circuit,

42

with the output of 1NVa being the reconstituted and restored square wave corresponding to the input control signal. The output of INV , - a buffered by 1NVc to INVf controls the Darlingt.'transistor TR2. When

TR2 is turned off TR3 conducts, and the GTO is turned on by a posi-tive gate current dependent on the network Rg, Cs and R10 . The required pulse of gate current flows initially from the +lOV supply to the gate via R10 and CS. When Cs is fully charged the lower steady-state gate current flows through Rg for the remainder of the on-period.

The GTO is turned off when TR2 is turned on and current is extracted via diode D2 into the smoothing capacitor C4 connected to the -12V supply. To ensure that the gate turn-off current is withdrawn rapidly, the inductance of the loop formed by the GTO gate-cathode Junction, D2, TR2 and C4 must be kept as low as possible-and certainly to less than 1 ~H.

The circuit design is such that the minimum allowable GTO on-and-off-periods are sufficiently low for the circuit to be suitable for PWM motor speed control. There is no lower limit to the switching fre-quency and it can turn-off a peak anode current of 6A over a swit-ching frequency range from DC to S kHz (50)

S.4 Power Supplies

The experimental circuit of Figure S.2 requires three different power supplies. These are the high voltage power supply connected across the bridge input terminals, the mains power supply which provides DC voltages of !16V for the control circuits and !12V for the gate drive circuits, and the mult~ple-output power supply which provides +lOV and -12V to the individual GTO drive circuits. The following sections concentrate on these circuits and their design.

43

5.4.1 High Voltage Power Supply

The high voltage power supply to the chopper bridge must be capable

of receiving power from the drive during regenerative braking, and

for this reason a 230V, 13A OC generator was used as the power supply rather than a 3-phase rectifier..

5.4.2 Main Control Circuit Power Supply

The circuit diagram for the main control circuit power supply is

shown in Figure 5.7. The 240V AC supply is fed to a step-down transformer Tl with a centre-tapped 30V secondary. The output vol-tage of this is rectified and smoothed, using two 470 ~F electrolytic capacitors, to provide a !16V unregulated supply. A! 12V isolated

power supply is obtained, using a fixed positive voltage regulator VRl (type 7812) connected to the +16V rail to provide the positive output, and a fixed negative voltage regulator VR2 (type 7912)

connected to the -16V rail to provide the negative output. ~. -

The common ground of this circuit is connected to the transformer core, the secondary centre-tap, and the input supply earth point. Fuse fl in Fig.5.7

provides overcurrent protection for this circuit.

5.4.3 Gate Orive Multiple-Output Power Supply

Figure 5.8 shows a simple and economical circuit suitable for providing the multiple isolated supplies required by the gate drive circuits shown in Figure 5.6!(50}

The oscillator OSC switches transistor TR1.on and off at about 60 kHz. Transformer Tl has a 1:3 turns ratio, giving a 65V peak-to-peak secon-

dary voltage. Transformers T2, T3 and T4 step this down to about 22V

peak-to-peak, with T2 and T3 providing power supplies for the two

GTOs in the upper section of the bridge, and T4 providing power to the two GTOs in the lower section of the bridge.

44

When transistor TR, is conducting diodes 05 to 08 also conduct, charging the capacitors connected to the positive supply of the gate drive circuit. Conversely, when TRl is turned off, diodes 09 to 012 conduct, and the energy stored in the cores of the transformers Tl to T4 charges the capacitors connected to the negative supply of these drive circuits. The voltage regulator diodes D13 , D14 and D15 limit the negative output to -12V.

5.5 Open-loop Trigger Pulse Generating Circuit

The circuit which provides the open-loop PWM control signals for the drive circuits is shown in Figure 5.9, and the waveforms at various points of the circuit are given in Figure 5.10.

For the modified unipolar control explained in Section 3.3, it is evident that when the desired motor direction is reversed, the chopper s ides exchange their triggeri ng sequence. However, it is vital that the two GTOs in.each chopper side, i.e. 1 and 2 or 3 and 4,

- . must not be turned on simultaneously, since this would create a supply short circuit. During complementary switching, an overlapping off-

- --- - .-------period, slightly longer in duration than either the turn-on o~the ~turn-~ff times of the complementary GTOs is therefore added to the bottom GTO.

The generating circuit for the trigger pulses uses three monostables to provide the two complementary control signals. Monostable 1 is clocked at the chopping frequency of the bridge (fc = 1 kHz, T = 1 ms), and provides a variable-width pul se with a duration between 25 \lS and 925 \lS, depending on the external resistance and the timing capacitance. For the 74LS123 dual monostable:

(5.1 )

where tw is the output pulse width (n5)

45

RT is the external timing resistance (kn) and Cext is the external timing capacitance (pF)

- ----------

and for monostable _1 , RT is a combination of a 5.6 kn resistor and a 100 kn trimmer, with Cext = 20 nF. Monostables -2 and .3 provide overlapping off-periods for the bottom GTO in the complementary side. The pulse duration, and therefore the off-periods is about 10 ~s and is achieved using RT = 10 kn and Cext = 2.2 nF. The output Q2 of monostable2· is ANDed with the output Ql of monostable 1 , to give the resultant pulse QA shown in Figure 5.10. The output pulse ~3 of monostable ·3 is also ANDed with the output ~l of monostable 1 , to give the pulse QB shown in Figure 5.10.

QA is applied to the upper GTO, with QS providing the complementary pulse for the lower section GTO. The gate drive for the permanently-off GTO is connected to ground, and the gate drive for the permanently-on GTO is connected to the +5V supply. rhe presence of a pulse trans-former in the gate drive circuit means that transformation of the 0 and +5V.DC levels is impossible, and to overcome this each control signal is modulated with a 1 MHz signal which is already available in the system. The modulation process is achieved by ANDing the high-frequency signal with each control pulse.

For reverse operation, the interchange of signals between the two

chopper sides is achieved using a tri-state octal buffer (74LS241) as a bus driver. The two enabling pins in a 74LS241 tri-state driver are complementary, and connecting them together to one control signal provides access for only one input group to the output at a time. The enable signal follows the state of direction command switch SW"i via a flip-flop debouncing circuit, to overcome any mechanical effects in the switch.

Switch SW2 provides an i nhi biti ng command faci 1 i ty to shut-down all the gate drive circuit inputs to logic 0, and consequently to turn-off

46

all the GTOs and to stop system operation. although the DC supply is still applied to the bridge.

Open-loop control of the drive is achieved using this circuit. The output voltage depends on the on-period of the upper-GTO in the complementary side of the chopper and this depends on the output of monostab1e 1. which is varied by the trimmer resistance. The bridge output voltage and therefore the applied motor armature voltage varies with the 25 ps minimum pulse width trimmer position. providing from almost zero up to about 90% of the DC supply voltage. with a corres-ponding speed change in the motor.

The circuit was constructed on a single board. as shown in Figure 5.11. and Figure 5.12 shows a photograph of the complete experimental power circuit discussed in this Chapter.

main 240V power

AC mains supply circuit

control circuits

DC power supply

multiple OIP isolated power supply circuit

GTO isolated 4-quadrant drive bridge circuit

PWM drive voltage

triggering pulses generating circuit

Fig 5.1 BLOCK DIAGRAM OF EXPERIMENTAL OPEN-LOOP CHOPPER DRIVE

motor and load

... ....,

48

L,

+ DC f, ~ ~~

ID L , GTO, 7

GT03

n ~ ~ 7 DRIVE SNUBBER DRIVE '- SNUBBER CIRCUIT CIRCUIT CIRCUIT CIRCUIT , , 3 .3

(DC ~ ~D, ~ ,...03 to

!!~ M GT02 GT0 4

n 7 ,. Z '- 7i DRIVE SNUBBER DRIVE SNUBBER

CIRCUIT CIRCUIT CIRCUIT CIRCUIT 2 2 4 4

~ .. O2 ~ ~ 0 4 ..

-DC f2 ,...,... -

Fig 5.2 EXPERIMENTAL 4-QUADRANT (HOPPER BRIDGE

49

A

G

K.

Fig 5.3 RCD SNUBBER CIRCUIT FOR SINGLE-ENDED CIRCUITS

+Dco-----~r_------r_----~---

+-----0 OUT

-Dc o-~----~------~--------

Fig 5.4 SIMPLE SNUBBER FOR BRIDGE CIRCUITS

50

OV

Fig 5.5 SIMPLE DIRECT DRIVE CIRCUIT

Fig 5.6 ISOLATED DRIVE CIRCUIT WITH A WIDE SWITCHING FREQUENCY RANGE (component values given in Appendix 4)

-..

l 0 f, switch T, DC ~ . -

a + 16V

C, b + 12V D VR,.I-I ---~

240V ...... AC MAINS c ov

C2 VR21 0 d -12V

N DIm 1 TaP -16V .....

Fig 5.7 MAIN POWER SUPPLY CIRCUIT DIAGRAM

U1 N

53

+ 16 V unregulat ed

• T, + 12V regulated

R2

74 a

R, osc. 3

6 R3 2s ,

(, (2 Rs

(4 OV

Fig 5.8 MULTIPLE-OUTPUT ISOLATED POWER SUPPLY (component values given in Appendix 4)

JlHI---O~IOV

~~-O~IOII

-12V

D'4 ov

+10V

+10V

-12V

;,.12V

1MHz modulation

MONOST ABLE 2 Tri-state driver signal (14LS123) (74LS241)

A2 I i J ~ 5V -:;b- lA, 1Y, , "' -.... .,..... 2A, 2Y) '--" '--MONOST ABLE 1 B2 0.2 1A2 1Y2

, , ~

(74LS123) I )R 2~ 2Y4 "--' fixed output pulse 110,,5). , , Al a l .Ja...

lA) 1Y) -"* ~[)a;

2A) 2Y, -+vIr

, Bl a l

lA4 1Y4 ,

~

~ ~~~~::IBLE 3 2A4 --....

1kHz 2Y2 clock variable-width r'\ aB ffj 2G

output pulse +vcc A3 rJ +vcc T +vcc B3 a 3 I )R I JR

fixed output pulse SW 1 R-S S SW~ , R-S (10,,5\ T I

)R Inhibit

Direction T I J R S f-F· F F.F command ""';- command -=E="

Fig 5.9 GATE TRIGGERING PULSES GENERATING CIRCUIT

55

1ms -------1\ ~--------_ r-------

Q,

O2 I ~ 10~s

Q3 I 10~s

QA I I I QB I I I

Fig 5.10 PULSE SEQUENCE FOR TRIGGERING CIRCUIT

, .

. ,

,

'.

, . , .

,.". 'I

FIGURE 5.11:

56

Experimental PC Board for Open-Loop Trigger Pulse Generating Circuit

57

FIGURE 5.12: Experimental PC Board for the Complete Power Converter Circuit

58

CHAPTER 6

DIGITAL CLOSED-LOOP CONTROL SYSTEM

In general, control algorithms may be classified as either linear or non-linear. Linear control algorithms are made up from a combination of one or more of proportional, integral, and derivative operations, leading to the most general case of a proportional-integral-deriva-tive (PlO) control algorithm. A recently developedlincremental control algorithm(8) is discussed in detail in Section 6.4, which ensures that the armature current does not exceed a predetermined maximum value.

Conventional digital closed-loop control systems for variable-speed drives usually contain an outer speed loop and an inner current loop, as shown in Figure 6.1. Advances in speed measurement techniques, as described in Chapter 7, enable speed information to be represented and fed back in digital form, whereas feeding back an armature current signal necessitates the use of an analogue-to-digital converter in the current loop.

6.1 DC Motor Transfer Function and Block Diagram

A separately-excited DC motor with armature voltage control is shown in Figure 6.2(a). The armature voltage equation is:

Ka' w + R i + a a

and the torque. balance equation is

(6.1 )

(6.2)

59

where the electromagnetic torque developed by the motor is

Rewriting equations 6.1 to 6.3 in the Laplace domain gives \ ..

\ i , I Me(s} = ML (s) + Fw(s} + Jsw(s}

/1

and ~M'I') • K, .1,1') From equation 6.4:

E (s) - K w(s} I a (s) = _a=---'R;--+-:---::sf'E-- =

a a

[Ea(s} - Kaw(s}]/Ra I + T S a

(6.3)

(6.4)

(6.5)

(6.6)

(6.7)

where Ta = La/Ra' the electrical time constant of the armature cir-cuit, while from equation 6.5

I '\ Me(s} - ML (s) w(s} = F + Js =

[Me(s} - ML(S}]/F 1 + TmS

where Tm = J/F, the mechanical time constant of the motor.

These two relationships are shown combined in block diagram form

(6.8)

in Figure 6.2(b}, where the feedback loop representing the back emf provides the moderate speed regulation inherent in a separately-excited

60

DC motor. From Figure 6.2(b). the change in speed w(s) due to varia-tions in the applied voltage Ea(S) and the load torque ML(s) can be obtained as

w(s) to> • (6.9)

where

(6.10)

G2(s) = - (1 IF l (6.11) 1 + 5Tm ,

H2 (s) -(Ka4» 2/Ra

= 1 + STa

6.2 The PlO Controller

The Proportional-Integral-Derivative (PlO) Controller, with the general form shown in Figure 6.3 is widely used in continuous-data control sys terns.

The function of the proportional term Kp is to achieve optimum accelera-tion in the system response. so as to achieve the demand value in the minimum time. The integral term Kl/s eliminates the steady-state error around the demand value. and the derivative term sKD provides an

61

anticipatory action to reduce any overshoot in the response. Although the 3-term combination is the general form of the controller, any 2-term combination is also possible when the coefficient of the third term is set to zero. Any desired response may be obtained from the system by adjustment of the three coefficients Kp' Kr and KO. l The time-domain re~ation between ,the output U(t) and the input e(t) of the controller 1S .

(6.12)

with the associated s-domain transfer function for the controller being

O(s) (6.13)

Since the 3-term control technique has proven so useful for continuous-time control systems, it is clearly desirable to develop a digital control algorithm which possesses similar characteristics.

6.2.1 The Oigital pro Controller

When the principle of conventional pro control is applied to digital control, there are a number of ways by which the integral and deriva-tive terms may be implemented(52,53).

The digital pro control algorithm may be written(7,54)

n U = K e + Kr I e. + Ko (e - e 1)

n p n j=O J n n- ~ (6.14)

62

where e is the error at the n th

controller output at the n

th 1 . . t t d U . th n samp lng lns an an n lS e sample.

The incremental or velocity form of the pro algorithm followsfrom equation 6.14 as:

or

U - U n n-l

n Le.

j=O J

[K e 1 + Kr p n-

(6.15)

When this general expression is interpreted in terms of the system under investigation, Un becomes the chopper output voltage Van applied to the motor armature at the nth sampling instant and en the speed error

bwn•

The equation representing the experimental system thus becomes

V - V ( 1) = (K + Kr + KO)bw - (K + 2KO)bw 1 + KObw 2 an a n- p n p n- n-

or

(6.16)

.,

63

where:

The chopper period can be divided into a number of equal intervals Tc'

with the ON-period Ton in each chopper period being given by the rele-vant number Dn of the Tc time intervals or

T = D T on n c

The chopper output voltage is then

T = E _c D =

T n (6.17)

where Kch is the chopper constant and E the DC supply voltage. Substi-tuting equation 6.17 in 6.16 and rearranging yields

D = D 1 + C46w - C5~w 1 + C66wn 2 n n- n' n- -~ .. (6.18)

where:

64

Equation (6.18) could be implemented by means of a binary arithmetic circuit •. the output of which can be directly interfaced to the chopper circuit as a PWM gate drive signal. The optimum system res-ponse can be achieved for any controlled system, as the control algo-rithm is not directly dependent on the controlled system constants.

6.3 Current-Limit Control Algorithm

The function of this algorithm is to provide a rapid acceleration/ deceleration rate when the speed error is unduly large, at the same time ensuring that the armature current does not exceed a preset maxi-mum value Iamax ' even if the motor is stalled. Limitation of the current is achieved by restricting the difference between the applied

armature voltage and the generated back emf to a maximum value .~Vmax' speci fi ed by

or (6.19)

If the rotational speed of the motor at the nth sampling instant is

wn' the armature voltage to be applied during this sampling interval is

(6.20)

which can be used as the basis for the current-limit algorithm.

65

Selection of the appropriate sign in equation 6.20 depends on whether the speed error is positive or negative, i.e. whether the system should accelerate or decelerate towards the demand speed. The sign adopted is the same as that of the error, and equation 6.20 can be rewri tten

(6.21 )

Equation 6.21 shows that Van is determined from the latest sample of motor speed and· does not depend on either the previous values of Va and w or the sampling period. This period can therefore be made as short as is necessary to obtain an accurate speed sample,while still performing the mathematical operations involved.

It follows from reference to equation 6.17 that the output number representing the width of the gate pulse driving the upper GTO in the complementary side of the chopper is

(6.22)

where wn is the motor angular velocity sampled at the output of the speed monitor.

T Cl = ~ Kb (a constant)

c

C - T lIVmax (a constant) 2-q

66

Different rates of acceleration and deceleration are obtained simply by varying C2, which in practice results in the variation of Iamax.

The control algorithm requires fast speed measurement, such that the sampling period is very small when compared with mechanical time con-stant of the system. This ensures satisfactory armature current 1imi-ting"during transient conditions, even when the drive is stalled.

6.4 ilncrementa1 Control Algorithm

Non-linear control may be developed for use as a second algorithm in conjunction with the current 1 imit control, so as, to provide smooth transient changes with no overshoot in the controlled speed variation when the speed error is smaller than a fixed value. This algorithm development is based on ensuring that the armature current does not exceed Iamax ' even at the instant the control system switches to the

I_incrementa~ ,a1gorithm~ At the (n-1th sampling instant Va{n-1) is determined by the current-limit algorithm as

(6.23)

and if the control system operation switches to thelincreme:tal~ algo-rithm at the nth sampling instant with the applied arm1ltur~61tage being incremented by AV, then

(6.24)

Substituting from equations 6.23 and 6.24 in equation 6.19

67

or

6V ~ Kb (w - w 1) n n- (6.25) \ ..

Multiplication of the right-hand side of this inequality by a factor that does not exceed unity ensures that 6V is never greater than 6V . . max Within the operating region of the non-linear controller, the speed error is smaller than a fixed value 6w, so that

or the magnitude of [(wr -w)/6w] is less than unity as Lal control. algorithm is operational. Use of this

in the equation for the control algorithm becoming

long as the:i ncrementa 1 , . ratio results

(6.26)

where 6w is the speed error limit determining which one of the two control algorithms is operating at each sample. Although this parame-ter may be determined from a tuning process, it seems sensible to relate 6w to 6Vmax through the logical relationship

6w = = R I a amax

Kb (6.27)

68

which results in the active operating zone of thelincremen~al.control algorithm having a variable width which depends on the value of Iamax' Comparison with equation (6.22) shows that this is the same parameter that controls the rate of acceleration/deceleration. It can be con-cluded from equation 6.26 that thelincremen{alcontrol algorithm is essentially a modified integral control process, in which the