Novel Zero-current-transition Pwm Converters

5/24/2018 Novel Zero-current-transition Pwm Converters

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NOV EL ZERO-CURRENT-TRANSITION PWM CONVERTERSGuichao Hua, EricX. Yang, Yimin Jiang, and Fred C. Lee

Virginia Power Electronics CenterThe Bradley Department of Elecmcal EngineeringVirginia Polytechnic Institute & State UniversityBlacksburg, VA 24061ABSTRACT

A new family of zero-current-transition ZCT) pulse-width-modulated PWM) converters are proposed. The newfamily of converters implements zero-current turn-off forpower transistor s) without increasing voltagekurrentstresses and operates at a fixed frequency. The proposedconverters are deemed most suitable for high-power applica-tions where the minority-carrier semiconductor devices suchas IGBTs, BJTs, and MCTs) are predominantly used as thepower switches. Theoretical analysis is verified ona 100kHz,1 kW ZCT-PWM boost converter using an IGBT.

I. INTRODUCTIONRecently, a number of soft-switching PWM techniqueswere proposed aimed at combining desirable features of boththeconventionalPWM andresonanttechniques l-161. Amongthem, the zero-voltage-transition (ZVT-PWM) technique isdeemed desirable since it implements zero-voltage switching(ZVS) for all semiconductor devices without increasing vol-tage/current stresses [ l l , 15, 163. This technique minimizesboth the switching losses and conduction losses and isparticularly attractive for high-frequency operation wherepower MOSFETs are used as power switches.Due to continuous improvement of switching characteris-tics,lowerconduction osses, andlower cost, IGBTsare gainingwide acceptance in switched-modepower converters/inverters.Since IGBT is a minority-carrier devices, it exhibits a currenttail at turn-off which causes considerably high turn-offswitchinglosses.TooperateIGBTsatrelatively high switchingfrequencies,either the ZVS or the zero-current switching ZCS)technique can be employed to reduce switching losses. Basi-cally, ZVS eliminates the capacitive turn-on loss, and reducesthe turn-off switching loss by slowing down the voltage riseandreducingtheoverlapbetweentheswitchvoltageandwitchcurrent. This technique can be effective when applied to a fast

IGBT with arelatively small currcnt tail. However, employingZCS technique eliminates the voltage and current overlap byforcing the switch current to zero before the switch voltagerises. ZCS is deemed more effective than ZVS in reducingIGBT switching losses, particularly for slow devices [17, 181.It iswell-known that the conventional resonant converters, such asthe parallel-resonant converter, series-resonant converter, andLCC-type converters, can achieve ZCS for power transistorswhen operated below the resonant frequency [20, 221.

Several ZCS techniques were reported [19-261.

0-7803-1243-0/93$03.00Q 1993IEEE

Constant-frequencyoperation is also achievable f phase-shicontrol is employed in full-bridge opologies [22]. Howeverthis type of circuits usually are operated with large amount ocirculating energy and thus require the power devices and othecircuit components to be rated for significantly higher V Aratings, as compared to their PWM counterparts.As a compromise between the ZCS resonant and PWMtechniques, the ZCS quasi-resonant (QRC) technique was firsproposed [23-261. While the underlying power conversionprinciple of ZCS-QRCs is similar to that of the PWM converters, a resonant network is employed to shape the switchcurrent waveform so that the power transistor is operated withZCS and the rectifier diode with ZVS. Compared with thconventional resonant converters, the ZCS-QRCs operate withless circulating energy. However, hese converters operate withsinusoidal current through the power switch(es) which resultsin high peak and rmscurrents for the power transistors and highvoltage stressesonthe ectifier diodes. Furthermore, when theline voltage or load current varies over a wide range, ZCS-QRCs are modulated with a wide switching frequency rangemaking the circuit design difficult to optimize.This paper presents a new family of zero-current-transitio(ZCT) PWM converters. The proposed ZCT-PWM converterimplement ZCS turn-off for the transistors without substantially increasing voltage/current stresses of the switches. Theyare particularly suited for high-powerhgh-voltagapplications where the minority-carrier semiconductor device(such as IGBTs, BJTs, and MCTs) are used as the poweswitches. In the following section, the ZCT-PWM boosconverter is used as an example to illustrate the principle ooperation.

11. PRINCIPLES OF OPERATIONThe circuit diagram and key waveforms of the ZCT-PWM

boost converter are shown inFig. 1.The converter differs froma conventional PWM boost converter by the introduction of aresonant branch, which consists of a resonant inductor, L,,resonant capacitor,C,, an auxiliary switch,S1, and an auxiliarydiode, D1. This resonant branch is active only during a shorswitching-transition time to create the ZCS condition for thmain switch. In the analysis, the boost inductor, L,, is assumeto be large enough to be considered as a current source,1;. Insteady-state, five operating stages exist within one switchingcycle (refer to Fig. 2):

538

i T I I


2

I

,: D current to decrease in a sinusoidal fashion. After a quarterof the resonant period, hlr he C, voltage reduces to zero,and the L, current reaches itsmaximum value, I-14 (1)dl =-T, ,and

. . .I Is1

To T i T2 T3 T4 TOFig. 1. Circuit diagram and key waveforms of the ZCT-PWh4 boost converter.

(a)TO - T l (b)T1 - T2 ( c )T2 - T3

(d) T3 - T4 (e) T4 - TO

Fig. 2. Equivalent circuits for five topological stages.(a)To-T,: Prior to To, he main switch,S, is conducting, and C,

is charged with certain negative voltage, -VaPc*. AtTo, heauxiliary switch, S1, is turned on, starting a resonancebetween C, and L,. This resonance forces the transistor

where T, = 2 m a n d , =m a r e he resonant periodand impedance of the resonant tank, respectively. It can beseen that to achieve ZCS for the transistor, IbF* has to begreater thanIi. After the transistor current drops to zero andits anti-parallel diode startsto conduct, the gate-drive signalof S is disabled at t = t + t d .

@)TI-T,: S1 is turned off shortly after S is turned off. Insteady-state operation as explained later), the resonantinductor current at T, is always equal to I;. Thus the timedelay between these two gate-drive turn-off signals, fadetermines the peak voltage of C,:Zr iv,q = cos(27cT,/Tr) 3)

if v c y . (4)In steady-state operation,V C r J ;annot exceed V,, since D1would otherwise conduct during operating stage T4-T,.Defininga 2 x ~q. (3) becomes:f2

Combining Eqs. (2-5) yields:2 i ,i---

k - osa

(5)

which means that as long as inequality (4) is satisfied, ZCSoperation will be guaranteed regardless of the input voltageand load current. In a practical design, Tdzcan be selectedat around 0.45T1,sothat ILF* is about 10 percent higherthanIi, and ZCS is ensured. When S1 is turned off at T1, both Dand D1 will st rt to conduct, and L, and C, continue toresonate until the I, current decays to zero at T,.

(c)T,-T,: AtT2,L, and C, complete the half-cycle resonance,and D1 is reverse-biased. This operating stage is identicalto the transistor-off stage of the PWM boost converter.(d) T,-T,: At T3, S is turned on, and the boost inductor ischarged by the input voltage. Meanwhile, C, and L, orm ahalf-cycle resonance through S and the anti-parallel diode ofS1, which reverses the polarity of the C, voltage.

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(e)T,-To: Operation of the circuit is identical to that of thetransistor-on period of the PWM boost converter. At To,S1is turned on again, and the switching cycle is repeated.It is interesting to note that in steady-state operation, theenergy stored in the resonant tank remains constant over theentire switching cycle. This canbeclearly seen from the abovedescription. During each topological stage (shown in Fig. 2),either the voltage across the resonant tank Lrplus C,) is zeroor the current bough it is zero, so there is no energy transferbetween the resonant elements and other parts of the circuit.Figure 3shows the state-plane rajectory of the resonant tankThe energy stored in the resonant tank, which is self-adjustedin accordance with line and load conditions, can be given by:

some energy into the resonant branch during this operatingstage. Therefore, the energy stored in the resonant branch willincrease until it reaches the balance point given by Eq. 7).

(a) l L r > i b) lL ciFig. 4. An additional topological stage occurs when the circuitoffsets the balance operating point: (a) I,>Ii, and (b)1,cIi at T,.

K j .........+............*.............+.............+.......... ..~

2E,,, = L r ( ) .cosa 7)

I lr0 8 K f

0 b K #

Fig. 3. State-plane rajectory of the resonant tank.This circulating energy increases as input current increases(when line voltage decreases or load current increases). In apractical circuit, since the resonant transition time is very shortwith respect to the switching cycle, the resonant inductance isvery small compared to boost inductance. Therefore, thecirculating energy of the ZCT-PWM converter is quite smallcompared to a conventional ZCS resonant converter, as illus-trated in the design example given in Section IV.

It wasmentioned that in steady-stateoperation, heresonantinductor current at T, is always equal to I i regardless of line orload condition change. Assuming that for some reason I, atT, is larger than Ii. In this case, there will be an additionaltopological stage inserted between mode (b) and mode (c) inFig. 2, as shown in Fig. 4 a). During this operating stage, theresonant branch transports energy to the load. Therefore theenergy stored in the resonant branch decreases. In contrast, ifI at T, is lower than Ii, there will be another topological stageinserted between mode (b) and mode (c) in Fig. 2,as shown inFig. 4 b). It can be seen that the boost inductor will pump

0 K * ........ +. +. +. .......1 8 s 70813 2 2 Y 2 1 s 2 b u s 2eor

, Y r ..................................................... .............,L YR ,

b K t

0 4 K f

.0 2 K i

Ia

t

I-.--Mll Y

0 ? K + . .............,.............+.............,1 8 s 2 0 " s 22 s 24 s 2hus 281150 )Fig. 5. Typical transistor V/I wavcforms of boost convertersusing an additional inductor for damping reverse-recovery of the diode: (a) with the ZCT-PWM tech-nique, and (b) with the PWM technique.

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From steady-state operation, it can be seen the ZCT-PWMtechnique implements ZCS turn-off for the power transistorwithout penalizing the voltage stresses of both the powertransistor and the rectifier diode. Although the main switchcurrent waveform exhibits a resonant peaking, it does notincrease the conduction loss, since the average current throughthe power switch (IGBT) is essentially the Same compared withits PWM counterpart. Another unique advantage of theproposed technique is that it has minimum circulating energy.Equation (7) reveals that regardless of the line and loadchanges,the energy stored in the resonant tank will alwaysbeadaptivelyadjusted so that it is only slightly higher than what is neededfor creating the ZCS condition.

For high-voltage applications (such as power-factor cor-rection (PFC)) where the boost diode suffers from a severereverse-recovery problem, an additional inductor (or asaturable inductor) in series with the rectifier diode or the mainswitch is usually used to suppress the reverse-recovery probllem. At transistor turn-off, this inductor invokes a high-voltagespike on the transistor due to high di/dt across the inductor. Tosuppress this voltage spike, a large dissipative snubber isfrequently used. For a ZCT boost converter with the sameadditional inductor, however, this voltage spike is muchreduced due to controlled di/dt across the inductor at transistorturn-off. As a result, a much smaller snubber can be used toabsorb this ringing. Figure 5 shows the simulation results oftypical transistor voltage waveforms in these two cases.Similarly, for isolated topologies, the ZCT technique cansignificantly reduce the transistor turn-off voltage spike causedby leakage inductance of the transformer.

III. EXTENSION OF THE ZCT CONCEPT TOOTHER TOPOLOGIES

From the circuit topological point of view, every reso-nant-type converter (including the conventional resonant,quasi-resonant, and multi-resonant [27,281) can be viewed asa variation of its PWM counterpart. By incorporating certaintype of resonant network, it creates aresonance to achieve ZVSor ZCS. For different resonant converters, of course, the typeof resonant network employed is different.One common characteristic of resonant-type topologies isthat they all employ a resonant inductor (sometimesa resonantinductor plus a resonant capacitor) in series with the powerswitch or the rectifier diode to shape switch voltage/currentwaveforms. Soft-switching is achieved by utilizing the reso-nance between this resonant inductor and certain resonantcapacitors,which are usually in parallel with the semiconductordevices. Due to the fact that these resonant elements are placedin the main power path, the resultant resonant converters arealways subjected toinherent problems. First, since theresonantinductor is subjected to bidirectional voltage, it inevitablygenerates additional voltage stress on the semiconductordevices. Second, sinceall he power flows through the resonantinductor, substantial circulating energy is always created,which significantly increases conduction losses . In addition,the energy stored in the resonant inductor strongly depends on

the line voltage and load current. Therefore, soft-switchingcondition is sensitive to line voltage and load current changes.This is why most resonant converters are unable to maintainsoft-switching for a wide line and load range.To alleviate the above-mentioned limitations, it is neces-s ry to remove the resonant element@) rom the main powerpath. Instead of using a series resonant element, an alternativeway is touse a shunt resonant network across the power switch.During the switching transition, the shunt resonant network isactivited to create a partial resonance to achieve ZVS or ZCS.When switching transition is over, the circuit simply revertsback to the familiar PWM operating mode. In this way, theconverter can achieve soft-switching while preserving theadvantagesof the PWM converter. In fact, the above-suggestedconcept of using a shunt resonant network has been adopted inthe ZVT-PWM converters E11,151. Apparently it can also beapplied to implement ZCT, as illustrated above.It should be mentioned that ZVT or ZCT can be implem-ented in perhaps a number of different ways. One should keepit in mind that it is desirable to minimize the circulating energyand circuit complexity. Furthermore, it also is desirable toimplement some form of soft-switching for the auxiliary switch

as well.The ZCT concept can be extended to any switched-modepower converters or inverters. Figure 6shows six basicZCT-PWM topologies. Figure 7shows a three-phase ZCT-PWM boost converter. In this PFC circuit, employing a smallresonant commutation network can implement ZCS for all sixpower transitors. The features of the ZCT-PWM convertersare summarized in the following:

(a) Buck (b) Boost

(c ) B U c k - h S t (d) Cuk

(e) %pic f) Zeta

Fig. 6. Six basic ZCT-PWM topologies.541


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ZCS for the power switch,low voltage/current stressesof the power switch and rectifierdiode,minimal circulating energy,wide line and load ranges for ZCS,constant-frequency operation.

Fig. 7. Three-phase ZCT-PWM boost converter.

IV. EXPERIMENTAL VERIFICATIONA 100 kHz, 1 kW ZCT-PWM boost converter wasimplemented to illustrate the operation of the new converters.The circuit is regulated at 400 V output with a 200-300V inputrange. The circuit diagram of the experimental converter isshown in Fig. 8.Inthe breadboarded converter, the main powerswitch is implemented by an IR fast-series IGBT, IRGPF40V,=600 V, I,40 A, &=37 S, and T 4 2 0 nS, rated for up to8 kHz switching frequency operation). Since the auxiliaryswitch only handles a little resonant transition energy, a smallMOSFET, IRF830, is employed. The small diode in series withS1 is used to block its slow body-diode from conduction. L,and C, are selected at 10uH and 8.2 nF, respectively. With thisdesign, the maximum circulating energy of the circuit, whichoccurs at full load and low line, is approximately lqW, whichis less than 2% of the output power.

Lf DHEXFRED-10-0.45mH +Cr 8.2 n F CO

47uF2% 400 vTI-D1WVP20600-300 V IRGPCWF S1IRF830i w 2 0 6I I I 1Fig. 8. Power stage circuit diagram of the 100kHz, 1 kWZCT boost converter.

It is shown that ZCS is always maintained when the linevoltage or load current changes in a wide range. Figure 9(a)shows the oscillograms of the ZCT-PWM boost operating at250 V input and 700 W output. Compared with waveforms ofthe PWM circuit operating under the same conditions, it canbe seen that the IGBT turn-off current tail is essentiallyalleviated. Figure 10shows the efficiency measurements oZCT and P W M oost converters. It can be seen that the ZCTtechnique significantly improves the efficiency. Due to thehigh turn-off switching loss of the IGBT device, the hard-switchedPWMcircuitisnotable ooperateabove800W outpupower. Table I shows the loss breakdown estimation for twoboost converters operating at 250 V input and 700 W outputFor the PWM converter, it can be seen that the major powerdissipation comes from the turn-off switching loss of the IGBTwhich is about 37 W. The estimation of IGBT switching lossesis based on the actual switch voltage/current waveformsroughly in agreementwith thedatagiven in the IGBTdesigner'smanual [29]. For the ZCT circuit, power losses involved in theoperation of the auxiliary resonant shunt are only about 1.8 W,which is less than 0.3% of the output power.

(a) Using ZCT-PWM technique

v,, 20 VldivV,, 200 V/div

Is, A P I V

VD, 200 v/div(b) Using PWM technique

Fig. 9. Oscillogramsof two IGBT boost converters operatingat Vi=250V, P,=700 W , and f = l OO kHz.

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Efficiency Yo)

resonant tankdiode conductiondiode switchingboost inductor

otherstotalefficiency

0 - r and C J

2 400 6 800 1000

NIA 0.2w1.7 W 1.7 W2.9W 2.9 W2.0w 1.9W1.5 W 1.5 W

51.4 W 21.0w93.1 97.1

Output Power (W)Fig. 10. Efficiency comparison of ZCT-PWM and PWMboost converters using IGBT.

Table I. Loss breakdown of PWM and ZCT-PWM boostconverters at P,=700 W and Vi=250 V.

IGBT conduction3.0 W

I IGBT turn-off switching 11 37.0 W I 5.2 Wauxiliary switch N/A I 1.0W II auxiliary diodes N/A I 0.6W I

V. CONCLUSIONSBecause of continuous improvement of switching charac-teristics, lower conduction losses, and lower costs, IGBTs regaining wide acceptance in today's power processing circuits,particularly in high-power applications. In practice, due to highswitching losses (mainly turn-off current tail), the switching

frequency of the hard-switched IGBT circuits has been limitedtolow tens ofkHz range. Performance (efficiency, size, weightand EM1 noise, etc.) of these converters can be improvedsignificantly by implementing ZCS and thereby boosting theoperating frequency. Although a number of ZCS esonantechniques have been presented to date, they all have severelimitations, suchas high circulating energy, limited load rangevariable frequency control, and complicated design.Based on the conceptof shunt resonant network, thispaperpresents a novel ZCT technique andanew family of ZCT-PWMconverters. The proposed converters feature ZCS turn-off fothe power transistor, while retaining the advantages of con-ventional PWM converters, such as low voltage/cunenstressesof the semiconductor devices and constant-frequencyoperation. The shunt resonant network operates only duringswitching-transition, thus during most portions of the switchingcycle, the converter operates like a PWM circuit. Conse-quently, the design of the power stage components and contro

is similar to that of the conventional PWM convertersFurthermore, ZCT operation is independent of line and loadconditions, and the circulating energy of the. converterisalwaysmaintained minimum. These features make the ZCT-PWMtechnique attractive for many applications where theminority-carrier devices such as IGBTs and MCTs areemployed.REFERENCES

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Novel Zero-current-transition Pwm Converters