On-Wafer Vector Network AnalyzerCalibration and Measurements

The Vector Network Analyzer orVNA has become the workhorse ofmost network measurements above 1GHz. Getting the best on-wafer mea-surement results requires a solidunderstanding of measurement systemcomponents and their interaction.This application note is intended tointroduce on-wafer vector measure-ments and provide references for fur-ther study.

VNA Calibration FundamentalsThe representative VNA block

diagram (see Figure 1) shows thekey elements of the analyzer. TheRF source at the top provides thedevice-under-test (DUT) stimulus.A forward/reverse switch directs theRF energy to either DUT port. Thetest set uses directional couplers orbridges to pick off the forward andreverse waves traveling to and fromeach port, and down-converts thesesignals to four IF sections. Each IFsection filters, amplifies, and digi-tizes the signals for further digitalprocessing and display.

A VNA measures vector ratios ofreflected or transmitted energy toenergy incident upon the DUT. AS

a stimulus-response measurement, aVNA measurement determines theproperties of devices rather than theproperties of signals. Signals wouldbe measured by instruments such asoscilloscopes or spectrum analyzers.

A significant challenge in stimu-lus-response measurements is defin-ing exactly where the measurement

CASCADE®

Port-l b_+Cable y3

D U T _

a2

) P o r t - 2

Figure 1. Main hardware blocks in a Vector Network Analyzer (VNA).

system ends and the DUT begins. erence plane. If we ignore any non-In on-wafer VNA measurements idealities of the cables, couplers,this boundary is known as the “ref-erence plane” of the measurementand will often be located at theprobe tips. A typical wafer probingVNA setup is shown in Figures 2and 3.

Ideally the system will measurethe characteristics of whatever isconnected to the measurement ref-

mixers, and so forth, then the ratiosof wave amplitudes (a’s and b’s)inside of the machine corresponddirectly to the DUT's S-parameters.For example:

b,/a, = S, 1 of the DUT

VNA calibration is the process ofmeasuring devices with known or

Figure 2. A typical RF device characterization setup is shown above. On the left is a vector net-work analyzer, in this case an Agilent 8510 system. Cables from the test set ports route over theprobe station to a pair of microwave probes which are aligned through the microscope or camerasystem onto calibration substrates and test devices. The right side of this picture shows the PCthat controls the prober and runs the calibration and measurement software. 1

partly known characteristics andusing these measurements to estab-lish the measurement referenceplanes. Calibration also corrects forthe imperfections of the measure-ment system. These imperfectionsnot only include the non-idealnature of cables and probes, butalso the internal characteristics ofthe VNA itself

The VNA is calibrated in muchthe same manner that the “zero”function on your ohmmeter sub-tracts out the resistance of the testleads. On an ohmmeter, when youactivate the “zero” function, itstores a resistance measurementwhich is then subtracted from allfuture measurements. The obviouserror model is simply a resistance inseries with the test port. CorrectedVNA measurements are referred toas “deembedded.”

A VNA does the same thing asthe ohmmeter but uses a morecomplex error model with severalterms for each frequency point. Themeasurement system is described asan ideal VNA with an error adapternetwork that models all of the sys-tem’s non-idealities: directivity ofthe couplers, imperfect match look-ing back into the reflectometer (testset ports), imperfect frequencyresponse of the reflectometers andthe transmission between ports, andthe crosstalk between ports. Thiserror model is shown in Figure 4.

VNA's rely on calibration foraccuracy even in measurementswith the reference plane defined atthe instrument front panel connec-

Figure 3. Close-upof microwaveprobes and cableson positionersproviding probeplanarity adjust-ment and cable

tors or at the ends of cables.Calibration functions allow theuser to store measurements of stan-dards, compute the error models,and automatically apply correctionsto DUT measurements.

Where the ohmmeter’s calibrationsimply used a short circuit (con-necting the test leads together) todetermine the extra resistance termin the error model, a VNA usesmultiple calibration standards -typically open circuits, short cir-cuits, loads, and through (thru)connections.

Measuring ImpedanceThe measured ratios of the ai and

bi wave amplitudes are calledS-parameters. S-parameters are justone type of network representationused for linear, small-signal, a.c.analysis.

A reflection ratio at a device port,with all other ports terminated, isalso known as a reflection coeffi-cient. The load impedance, Z,, isrelated to the reflection coefficient,I?, by the bilinear transform pair asfollows:

Figure 5. The Smith Chart is a polar graph ofreflection coefficient with grid lines showingnormalized impedance values.

zL1 +l-=zo -

[ 1 = r+ix1 -I-

r= V reflected zL - zll=

V incident Z L + z.

where Z,= the reference imped-ance of the measurement system

The Smith chart (see Figure 5) isa polar graph of the reflection coef-ficient with grid lines showing thenormalized impedance values. Allpassive impedances are representedin a compact unit circle reflectionformat. Typically the v n a auto-mates the display of a wide varietyof formats: Smith Chart, polar, andlinear and log magnitude formats.

Calibrated VNA AccuracyAs with any corrected measure-

ment, the absolute accuracy of acalibrated VNA measurement isdetermined by the techniques andcompleteness of the error modelused, the accuracy of the descrip-tion of the reference devices (cali-bration standards), and the repeata-bility of the measurement system.

ForwardP 1

ErrorAdapter

Port 2

DUT3Pl

Figure 4. Error model for a typical network analyzer. All errors are modeled by an error adapterin front of an otherwise perfect system.

2

If performed improperly, calibra-tion can introduce errors. Theimpedances used in the calibrationmust be accurately known andentered. Assumption of idealbehavior of standards is a recipe forerrors, but worse things can hap-pen. For example, improperlyentering a description of a shortcircuit or open circuit standard willlead to useless results that may notalways be obviously wrong. TheVNA will believe everything thatyou tell it. Accurate descriptions ofthe electrical behavior of the vari-ous calibration standards must besupplied to the VNA.

Corrected measurements rely onthe repeatability of the measure-ment system. Just as an ohmmetercannot correct for a changing resis-tance of its test leads or drift in theohmmeter circuitry, a VNA cannotcorrect for random errors such asnoise or dynamic range, cablerepeatability, or instrument drift.Any change in the measurementdue to, for instance, drift of theVNA test set, thermally inducedcable length changes, or even theeffects of noise due to VNAdynamic range can invalidate thecorrection. Sensitivity of cable elec-trical performance (such as phasedelay) to environmental changes isa significant element of quality andsuitability for VNA use.

VNA error models are basedupon the use of S-parameter repre-sentations of network properties. S-parameters are signal flow andtransmission line based. Only a sin-gle mode of propagation at deviceterminals is assumed. Situationsthat violate this assumption such asusing waveguides that can propa-gate multiple modes, radiation, orparasitic coupling between net-works other than at the device ter-minals are not properly handled.

If a second mode, radiation, orextra parasitic doesn’t change forany and all devices being measuredthen it will drop out with the cali-bration. Usually, however, theseextra modes have behavior that isDUT dependent and VNA calibra-tion will not account for theeffects. A clean, well-designedprobing system (including die padpattern) with good quality trans-mission-like interconnections willminimize these errors as much aspossible (see Figure 6).

An extensive discussion of VNAcalibration and accuracy is includedin Calibration and AccuracyFactors Summit High-FrequencyProbe Station Reference Manual. l

ground

signal

ground

contact pitch100 - 150 pm

100 - 150 pm

VNA Calibration Options andStandards

The VNA will measure theuncorrected S-parameters if not cal-ibrated, though not very accurately.Uncorrected measurements arerarely used (see Figure 7).

l Convenientl Generally not accuratel No errors removed

Figure 7. Uncorrected VNA.

A “response” calibration is simplya vector magnitude and phase nor-malization of a transmission orreflection measurement, used onlyat low frequencies (See Figure 8).

0thru

0

l Use when highestaccuracy is not require’

l Removes frequencyresponse error

Figure 8. Response calibration.

Typically a full calibration of all theerror parameters is used as a refer-ence or to assure the highest accuracy(see Figure 9). 2

4 DUT I-0I I

Highest accuracy for P-portdevices

Removes these errors- directivity- source match- load match- reflection tracking- transmission tracking- crosstalk

Figure 6. Recommended microwave pad pattern for each measurement port. Figure 9. Full 2-port calibration.

Figure 10. The LRM Impedance Standard Substrate (ISS) has multiple repeated patterns ofShort-Load-Thru Standards and alignment marks for setting proper probe separation. A set oftransmission lines are also provided for TRL Calibration.

Calibration StandardsOn-wafer calibration standards

most often are precision thin-filmresistors, short-circuit connections,and 50 ohm transmission lines fab-ricated either on the wafer contain-ing the DUT or on a separateImpedance Standard Substrate(ISS). Cascade Microtech trimshigh-performance ISS load resis-tors to within 0.3% of theirdesired d.c. value (usually50 ohms). Open circuit standardsare normally implemented by rais-ing the probe in air above thewafer by 250 pm or more.

A true 'thru' standard does notexist for on-wafer measurementssince probes cannot directly con-nect to each other and mustinstead use a very short transmis-sion line as a thru standard. A thruline standard may be referred toeither with the label of thru orline. For the Cascade MicrotechLRM ISS (see Figure 10) the thrulines have 1 ps delay for the ver-sion suitable for probes with con-tact pitch (center-to-center spac-ing) of 250 pm or less, and 4 psdelay for the wide-pitch versionused for probes with more than250 pm pitch.

It is important to note that thespecific electrical behaviors of thestandards depend upon the probe

pitch used. The calibration data istherefore, supplied with the probe,rather than being a single value fora standard as would be the case forcoaxial standards where there is noambiguity about what connectionscan be made to it. Calibration datais normally specified for a specificprobe spacing used with a particu-lar impedance standard substrate,and is only valid for a particularprobe-ISS combination. The con-tact pattern, e.g., ground-signal-ground or ground-signal, (seeFigure 11) will also impact calibra-tion data (see Table 2).

Figure 11. Microwave probe contact pat-terns. The Ground-Signal-Ground (GSG) pat-tern has higher performance than theGround-Signal (GS/SG) configuration.

On-wafer standards, fabricatedon the same wafer as the DUT, aresometimes desirable since theprobe-to-standard transition can bedesigned to be very similar to thetransition to the DUT. This ishelpful at frequencies above20 GHz since a primitive transi-tion may introduce extra parasiticsor modes which, since they areoccurring at the probe-tip refer-ence plane, may not be correctedby calibration. In many cases thesetransition errors may be deembed-ded using simple lumped elementequivalent circuit models.3

The use of off-wafer standards onan Impedance Standard Substrateis very practical and CascadeMicrotech has demonstrated cali-bration results comparable toresults obtained by using the spe-cial on-wafer standards methodsrecommended by the United StatesNational Institute of Standards andTechnology. In cases where thetransition on-wafer is dramaticallydifferent than that of the ISS, suchas on silicon substrates, the use ofa set of dummy pads on-waferallows the removal of additionalpad parasitics. Pad parasiticremoval algorithms are discussed inOn the Characterization andOptimization of High-Speed SiliconBipolar Transistors.4

A very useful feature to look foron probe stations is an auxiliarychuck or independent vacuumhole pattern specifically for hold-ing small substrates such as an ISS.The ability to have both the testwafer and the calibration standardsimmediately accessible eliminatesfrequent swapping of the waferand the ISS. In Cascade Microtechthermal chuck systems the auxil-iary chucks are insulated from thethermal system, allowing bettercontrol of the standards. The thinfilm load resistor has a temperaturedependence, making the auxiliarychuck system a must for obtainingthe most accurate results.

Transistor CharacterizationMeasurements

In characterizing devices on-wafer, the goal is to obtain the elec-trical behavior of the intrinsicdevice. This is the transistor with-out the parasitics associated withbond pads and interconnections.

Cascade’s WinCal software sup-ports pad parasitic removal, whenmeasurements of identical pad con-figurations without an active deviceare used to deembed intrinsicdevice behavior (See Figure 12 &13).

Magnitude of l-l,, (db)30 ,

-20 ------1.8GHz 3.3 GHz

0.1 1 10

Frequency (GHz)

Figure 13. Apparent bandwidth improvementdue to removal of pad parasitics.

After a probe-tip calibration, theopen-circuit admittance parametersof open-circuited pads are subtract-ed from device measurements toeliminate shunt capacitances andconductances. Also, series resistanceand inductance are eliminated bysubtraction of short-circuit imped-ance parameters obtained frommeasurements of shorted pads andinterconnections.

Apparent device fr can be signifi-cantly increased by eliminating padparasitics. But be wary of largeincreases in fT when residual errors

Ld

GSG Test Device Open Pads & Metal Shorted Metal(Remove Y pad) (Remove Zmetal)

Figure 12. Pad Parasitic Removal structures.

in pad measurements are signifi-cant. Also, series inductance fromlayout inductance and/or probeplacement errors can resonate withdevice output capacitance and dis-tort f,. results (See Figure 14).

Calibration Standards forPyramid ProbeTM Cards

Calibration standards for use withPyramid Probe Cards differ littlefrom those used in standardmicrowave probing. In many casestraditional ISS's may be used. Thefixed input to output spacing ofprobe cards may be inconvenient,however. Conventional separateprobes can be adjusted to land onpairs of standards simultaneously.For probe card contact separationsdiffering from the distance betweenstandards on ISS's, separate land-ings for each port are required.Additionally, the thru line may notbe the appropriate length to con-nect the input and output.Transmission line standards on theISS may be used as the thru if theyhappen to be the length thatmatches your probe contact separa-tion. The delay and loss of the thrumust be accounted for in your cali-bration.

Effects of increasing L, Cd,

log f

Figure 14. Inductance, Ld, due to layout inductance can resonate with device outputcapacitance and distort the measured frequency response.

The General Purpose MembraneISS (see Figure 15) addresses theproblem of thru length by provid-ing a number of microstrip andcoplanar waveguide (CPW) lines

Figure 15. The General Purpose MembraneImpedance Standard Substrate.

with a variety of straight and rightangle lengths. If no line fits wellenough, the next larger line can becut to length. Loss and delay datafor the straight CPW lines on theGP Membrane ISS are shown inTable 1. Loads trimmed to 1%accuracy are provided in openingsin a large field of metal which isalso used for the short circuit stan-dard.

The RFC-CAL is a custom mem-brane with specific length thru con-nections mounted on a glass slide.It is sometimes useful to use a thru

Line # Delay (ps) ; 10 GHz Loss (dB)1 3.8 ! 0.0252 5.8 i 0.0443 11.5 I 0.0934 17.1 0 1 45 22.8 -tm--PO:,96 28.3 0.247 34.2 0.28

-8 39.7 0.34.~9 45.3 0.3810 52.7 0.4411 60.4 0.5212 67.7 0.59

Table 1. Properties of CPW thru lines on theGeneral Purpose Membrane ISS.

5

structure with an inverted match-ing network for calibration ofimpedance matching probes.

A fully custom ImpedanceStandard Substrate is yet anotheralternative. This ISS can have pat-terns that provide each standardtype to all of the microwave portssimultaneously. A custom ISS maybe particularly valuable whenautomating calibration. ContactCascade Microtech for details onthese options.

Short-Open-Load-ThruCalibration

By far the most commonly usedmethod, the Short-Open-Load-Thru (SOLT) calibration is avail-able on every commercially avail-able VNA. This calibration is thecombination of two one-portShort-Open-Load calibrations withadditional measurements of a thrustandard to complete the two-portcalibration. All of the standardsmust be fully known and specified.

The SOLT standards are reason-ably well modeled with simplelumped elements: open-circuitcapacitance, short-circuit induc-tance, load inductance, and thrudelay (and loss). Often the open-circuit capacitance will have a nega-tive value since the probe lifted inair has less tip loading than when itis in contact with a wafer.

All standards must be accuratelycontacted physically, since theinductance values are very depen-dent on probe placement on thestandard. A 1 mil (25.4 pm) longi-tudinal change in the overlap of theprobe tip over the standard canresult in a significant change in theinductance value (see Figure 16).

Typical values of calibration coef-ficients for individual microwaveprobes (Cascade Microtech ACPstyle) are shown in Table 2.

Calibration coefficients must becorrectly entered into the VNA todo any good. This would seem to

6

be an easy task with only threecoefficients per probe, but improp-er calkit entry is one of the largestsources of error in on-wafer VNAmeasurements. VNA front paneldata entry of calibration coeffi-cients can be tedious and errorprone.

Some VNA's do not allowlumped element models for allstandards. The use of an offsettransmission line based equivalentcircuit is required. The lumped ele-ment series inductance representedby the load (term) or short can beeffectively modeled by a short sec-tion of high impedance transmis-sion line offset. Normally the offsetimpedance is set to the maximumvalue allowed by the VNA (e.g.,500 ohms for the HP 8510) andthe corresponding delay is deter-mined using the equationT, = L/Z,.

For example, the 4.8 pH shortcircuit inductance of a 150 pmpitch, ground-signal-ground con-figuration ACP probe would beentered into an HP 85 10 calkit as ashort standard with a 9.6 fs (4.8pH/500 0) long 500 0 offset.

Another common source of erroris to forget to enter the delay of the

thru line standard. A 1 ps thru isnormally used with the standardprobe pitch ISS versions (PNs101-190 and 103-726) while a 4 psthru is used with the wide probepitch ISS versions (PNs 106-682and 106-683). When using longthru lines such as for PyramidProbes it is important to use anaccurate delay value and to providean accurate loss value. The offsetloss term (in GflZ/s at 1 GHz) forthe HP 85 10 is given by:

CROffsetLoss =

d&w 1, GHz c @oS 1GHz 10 log,,,(e)e

where dBloIJ = measured insertionloss at 1 GHz, 20 = thru lineimpedance (normally 50 Sz), andI= physical length of the offset.

For a standard specified by lossand delay at a particular frequency,the offset loss term calculation sim-plifies to:

where dBlo, is the measuredinsertion loss (dB) at frequency f(GHz) for a line with offset delay =Delay (ps) and offset impedance =2, (usually 50 0). For example,table 1 shows that the CPW line #6

25pH 25pH

-r20 fF

1-

delay = E = I ps

15fF

-delay = 1/2 2, c = 0.375 ps

Figure 16. Improper probe placement will impact the electrical behavior of a measured standard.Use of alignment marks to set probe separation and reasonable care in placement will providegood results.

on the general purpose membraneISS has 0.24 dB loss at 10 GHzand 28.3 ps delay so this equationgives:

OffsetLoss~, , = (230.3).(50).(0.24) = 30.9m(28.3).x6 s

An offset loss value of 31 GR/sshould provide good results for allbut the shortest of the CPW lines onthe general purpose membrane ISS.

The thru standard loss parameterentry with the requirement fornearly ideal behavior of the thrustandard is a major limitation ofthe SOLT calibration method. Anyloss behavior that doesn’t match theideal loss model built into the VNAor extra reflections, such as thosegenerated by a right-angle bendwhen connecting orthogonally ori-ented probes, will introduce signifi-cant error. Similarly, excess lengthof a thru line over the spacingbetween RF contacts on a PyramidProbe will add parasitic shunt stubsfrom the excess line length. Theseact as open circuit stubs and willspoil the accuracy of an SOLT cali-bration.

An alternative to the SOLT cali-bration is the SOLR5 (Short-Open-Load-Reciprocal) advanced calibra-tion. This algorithm is similar tothe SOLT calibration for one-portcorrections, but the redundantinformation available in the stan-

dards set allows the R standard tobe any reciprocal two-port. Noother knowledge of the standard isrequired to complete the calibra-tion.

The SOLR algorithm is verypowerful when only non-idealthru standards are available:orthogonal probes, excess thru linelength, lossy or mismatched thruline, incompatible port geometries(pitch), etc. The SOLR calibrationalgorithm is an advanced calibra-tion method and is not directlyavailable on any VNA presentlymade. SOLR requires special soft-ware - such as CascadeMicrotech’s WinCal - running ona PC communicating with theVNA via IEEE-488 instrumentbus commands. Additionally, theVNA must be one that supportstrue TRL calibration (a four sam-pler VNA) in order to be able toprovide enough information forthe SOLR algorithm to be calcu-lated. Calibration of right-angleoriented probes using SOLR isdiscussed in An SOLR Calibrationfor Accurate Measurement ofOrthogonal On- Wafer DUE.6

A standard SOLT calibration canpresent the dilemma of conflictinginformation to the VNA. Thenumber of measurements andstandard definitions exceeds thenumber of unknowns in the two-port error model. This mathemati-

cally ‘over-determined set of equa-tions will not necessarily have a sin-gle entirely self-consistent solution.The implications of this lack ofself-consistency are subtle, numer-ous, and unavoidable. One examplewould be measuring slight differ-ences between S 12 and S,, for apassive reciprocal structure. TheSOLR calibration is not over-deter-mined and is inherently self-consis-tent.

Advanced Calibration MethodsS-parameters require measure-

ments with each port of the mea-sured device stimulated in turn. Fora two-port VNA this means thatthe microwave source power mustbe switched to each port (as shownin Figure 1). When the switchchanges position the error model ofeach port changes due to the non-ideal nature of the microwave trans-fer switch, the source outputimpedance, and the response termi-nation load impedance. The fulltwo-port error model provides forthese switching terms by the use oftwo different error models - onefor forward and one for reversemeasurements. The error modelshown in Figure 17 is consistentwith the SOLT calibrationmethod.2

Normally in on-wafer measure-ments the isolation terms, E,, areneglected since they are relatively

Table 2. Air-Coplanar Probe (ACP) calibration coefficients. Coefficients for VNA calibration depend on the style and pitch of the probe, as well asthe ISS used. The wide-pitch ISS's (106-682 and 106-683) use a 4 ps long thru line while the standard-pitch ISS's (101-190 and 103-726) have a 1ps thru line.

Ex:____________-________________________________~__-_______;

ER S 21 E, :. L L L:,v Fa0 4

ED ES S S11 22A 11

b0 1 u 12

DUT

Figure 17. Signal flow graph of the full two-port VNA error model without switch corrections. Theforward stimulation case is shown, a similar error model is used for reverse stimulation.

small and are not the same for thestandards and DUT.

VNA's that have the ability tosample the energy reflected fromthe switch termination as well asthe stimulus, transmissions, andreflections - the four sampler archi-tecture - allow removal of theswitching terms from raw measure-ments. This results in a reducederror model shown in Figure 18which has error boxes at each portthat contain all of the VNA andinterconnection errors. A great dealof study has gone into variousmethods for determining the coeffi-cients of the error boxes from mea-surements of known and partiallyknown structures.

The most fundamental of thesemethods, the Thru-Reflect-Line(TRL) calibration algorithm, usesthe characteristic impedance of atransmission line as the referencefor calibration.7 Line pairs differingonly in length are measured alongwith a reflect standard that isknown only to the sign of thereflection coefficient. The linelength difference must not be near180 degrees of phase since a halfwavelength transmission line mim-ics a zero length line and providesno additional information. In prac-tice a line pair will only provideuseful results over a 20 to 160degree phase difference range.Wider bandwidths require multipleline lengths to provide suitable

phase differences over a wide fre-quency range.

The TRL calibration process itselfdoes not determine the characteris-tic impedance that is the referencefor the resulting S-parameters.Additional assumptions and mea-surements may be used to deter-mine the 2, and renormalize theS-parameters to 50 ohms (forexample). Often this is not doneand the frequency dependence andimaginary part of Z, due to lossare disregarded. At low frequencies,where the per-unit-length inductivereactance is small compared to theohmic loss, the imaginary part ofZ, can be large. This effect can bepresent at moderate frequencies(perhaps up to 1 GHz) for relative-

ly high resistance transmission linessuch as are found in thin film cir-cuits.

In the TRL calibration (and itsvariations) the thru line must befully known to allow the measure-ment reference plane to be movedto the probe tips from the mid-point of the thru line. In someapplications using on-wafer stan-dards, the DUT is effectivelyembedded at the thru mid-point,eliminating the need for this refer-ence plane change.

The TRL calibration method isnot practical for Pyramid Probesand other probe cards since it isnot possible to measure multipleline lengths with fixed spacingprobes. Even with individualprobes, the TRL calibrationrequires programmable probe posi-tioners for fully automatic calibra-tion. An alternative to the TRL isthe Line-Reflect-Match (LRM) cal-ibration. The LRM calibration usesTRL-like mathematics but uses abroadband match standard to setthe system impedance. The matchessentially acts as an infinitely long,infinitely attenuating transmissionline.

The LRM calibration standardsare suitable for fixed probe spacingand can be used with probe cardsor manual positioners. The stan-dards are a subset of those used for

DUTL‘S21

S SI&1

Figure 18. Signal flow graph of the switch corrected two-port VNA eight-term error model.Since S-parameters are ratios, only seven unknowns must be determined to complete the cali-bration.

8

SOLT since either the short oropen can be used as the reflectstandard. In on-wafer measure-ments the open standard is pre-ferred since it has no probe place-ment issues to effect repeatability.The conveniently available stan-dards combined with superior per-formance make the LRM calibra-tion a natural first step to improvedcalibration accuracy over usingSOLT.

The one weakness of the LRMcalibration is that the inductanceof the load standards becomes partof the reference impedance of themeasurements. Methods for auto-matic determination of the loadinductance and correction of mea-surements are available in CascadeMicrotech’s WinCal VNA calibra-tion software.8 These more sophis-ticated versions of the LRM cali-bration are only suitable for rela-tively small thru line lengths (a fewps) but when applicable can yieldperformance comparable to themost advanced TRL calibrations.

LRRM Calibration withAutomatic load InductanceExtraction

A variation of the Line-Reflect-Match, the Line-Reflect-Reflect-Match advanced calibration pro-vides calibration accuracy compa-rable to the NIST multiline TRLalgorithm described below withoutthe need for changing probe sepa-ration.9 The LRRM calibrationalgorithm is available only inCascade Microtech’s WinCal VNAcalibration software running on aPC.

The standards used for LRRMare identical to those used forSOLT but without the require-ment for careful specification ofthe short and open standardbehavior. In LRRM, only the thruline delay (and loss for longerlines) and d.c. resistance of oneload standard must be specified.

The inductance of the load isextracted from the redundancy ofinformation provided by the extrastandard measurements. By using

only one Match standard anyambiguity in load inductancebetween ports is eliminated, reduc-ing the sensitivity to probe place-ment since the actual value is direct-ly extracted.

LRRM provides high-perfor-mance calibrations with fixed probeseparations, allowing simple, auto-mated calibrations.

A Cascade Microtech Summitprobe station using WinCal soft-ware allows highly repeatable, fullyautomated VNA calibrations initi-ated with a single mouse click. Seethe Cascade Microtech TechnicalBrief Technique Verifies LRRMCalibrations for GaAsmeasurements. 10

NIST MultiCalResearch by the Microwave

Metrology Group of the U.S.National Institute of Standards andTechnology (NIST) has advancedthe quality of rigorous on-waferVNA measurements. The focus ofthis work has been the develop-ment and application of a multilineTRL calibration algorithm usingthe full redundancy of the stan-dards. While normal TRL selectsline pairs for applicable frequencyranges, the NIST multiline TRLuses a continuously varying opti-mally weighted average of all of thedata. 11, 12

The multiline TRL algorithm isavailable in NIST's MultiCal soft-ware. This HP BASIC programruns on HP 9000 workstations andPCs and is compatible with theHP 8510B, HP 8510C, andHP 8700 and Wiltron 360 seriesVNAs

MultiCal allows high-perfor-mance calibration using on-waferstandards and computes measure-ment error bounds from calibra-tion comparison. 13

One-Tier vs. Two TierCalibrations

In most on-wafer VNA measure-ment applications a single set ofcalibration measurements are madeto characterize everything at once:the VNA errors, cables, andprobes. There are situations whereyou may want to make this a two-step process. A two-tier calibrationis obtained by performing one cali-bration - the first tier - and fol-lowing it with a second calibration- the second tier - that uses firsttier corrected standards measure-ments. Two-tier calibrations nor-mally require special software, suchas Cascade’s WinCal, since thesefunctions are generally not provid-ed in the VNA.

The second tier actually obtainserror boxes that represent thechange in the system between twocalibrations. If a cable calibration isfollowed by a second tier probe-tipcalibration the second tier errorbox will be the network parametersof the probe (see Figure 19).

Separation of the cable cal fromthe probe details can be useful incomplex multiport situations. Theerror box S-parameters of the vari-ous probe contacts are determinedand the data mathematically cas-caded with the cable calibrationerror boxes. This technique is usedin some microwave production testsystems to reduce the number ofon-wafer standards needed formulti-port calibration.

i On-Wafer Reference Planes

VNA & CablingErrors HDUT HProbe Errors MEEt Cabling /

Coaxial Reference Planes i

Figure 19. Error boxes for a two-tier calibration. A coaxial first tier is followed by an on-wafersecond tier.

S,, 8 Ml logREF 0.0 dB

5.0 dB/A

MAG

Start 0.045000000 GHzStop 40.000000000 GHz

‘--*-++IGood Marginal Poor

Figure 20. The relative magnitudes of uncorrected measurements of a short and a load providea good check on proper system function.

Another application of second tiercalibrations is for determination oferror bounds due to measurementsystem repeatability. A first-tierprobe-tip calibration followed by asecond tier calibration at the samereference plane will determine errorboxes that represent the change inthe system due to drift between thecalibrations. This information canbe used to calculate bounds on therepeatability errors impacting mea-surements that were made betweenthe two calibrations, providing auseful measure of one of the com-ponents effecting overall measure-ment accuracy.

Measuring Cal StandardsThe process of completing the

calibration entails placement ofprobes on standards and directingthe VNA measurement system toacquire data. This process is some-what error prone since it is notalways easy to determine if the stan-dard is properly connected.

While uncalibrated or ‘raw’ S-parameters from a VNA are oftenfar from accurate they can be usedto help verify proper contact withthe calibration standards.

The raised probe open measure-ment does not have any contactissues and can be used as a refer-ence. The measurement of a shortstandard will show an ideally180 degree phase rotation of thereflection coefficient. This can beseen as a rotation of the raw dataSmith chart constellation or a shiftin a phase display occurring whenthe probe makes contact. In prac-tice second order terms in the errorbox will also somewhat impact theactual change of observed reflection,but the shift will be very noticeable

Open stub

and provide a good indication con-firming visual identification ofproper alignment and contact.

The load standard shows a signifi-cant reduction in the magnitude ofthe uncorrected reflection coeffi-cient when contact is made.Contact with a 50 ohm load willtend to reduce the Smith chart dis-play toward the ideal center dot.Particularly at higher frequenciesthe various reflections fromadapters, cable connectors, andprobes can degrade the overallreturn-loss of the load standard.The measured reflections will nor-mally be reduced by more than10 dB when the load is in contact.Smaller reductions are signs of poorsystem return loss that may requirecorrection (see Figure 20).

The thru line measurement willexhibit the matching characteristicsof the load and will also show anincrease in the raw S,, transmissionmeasurement. A few experimentswill provide a good idea of what toexpect from your fixturing andVNA measurement system.

Calibration VerificationVerification of proper calkit entry

and calibration can be demonstrat-ed by measuring a long open circuitand/or long transmission line. Thereflection coefficient of the long

Pad

Line (delay)

Figure 21. Typical on-wafer verification elements. In many cases measurement of such struc-tures will catch the most common errors and problems.

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open circuit should exhibit asmooth monotonic inward spiralon the Smith chart. Look for linearphase in the transmission responseof the long transmission line verifi-cation standard. High Q-factorinductors or capacitors can similar-ly be used for calibration verifica-tion. It is a good practice to per-form one or more of these verifica-tion methods after completingevery calibration. Some commonverification standards are shown inFigure 2 1.

Verification is particularly impor-tant with the SOLT calibration toensure both proper calkit entry andsuccessful probe calibration stan-dard measurements. Do not remea-sure a calibration standard as a veri-fication, since the only thing thatwill be revealed is the repeatabilityof the measurement. If the systemis repeatable then the remeasure-ment will simply show you whatyou entered into the calkit for thatstandard. A VNA cannot recognizethe errors caused by using incorrectimpedance standards. A load resis-tor of the wrong value will not be

recognized: it will be accepted as50 ohms. The VNA will evenaccept interchanged open and shortstandards without complaint.Either mistake can result in signifi-cant measurement error.

Measurement of calibration stan-dards for the advanced calibrationswhere standards don’t have to bespecified is useful for verification.Examining the open that is usedfor the reflect standard in a TRL orLRM calibration can identify prob-lems. After calibration, the openmay not have perfectly zero returnloss. The probe in air will actuallyhave slightly less loss then theprobe in contact with the wafer, soin addition to some negative capac-itance a small amount of returngain from the calibrated open maynormally be observed. The specificstrength of this effect is very depen-dent on the probe design and willvary for different probe types andthru losses.

GSG pads shield like CPW

GS pads fringe to the ground plane or chuck

Figure 22. GS pads fringe to the ground plane or chuck.

The thru line standard is a usefulverification element for an SOLRcalibration. Again the verificationshould reflect expected behavior sothis is only useful when the behav-ior is predictable. The SOLR cancalibrate successfully even with ahighly reactive and complicatedfrequency response of the recipro-cal thru which won’t be pre-dictable. However, a good linestandard will exhibit attenuationwith square-root of frequencydependence consistent with thin-film transmission lines. Excess linelength beyond the probe contactswill act like capacitive stubs result-ing in increased attenuation athigher frequencies. Linear trans-mission phase would be expectedunless the standard has dispersivecharacteristics.

A golden standard device - adevice that has been characterizedusing a (hopefully) known goodtest system - is another way to vali-date a test system. It is difficult tovalidate a calibration and measure-ment method this way however.Slight differences in calibration ref-erence planes may appear as signif-icant error. Differences in circuitbias or power supplies can alsoappear as errors in the microwavemeasurement. A golden standardcan, however, be an effective andsimple test for qualifying a new sys-tem that uses identical equipmentand methods as a reference mea-surement system.

GSG vs. GS PadsAfter probe positioning and gross

cal calibration errors, the most sig-nificant inaccuracy in microwaveprobing is associated with parasiticcoupling at the probe tip (see Figure22).

A ground-signal-ground interfaceterminates field lines effectively, butground-signal pads allow the openside fields to terminate on the waferground plane or prober chuck. Thedegree of this coupling is dependenton substrate thickness and padspacing.

An undesirable mode of propaga-tion where all top side conductorsform a microstrip with the bottomof the wafer can occur in thesestructures. The best way to dealwith this problem is usually to min-imize the excitation of that mode.

The GSG pad configurationexcites the undesired microstripmode less than the GS configura-tion. A reasonable rule is to avoidGS pads for good results aboveabout 10 GHz. GHz.

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GS Probing ParasiticsFigure 23 depicts a GS probe tip,

two connection pads, and a DUT,along with the main reactive para-sitics of the connection. Theground is the back side of thewafer, the wafer chuck, or a high-conductivity substrate such asSilicon.

Figure 23. GS probing parasitics.

The desired transmission line usesthe inductances in series with theDUT and the corresponding bridg-ing capacitors. The resistor repre-sents polyiron material used onCascade Microtech single lineprobes that acts to absorb energy inthe undesired mode. This attenu-ates energy propagating up theground which could resonate withthe probe holder.

Proper selection and use of amicrowave probe will allow the par-asitic CSProbe and CGProbe to besmall and repeatable. However, dif-fering pad sizes or conductorgeometries on test structures cancause CGDur or CSDut to be differ-ent from one structure to another,or from the calibration standards.

Only when these values are equalfor calibration standards and mea-surement devices will these para-sitics be removed by VNA calibra-tion. But changes in metal configu-ration will often occur. For mea-surements above 10 GHz, the GSconfiguration is inadequate and thesuperior shielding of GSG is pre-ferred.

Device Bias ConsiderationsPower supplies may be supplied

through the VNA bias ports whichcombine the DC and RF signals

Figure 24. Calibrated measurement of short circuit standard before and after distressing acable. A change has occurred in the cable electrical behavior resulting in a 20 GHz resonance.

together at the measurement portsusing bias tees. If the performanceof the VNA internal bias tee isinadequate, or your VNA does notprovide one, then an external biastee may be preferred.

A bias tee is a three-port networkwith the properties of a diplexer;that is, a wide bandwidth port iscoupled to the bias port via a low-pass filter and to the RF portthrough a high-pass filter.Important selection parameters of abias tee are: cross-over frequency,bandwidth, current and power rat-ing, resistance and loss, and con-nector type.

The series and shunt resistance ofthe bias tee will be significant whenmeasuring individual transistors orother devices where a precise biaspoint is desired. For moderate tolarge currents the series voltagedrop will make a change from theexpected bias voltage. Expecteddevice currents will vary from actu-al due to the current path providedby shunt resistance. This resistanceis high, but not high by the stan-dards of parametric measurementswith currents possibly ranging from1 fA to 100 mA. Probes, cables,and probe stations designed specifi-cally for dc parametric measure-ments are generally preferred forthese cases.

For microwave integrated circuitsthe bias and power lines will oftenuse separate pads from the RF con-nections. For these circuits, either amulti-contact probe, hybrid RF

and needle probe system, or aPyramid Probe Card is needed. Lowimpedance is the primary desirablecharacteristic in a power supply andis a major feature of the PyramidProbe Card with its bypass capaci-tors very close to the contacts andexcellent grounding.

Needles, either used in a hybridprobe card or as additional probeson separate positioners, have highinductance since they are not close-ly coupled to a ground return.Inductance on supplies can causepower supply ‘bounce’ in digital cir-cuits and instability or gain flatnessproblems in analog circuits.Adjacent needles may be inductive-ly coupled or even by virtue of shar-ing a common ground return pathmay create signal paths betweenvarious portions of a circuit by wayof power connections.

Cable SensitivityCorrecting the network analyzer

measurement through a lossy cableis similar to the correction we sub-consciously use to see an imagethrough a dirty window. We correctfor the attenuation due to dirt onthe window as well as reflectionsfrom the window’s surfaces. It isimportant that the cables be highlyrepeatable as well as reasonably lowloss, so that there is enough dynam-ic range after looking through andcorrecting for the lossy cable.

Calibrated measurements of ashort circuit standard before andafter the cable has been significantlyflexed, shown in Figure 24, demon-

SymptomMeasurements do notrepeat at any freq

Measurement noisy at allfreqs

Measurement noisy atcertain freqs

Measurement noisy athigh freqs only

DUT measurements noisyat low freqs only, orTransistor S 21

compresses at low freqs(passive elements ok)

Passive elementreflection coefficientoutside Smith chart

Table 3. Troubleshooting guide.

strates the importance of measure-ment repeatability. The resonanceat about 20 GHz would signifi-cantly distort further DUT mea-surements.

When troubleshooting theresponses of a VNA, it is useful tokeep in mind that impedances thatvary rapidly with frequency cannotbe due to errors near the DUT.When wafer probing there is insuf-ficient electrical length on thewafer or in the probes to cause res-onances or impedances that changerapidly with frequency.

TroubleshootingTable 3 may be helpful in trou-

bleshooting VNA measurementproblems (some details apply onlyto Agilent 8510 systems).

Likely ProblemCables or adapters are loose orfaulty

Analyzer fault

Probe Contact fault

Probe head fault

Oscillating DUT

Insufficient averaging onHP8510

Poor ramp sweep repeatability

Sweeper band switching

Loose connection

Faulty probe resonating

Poor return loss at high freqs

DUT is being biased by rectifiedRF, or RF is compressing DUTgm

Incorrect cal coefficients

Poor probe placement onstandards

References

Calibrate in coax with and without cables andadapters

Calibrate in coax directly at test set port

Verify probe contact visually or at dc

Calibrate in coax at probe connector; if good,replace probe head

Use different bias condition to verify; changelow-frequency DUT terminations

Increase averaging to at least 4 in rampmode, or 128 in step mode

Use step mode

Increase averaging, use step mode. Changesweep rate

Tighten connectors

Check open stub on ISS - should be smoothinward spiral

Measure ISS load with corrections off -should be at least 10 dB below the open orshort-trace back to test set using coaxialload and short

Reduce RF power level, add device input portattenuation, use power slope

Adjust cal coefficients

Use alignment marks to improve placement

1. Calibration and Accuracy FactorsSummit High-Frequency Probe StationReference Manual. Cascade Microtech,part number 103-375-A

2. D. Rytting, “Appendix to an Analysisof Vector Measurement AccuracyEnhancement Techniques,” RF &Microwave Symposium and Exhibition,Hewlett-Packard Inc, 1986.

3. D.K. Walker, D.F. Williams,“Compensation for Geometrical Variationsin Coplanar Waveguide Probe-TipCalibration, IEEE Microwave and GuidedWave” Letters, Vol.7, No.4, pp. 97-99,April 1997.

4. Paul J. van Wijnen, On theCharacterization of. . Cascade Microtech

5. A. Ferrero, “Two-Port NetworkAnalyzer Calibration Using an Unknown'Thru,'" IEEE Microwave and Guided WaveLetters, Vol. 2, No. 12, pp. 505-507, Dec.1992.

6. Hayden and Basu, "An SOLRCalibration for Accurate Measurement ofOrthogonal On-Wafer DUTs," IEEEMTT-S Digest, 1997.

7. C. Hoer, G. Engen, “Calibrating adual six-port or four-port for measuringtwo-ports with any connectors,” IEEEMTT-S Digest, pp. 665-668, 1986.

8. A. Davidson, K. Jones, E. Strid, “LRMand LRRM Calibrations with AutomaticDetermination of Load Inductance,” 36thARFTG Conference Digest, Nov. 1990.

9. J. Pence, “Verification of LRRMCalibrations with Load InductanceCompensation for CPW Measurements onGaAs Substrates,” 42nd ARFTG ConferenceDigest, Nov. 1993.

10. Cascade Microtech Technical BriefTechnique Verifies LRRM Calibrations forGaAs measurements.

11. R Marks, “A Multi-line Method ofNetwork Analyzer Calibration,” IEEETrans. on MTT, Vol. 39, No. 7, pp. 1205-1215, July 1991.

12. D. Williams, J. Pence, A. Davidson,“Comparison of On-Wafer Calibrations,"38th ARFTG Conference Digest, Dec. 1991.1.

13. For more information contact DylanWilliams, NIST 813.06, 325 Broadway,Boulder, CO 80303.

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