NASA Contractor Report 194922

Y" i J "Z "/

Switched-Beam Radiometer Front-End

Network Analysis

IL J. Trew

Case Western Reserve University, Cleveland, Ohio

G. L. Bilbro

North Carolina State University, Raleigh, North Carolina

(NASA-CR-194922) SWITCHED-BEAM

RADIOMETER FRONT-END NETWORK

ANALYSIS Final Report (North

Carolina State Univ.) 32 p

N94-35247

Unclas

Grant NAG1-943

June 1994

G3/32 0012659

National Aeronautics and

Space Administration

Langley Research CenterHampton, Virginia 23681-0001

https://ntrs.nasa.gov/search.jsp?R=19940030741 2018-09-08T13:12:29+00:00Z

Table of Contents

If.

Ill.

IV.

Introduction

Delay Line Investigation

Delay Line Implementation

Antenna Element Model

Simulation Results

Delay Line Architecture

Noise Figure and Combiner Loss

Monte Carlo Tolerance Simulation

Phase Shifters

Conclusions

Pa_a

1

3

3

4

7

8

13

13

17

27

Addendum 28

I. Introduction

Switched-beam networks offer the potential to fabricate

electronically steered antennas with RF performance characteristics

superior to mechanically steered configurations. This concept has been

widely developed and employed for phased-array radars, but has not

been thoroughly investigated for radiometer applications. The radiometer

application is, in many regards, much more difficult for switched-beam

networks than encountered in phased-array radars for a basic reason. In

a radar a coherent signal is transmitted to a target and the return signal

compared to the original to determine certain parameters of interest

about the target, such as range, velocity, etc. The active signal is the

main information carrier and since it's characteristics such as frequency,

phase, and amplitude are known, it is only necessary to determine how

the transmission path and target modify the signal to extract the

information of interest. Noise acts to contaminate the signal, but

primarily serves to place a limit upon the useful operating range of the

radar. Radiometers, conversely, operate on a fundamentally different

principle. They do not transmit a source signal, but measure the natural

radiation emissions from the target scene. As such, they measure

whatever radiation emissions are present within the antenna beamwidth.

Noise generated within the path between the target scene and radiometer

or within the antenna/network circuit will contaminate the desired

radiometer signal. Since the instrument is basically measuring noise, it

is, in general very difficult to differentiate between the path/system noise

and the desired target emissions.

The noise contamination problem can be minimized by proper

calibration procedures. In this manner it is possible to measure or

calculate the characteristics of the noise generation mechanisms between

the target scene and the instrument read-out. Once the noise

characteristics are known, they can be subtracted from the composite

signal to yield the desired target scene. As long as the unwanted signals

are deterministic it is possible to improve the instrument performance by

calibration. Unfortunately, certain signals are not deterministic and

cannot be removed from the composite signal by calibration. These

signals can prove very troubling and will present limitations to system

sensitivity. Examples of this type of signal include anything that has a

basically random performance characteristic. For example, a switch thatalternates between two positions is deterministic only if it returns to

exactly the same position every time it is activated. Since a physicalswitch will have tolerances associated with its operation, it will not have

exactly the same characteristics every time it switches. The tolerances

depend upon both mechanical and electrical parameters. The tolerance

problem ultimately will define the sensitivity of a radiometer, since smallvariations in the device impedance will produce slight impedance

discontinuitie_ that will generate noise. The noise generated from the

performance tolerances is not deterministic and cannot be calibrated outand will, therefore, place a limitation upon the radiometer sensitivity.

The problem is fundamental to the use of switched-beam networks

for radiometers. The purpose of this study is to investigate this issue.

Specifically, a particular combiner network is modeled and investigatedin order to determine its noise performance. The network consists of a

series of parallel delay lines with an ideal antenna element at one endand connected together by means of an ideal combiner. The noise

performance of a lossy combiner and the use of such an element inswitched-beam radiometer front-ends is also considered and reported.

The effects of delay line fabrication were also considered and preliminary

work directed towards investigation of the noise performance of example

phase shift elements is reported.

11. Delay Line Investigation

For this investigation a series of delay lines fabricated from

microstrip were simulated using the Hewlett-Packard Microwave Design

System. The lines were operated for a band of frequencies centered at fo -

4.3 GHz. The wavelength in free space corresponding to this frequency is

k = 2.75 inches or k = 0.229 feet. Arrays with 2 < N > 16 elements placed

on k/2 centers were simulated. The investigation considers one

dimensional angular sweeps. The angle was allowed to vary from on-axis

boresight (zero degrees) to +/- 45 °. Since the array is symmetric, it is

sufficient to consider only positive angles as the negative angles will be

simple mirror images of the positive angle results. The angle is measured

in terms of the free space distance between adjacent elements. This

length is extended out to k/2, which corresponds to an "end-fire"

orientation of 90 ° . The results are presented so that the center of the

plots corresponds to the 45 ° angular location. Narrow bandwidths of a

few percent are considered so that dispersion has negligible effects upon

the performance of the array. Although most of the work considered the

use of microstrip transmission lines, other transmission media, including

TEM transmission lines, were considered. The TEM transmission lines,

however, are considered impractical for lengths of a few inches that are of

interest for the frequency and array size investigated in this work.

Delay Line Implementation

The microstrip model used in the simulation is selected so that the

signal is attenuated at a rate of 2.5 db/foot, which is significantly lossier

than obtained with coaxial transmission lines. This is not restrictive,

however, since the simulations predict that for small arrays the

attenuation is negligible compared to other effects in the system. High

attenuation rates result in the noise figure being dependent upon the

steering angle, but this effect is not significant for losses up to about 5

db/foot. The delay lines are assumed to be switched by PIN diodes which

introduce negligible noise compared to other effects in the system.

An alternative to delay lines are electrically controlled filters with

specifically designed phase characteristics. In this type of network the

delay is established primarily by varactor diodes, which can be tuned bybias to present the correct impedance to generate the desired delay. This

operation is in contrast to the use of the varactor diodes as switches

where they simply switch lengths of transmission line in or out of thenetwork. The varactor diodes in the filter network will have reactance

characteristics that are strongly determined by the electronic bias andthe reactance tolerance will be directly dependent upon the .bias

tolerance of the source. Unless extremely stable bias sources are

employed this type of network may contribute significant noise. Theamount of noise will be a function of the diode resistance as well as the

bias source stability. This type of network has the advantage of small sizeand easy integration, however, and may be useful for the switched beam

application. Due to time limitations it was not possible to completely

investigate these networks at this time, but they should be investigatedin more detail due to the potential advantages offered.

Antenna Element Model

Several circuits were evaluated for the switched-beam radiometer

front-end network. Two general models were investigated in detail. Both

network models use arrays of delay lines with realistic loss and both use

power combiners to combine the output from the delay lines into a 50

load. The two models differed in the details of the method used to

simulate the antenna and free space propagation. One model used power

splitters to distribute the output of a source to lossless delay lines used

to model free space propagation. That model was found to not accurately

model the desired network for reasons that will be discussed. A model

that accurately simulated the desired network was constructed by use of

voltage-controlled voltage sources with specified phase. These elements

permitted the free space wave to be simulated for any incident angle by

control over the individual phase at each source. The resulting circuit for

a 4 element network is shown in Fig. 1 and was used for this

investigation.

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Fig. 1 Switched-Beam Network Model

Originally, attention was directed towards investigation of the noise

performance of the delay lines and it was desired to isolate the delay line

effects from those of the combiner. Therefore, the array was modeled with

a series of delay lines connected by an ideal power splitter on the input

and an ideal combiner on the output. The power splitter and combiner

had no losses and contributed no noise to the network. When the noise

performance of the network was calculated a very surprising result was

obtained. It was observed that the lowest noise performance was

obtained when the array was tuned to the extreme angular positions at

the +/- 45 ° positions. This result is contrary to intuition since the

extreme angle positions require longer delay lines with their

corresponding loss. The extra length of Iossy line should produce more

noise so that the highest noise is expected at the extreme angle positions.

The reasons for this performance were investigated. It was

discovered that the anomalous behavior was related to the model used in

the simulation. The power splitter on the array input does not accurately

model the performance of the desired network. In fact, the power splitter

creates a simple parallel topology of delay lines, rather than a suitable

model of a switched-beam network. When the array using the input

power splitter is tuned to the extreme angle position the delay line on one

end is tuned to maximum delay and the delay line on the other end of

the array is tuned to zero delay. The array of delay lines operates as a

group of parallel resistors and the noise performance is defined by the

net equivalent resistance. Since the end line tuned to minimum delay

has the lowest resistance it will, therefore, dominate the parallel

combination and the minimum net resistance will be obtained when the

array is tuned to extreme angle positions. Conversely, the largest net

resistance (and the greatest noise performance) is obtained when all the

delay resistors are the same, which occurs when the network is tuned to

the zero angular position. This behavior explains the observed

performance and indicates that the power splitter model is not an

accurate representation of a switched-beam input network.

A suitable model for the switched-beam network is obtained by

removal of the input power splitter and locating a voltage source on the

input of each delay line element as shown in Fig. 1. The binary power

combining strategy is also shown in Fig. 1 and demonstrated for an array

of four delay line elements. Removal of the input power splitter eliminates

the paralleling effect of the lossy lines. Each element is modeled as avoltage-controlled voltage source with a specified phase and animpedance of 50 £2 at the frequencies of interest around the center

frequency of 4.3 GHz. The phase of the n th element is

A,, = B u • I v •(n - l)

where Bu = 2rcf/c is the free space wave number which depends upon

frequency f- 4.3 x 109 and c = 9.84 x 10 s feet/sec. The attenuation

coefficient of the microstrip delay line is KL - 0.69�foot, which

corresponds to 2.5 db/foot.

Networks of varying complexity are formed by using a series of 2x 1

combiners as shown in Fig. 1. By cascading the combiners network

complexity will increase by a factor of two. That is, one combiner will

have two lines. A network of 4 lines, however, will require the use of three

2xl combiners. For this work combiners with loss are considered since

the combiner will be a significant noise source as the number of lines

increase. The attenuation of the power combiner is described in terms of

the loss of a corresponding power splitter. This loss was varied from 3.1

db (3 db of ideal combiner loss and 0.1 excess resistive loss) up to 5 db

loss (3 db ideal combiner loss and 2 db excess resistive loss) in order to

determine the effects of the loss upon overall system performance.

Simulation Results

Three different simulation experiments were performed. First, two

different delay line architectures were investigated and the noise figure

performance of each was determined. Second, the dependence of noise

figure on the loss in the power combiners was investigated. It is

demonstrated that small losses in the combiners have a limiting effect

upon the optimal size of the array. Third, the effect of uncertainties in

the microstrip delay lines were considered. The effects of line tolerances

were investigated by means of Monte Carlo simulation techniques. This

effect is very significant in establishing radiometer sensitivity since the

8

resulting noise is not deterministic and cannot be calibrated out, as

previously discussed.

Delay Line Architecture

Two delay line architectures based upon the network shown in Fig.

1 were considered. The first is an arrangement in which there is no delay

on axis and as the beam is steered to end-fire, delay is added to

compensate the free space delay according to

where Lu is the free space delay length, Bu = 2zf/c is the free space wave

number, BL = 2rtf/vt. is the wave number In the microstrip, and vL=O. 7c is

the phase velocity in the microstrip. This results in low noise figure on

axis and high noise figure off axis as shown in Fig. 2, which shows the

noise figure performance for arrays fabricated using 6 db power

combiners. The power combiners include 3 db of resistive loss. In Fig. 2

the horizontal scale is the delay line length in feet with the left hand side

indicating zero delay and the right hand side indicating +90 ° phase,

which occurs at 0.116 ft. or 0.5_. for operation at 4.3 GHz. The vertical

scale is the combiner noise figure with a maximum of 4 db indicated.

Array sizes of 1, 2, 4, and 8 elements are shown. As indicated, the noise

figure for the arrays increase with phase angle. At zero phase the arrays

have only the 3 db ideal combiner loss. As delay line length increases

more resistive loss is encountered and noise figure increases. As

expected the maximum loss and greatest noise figure is encountered with

the largest array size.

The second architecture is designed to reduce sensitivity of the

noise figure to angle. The delay line lengths on axis are selected to be the

average length value determined by

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10

and are either shortened or lengthened to steer the beam. As shown in

Fig. 3, this increases the average noise figure compared to the otherarchitecture, but essentially eliminates the angular dependence of noise

figure for the re'rays considered (i.e., N < 8).

Fig. 3 indicates that the noise figure of the second design falls

slightly with increasing steering angle even though a comparison of the

performances shown in Figs. 2 and 3 shows that the sensitivity of the

network performance to steering angle is greatly reduced in the second

design. For the first design the noise figure indicated in Fig. 2 .rises

rapidly with steering angle as expected. But for the second design with

on-axis delay, there is a weak reverse dependence upon steering angle:

the noise figure falls with angle, contrary to expectation.

For very lossy delay lines, this reverse dependence on steering

angle is clearer, as shown in Fig. 4 where the noise figure for two 8

element arrays is presented. The arrays differ in the amount of loss. One

array has a loss factor of KL = 0.69�foot and the other a loss factor of KT. =

2.5�foot. The second loss factor is much greater than expected in

practice and is shown only for illustration purposes to indicate trends. It

indicates that only for very lossy lines does the noise figure of the array

depend significantly upon the beam angle. The maximum noise figure

occurs on-axis. This result is explained in terms of simple circuit theory

for parallel reslstive circuits. However, the gain of the antenna is

correspondingly reduced for large steering angles, so that the output

signal is also lower at high angles.

For larger arrays, lossier lines, or lower frequencies, attenuation

increases and more structure is obtained in the noise figure versus angle

performance characteristics as shown in Fig. 5 for array sizes up to

N= 16. The noise figure dependence results from the excess losses in the

power combiners. The optimal number of elements with respect to noise

figure can be determined for various power combiner losses. For the

largest array (N=16) the noise figure is greatest at the zero delay line

length posiUon and decreases as delay line length is increased.

11

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13

Noise Fi.qure and Combiner Loss

The effects of combiner loss upon the network noise figure were

investigated for array sizes up to 16 elements. The results of the

simulations are shown in Fig. 5. The number of delay elements is shown

on the horizontal scale and the network noise figure is indicated on the

vertical axis. Noise figure was calculated for combiner resistive losses of

0.1 db, 1 db, 2 db, and 3 db. As indicated, the network noise figure is

always reduced as array size increases for low resistive loss values. For

example, for resistive loss of 0.1 db and array sizes up to N= 16 the noise

figure is still being reduced. As resistive loss increases an optimum array

size is found to exist, with the optimum number of elements being

reduced as loss increases. For resistive loss of 2 db the optimum array

size is 4-8 elements. For resistive loss of 3 db, however, the minimum

noise figure is obtained for the minimum array size (N=2 in this case).

Monte Carlo Tolerance Simulation

In order to investigate the sensitivity limitation imposed by

component tolerances, a Monte Carlo approach was employed.

Simulations on the on-axis delay line network were performed using

component tolerances of 1%, 2%, 3%, 4%, 5%, and 6%. The actual line

length for each angle was randomly selected to lie within the desired

value to the allowed tolerance. The noise figure for the array was then

calculated. The procedure was repeated until sufficient data was

obtained to determine the resulting spread in network noise figure. This

information is very important since the noise figure can not be resolved

to accuracy greater than the resulting spread in network noise figures.

Since this type of variation cannot be removed by calibration, it

ultimately determines the sensitivity of the radiometer.

14

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Combiners (KL = 0.69/foot, Frequency = 4.3GHz)

15

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Line Propagation Length for an 8 ElementCombiner with Line Length Tolerances of 1-6 %.

16

The results of the simulations are shown in Fig. 6. The horizontal

scale indicates delay line length in wavelength, and the vertical scale

indicates network noise figure. The noise figure is calculated for an on-

axis array of N=8 elements with a resistive combiner loss of 2 db. Thenon deterministic noise figure variation is indicated by the width (i.e.,

thickness) of the noise figure plots. The uncertainty in noise figure and

resulting noise temperature is directly related to the line length toleranceand the results are indicated in Table 1.

Table 1

Uncertainty in Network Noise Figure and Noise Temperature

due to Component Tolerance

Component Noise Figure Noise Temp.Tolerance Uncertainty Uncertainty

1%

2%

3%

4%

5O/o

0.01 db

0.025 db

0.06 db

0.1 db

0.16 db

0.67 OK

1.67 OK

4.03 OK

6.75 OK

10.88 OK

6% 0.2 db 13.67 OK

17

The significance of this result is that if the delay line lengths for this

array could not be determined for each beam angle position to an

accuracy tolerance better than the values indicated in the table, thenetwork alone would limit the radiometer sensitivity to the temperatures

indicated. The situation gets worse for greater tolerance variations. For

this reason it is very important to use delay line elements with extremely

repeatable characteristics.

rrl. Phase Shifters

The

allows the

fabricated

switching

itself, can

The most

transmission delay lines create a phase shift function that

antenna beam to be electrically steered. Phase shifters can be

in a variety of configurations, but generally require the use of

elements to change the amount of phase delay. The delay,

be generated using transmission lines or reactance devices.

common switching element used is a PIN diode. The noise

generated by the diode will vary, depending upon the state the switch is

in. The resistance of the diode will vary from a low value in the forward

bias, or 'on' state, to a very high value in the reverse bias or 'off' state.

Since the switch is fundamental to phase shift networks it is important

to understand it's noise characteristics. In this section the noise

performance of three phase shift implementations is investigated. The

phase shifters are fabricated in microstrip using a commercial PIN diode

(MA47899-030). This diode is not necessarily the best diode to use, but is

typical of devices commonly employed in phase shifters. The noise

performance of the phase shifters is typical of experimentally obtained

results.

The three phase shift implementations investigated consist of one

reflection design and two transmission type (switched-line and loaded-

line) phase shifters. The phase shifters were designed to operate at a

center frequency of 4 GHz and were fabricated in microstrip using an

alumina (AL203) substrate. The reflecUon phase shifter uses a hybrid 90 °

coupler in order to operate as a two-port device. All three phase shifters

were designed to provide one bit 45 ° phase shift.

18

The equivalent circuit for the PIN diode is shown in Fig. 7. The

equivalent circuit is applicable for both forward and reverse bias states.The equivalent circuit can be used for other PIN diodes by determination

of the equivalcnt circuit parameters, which can be established by

parameter extraction techniques from measured data. The phase shiftercircuits are shown in Figs: 8a, 8b, and 8c, for the reflection-type,

switched-line, and loaded-line phase shifters, respectively. The RF

performance of the three phase shifters are shown in Figs. 9a, 9b, and

9c, respectively, for the three phase shifters. The RF performance isshown for a frequency band extending from 3 GHz to 5 GHz. The lefthand side vertical scale indicates the return loss and insertion loss for

the phase shifters, and the right-hand side vertical scale indicates the

phase shift angle and noise figure. The RF performance characteristicsfor the three phase shifters are summarized in the following tables.

Table 2

RF Performance of the Three Phase Shifters

Phase

Shifters

Switched-

Line

Loaded-

Line

Reflection-

Type

BW(%)

20.0

11.0

6.8

RL°(db)

-34

-5O

-38

RL l{db)

-.36

-43

-32

ILo{db)

-0.193

-0.020

-0.138

IL 1 (db)

-0.194

-0.082

-0.287

19

CMP4L

o

L-Lp nH

CMP1 k 1=0

spdtk 1 0 3

kl-1

MA47892-109 CMP6EQUATIONCj=I R

EQUATION R{=0.4

EQUATION Rr=O.5

EQUATION Lint=@.3

EQUATION Cp=8.88

EQUATION Lp=O

CMP5

C

)tC=Cp pF

CMP2

R

R=Rf OH

-t-O

t,..

rv CMP3rv C

)tc-cj pF

CMP7

L

L=Lint nH

Fig. 7 PIN Diode Equivalent Circuit

2O

Table 3

Noise Figure Performance of the Three Phase Shifters

Phase Shifter

Switched-

Line

Loaded-Line

Reflection-

Type

NFo{db)

Center

0.192

0.020

0.137

NF 1 (db)

Center

0.192

0.082

NF°(db)

Max

0.204

0.024

0.284 0.142

NF 1{db)

Max

0.196

0.112

0.287

In Tables 2 and 3 the ,o, and ']' correspond to the two states of the

phase shifter bit. The percent bandwidth is defined as the range of

frequencies with phase shift error less than 10 ° and return loss less than

-20 db.

As indicated in the Figs. 9a, 9b, and 9c and summarized in the

tables it is seen that the loaded-line phase shifter has superior noise

performance over an acceptable bandwidth. The switched-line phase

shifter has a considerably larger bandwidth, but the noise figure

throughout the band is significantly higher. The reflection-type design

provides the least desirable performance ....

The offset between the noise figure values at different positions of

the phase shift angle is an important consideration. In a phased array

application the offset results in uncertainty in the noise figure of the

network and serves to degrade system sensitivity in much the same way

21

OiPlll

SUUST,-dG88G-T .... E3_.6

•._T :%66 mill _!Ct_l_)-5.6e7

H,,,I6 ed I _ um

,L TFV'4I)-O.08C2

o,p7NSl'l.

21_,,,_51_ mi,_-u35b rail

_G_

(_NP4t

_6b-9.95 EQUATION

quir tb-_. $$OeO/4e)/sqrt (9.G)/4N !80/2.54_ | HOVkbkb-l.O0 EOL_TION

[(IJ_TIOH Lshort-OS.B

EOUQTIOH Lm-B,UEI

011'1i 01PIZ

SU'BST,,dGDBE oqr$ SLJES'I",,dG66G m,4_Hl,,,v5Ob railI(2_5b millS"ll. Hl_5b m_81_ rail

L_ Impic]uartbmi I Im_Lbumil

__ _ 14,.u35bmi I H-wS_ rail

_lDST'dGl_m3Sb mH!l,-uSgb rail

L'mquurtb mi I _JDST-cIGOB5H'-_5b mi I

OqF'15

-...---..---...Ea--- ---_]

pinbk:

,u | m

EQU_TION_=I

QGROUND

$RGROUNB

_JEST=d688£ _/

L.d.Jhort @G_OUN]3

kl_ mi I

04P5|

JI,_=dGO86

C-L_hor t @Gi_OUND

Fig. 8a Reflection-Type Phase Shifter

22

mgm.

I,Br+3 m

T"H-16 mi I

HSSLmSTR_T[

5UBST-sI h i IER-S. 6 '---MUR- I _ _

CONI)-I.6E+396 r';

TAND-B.O

Switched-Line Phase Shi.Fter

o4P't7 o_!

oe'J! SUBST slBmil l lH-.i milEOUATION LI=IBB(731(20Blr_tm_ L=LI mil L.r.JEOUATION L2-IBB(BTG(2BBB H=uz milEOUATIONua=9.85

EOUATION ws,,wl ] '._:_..Imi_l] pi.i_

EOUATIONk l,,r _OUATIONk3-_|

'FLLEIUATION ,'B _'_ mmJ|

EQUATIONl,,_-l-r _ EOUATIONk4-I-r

:'"L'I :"21_-" _-"

W-vb mi I _ ot_|5UBST-s Ih i I

5UBST=zlSmi I -- -

L-L2 mi ] "_&m_t

-.. _..,, ;Zi

Fig. 8b Switched-Line Phase Shifter

23

I

HI,l=I.8[+3 m

! ,!_}"T'

H,',I B rail,l

(:NP1I,ISSUL_"R_ITE

SUBST-= IBmi 1

ER-9. G

NOR- 1COND-1. BE+3B6ROUGH-{} um

TAND-8.{}

(_lS

SUBST-: 18mi I (:_Hl=w: mi I H2-wa mi 1 I_TL

_ SUBST,,= l_i IIm,illa mi I

I1":

EQUATION k

. .@._,!

..j ul .,4 "l-

1,,1

• pinbkl]

AGROUND

Loaded-line 45'Phase ShiTter Bit

EOUATION LA= 100<2BB. B<2880

EOUATION LB= 1B8<433.0( 2888

EOUATIONXA-I< 11.61<28

EQUATION HB=(}. 1( 1.41<28

EOUATION w:-9.95

OVpllllb-rE

SUBST-= IBmiI

Nl-wa railX;_'w: rail

Jia-wa/4 rail

r

II

i _

I l

._1-1.

EOUATION ,?-I

•__ I:_

pink2

II

AGROUND

Fig. 8c Loaded-Line Phase Shifter

24

RL IL I

I

4

----] Ph.Shift NF98 I

PICI

SI

8

Fig. 9a Reflection-Type Phase Shifter Performance

25

RL IL

8 B 1 TL

"_'_ 7",,I

\/" I__i_ 4

,/

.... ! /'

/ i t

98

,Cl

5

NF

0.5

0

f(GHz)

Fig. 9b Loaded-Line Phase Shifter Performance

26

RL IL0 8

iI

NF1

f(GHz)

Fig. 9c Switched-Line Phase Shifter Performance

27

as discussed for the tolerance considerations. The switched-line phase

shifter appears to have the most desirable characteristics in this regard.

IV. Conclusions

The noise figure performance of various delay line networks

fabricated from microstrip lines with varying number of elements was

investigated using a computer simulation. The effects of resistive losses

in both the transmission lines and power combiners was considered. In

general, it is found that an optimum number of elements exists,

depending upon the resistive losses present in the network. Small

resistive losses are found to have a significant degrading effect upon the

noise figure performance of the array. Extreme stability in switching

characteristics is necessary to minimize the non deterministic noise of

the array. For example, it is found that a 6% tolerance on the delay line

lengths will produce a 0.2 db uncertainty in the noise figure which

translates into a 13.67 °K temperature uncertainty generated by the

network. If the tolerance can be held to 2% the uncertainty in noise

figure and noise temperature will be 0.025 db and 1.67 °K, respectively.

Three phase shift networks fabricated using a commercially

available PIN diode switch were investigated. Loaded-line phase shifters

are found to have desirable RF and noise characteristics and are

attractive components for use in phased-array networks.

28

Switched-Beam Radiometer Front-End Network Analysis

Addendum

After submission of this report some additional calculations were

performed. The purpose of these calculations was to investigate the

uncertainty in the noise figure and noise temperature for a combiner

structure that had lower loss than the combiner previously investigated

and reported. The original calculations are described starting on page 13

in the Monte Carlo Tolerance Simulation section. The original

calculations were performed for an N=8 combiner array with 2 db

combiner loss. "These calculations resulted in the the uncertainty values

presented in Table I on page 16. As indicated in Table 1 the noise

temperature uncertainty varies from 0.67 °K for 1% line length tolerance

to 13.67 °K for 6 % line length tolerance.

The new calculations reported in this Addendum were performed

for the identical combiner array, except that the combiner loss was

reduced to 0.5 db. The results are presented in Table 4.

As indicated in Table 4, the decrease in the combiner loss from 2

db to 0.5 db produces a slight reduction in the uncertainty in the

network noise figure and noise temperature. However, the reduction is

not in proportion to the combiner loss reduction. For example, for 1%

line length tolerance the noise temperature uncertainty is reduced from

0.67 °K to 0.58 °K, and for 6% tolerance the noise temperature

uncertainty is reduced from 13.67 °K to 12.95 °K.

These calculations indicate that the line length variation due to

the tolerances is far more significant in determination of the network

noise temperature uncertainty than is the actual magnitude of the

combiner loss. Since the uncertainty in the noise temperature cannot be

removed from the system noise temperature by calibration, it represents

a lower limit to radiometer sensitivity. It is believed that this issue is

fundamental to understanding the use of switched-beam networks in

radiometry. Although this study only considered a simple combiner

network the physical principles revealed are significant and additional

work should be performed to more clearly understand the problem.

29

Table 4

Uncertainty in Network Noise Figure and Noise Temperature due to

Component Tolerances for an N=8 Array with 0.5 db Combiner Loss

Component Noise Figure Noise Temp.Tolerance Uncertainty Uncertainty

+1%

±3%

0.008 db

0.04 db

0.58 oK

2.89 oK

±6% 0.19 db 12.95 oK

Form Approved

REPORT DOCUMENTATION PAGE OH8No070 -0,aam ,, i i..

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atherl and maintaining the data needed, and comptet=ng and rev_ew,ng the collection ot _nTormatlon .ena comments re_armng tn_s Duraen estimate or .a y o_ .... _ = __cgoallectio_Jnof information, including suggestions for reducing thts burden, to Wash=ngtOn HeadQuarters Services. ulrec_orate tot Information uperat=ons ana _epor_s, i=]_ JeT_e._un

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I. AGENCY USE ONLY (Leave blank) 2. REPORT DATE 3. REPORT TYPE AND DATES COVERED

4.TITLEAND SUBTITLE

Switched-Beam Radiometer

June 1994

Front-End Network Analysis

6.. AUTHOR(S)

R. J. Trew and G. L. Bilbro

7. PERFORMING ORGANIZATION NAME(S) AND ADDRESS(ES)

North Carolina State University

Department of Electrical and Computer Engineering

Raleigh, NC 27695-7911

9. SPONSORINGIMONITORINGAGENCYNAME(S)ANDADDRESS(ES)National Aeronautics and Space Administration

Langley Research Center

Hampton, VA 23681-0001

Contractor Report5.FUNDINGNUMBERS

G NAG1-943

WU 505-64-12-02

B. PERFORMING ORGANIZATIONREPORT NUMBER

10. SPONSORING/MONITORINGAGENCY REPORT NUMBER

NASA CR-194922

11. SUPPLEMENTARY NOTES Final ReportTrew: Case Western Reserve University, Dept. of Elect. Eng. and Applied Physics,Cleveland, OH; Bilbro: North Carolina State University, Dept. of Elect. and

Computer Eng., Raleigh, NC Langley Technical Monitor: C. P. Hearn12b. DISTRIBUTION CODE• 1

12a. DISTRIBUTION IAVAILABILITY STATEMENT

Uncl assi fied-Unl imi ted

Subject Category 32

13. ABSTRACT (Maximum 200 words)

The noise figure performance of various delay-line networks fabricated from microstrip lineswith varying number of elements was investigated using a computer simulatiorL The effects ofresistive losses in both the transmission lines and power combiners were considered. Ingeneral, it is found that an optimum number of elements exists, depending upon the resistivelosses present in the network. Small resistive losses are found to have a significant degradingeffect upon the noise figure performance of the array. Extreme stabilt W in switchingcharacteristics is necessary to minimize the non-determml.Cdc noise of the array. For ,example, it is found that a 6 percent tolerance on the delay-line lengths will produce a 0.2-dbuncertainty in the noise figure which translates into a 13.67°K temperature uncertaintygenerated by the network. If the tolerance can be held to 2 percent, the uncertainty in noisefigure and noise temperature will be 0.025 db and 1.67°K. respectively. Three phase shiftnetworks fabricated using a commercially available PIN diode switch were investigated.Loaded-line phase shifters are found to have desirable RF and noise characteristics and areattractive components for use in phased-array networks.

ii

14. SUBJECT TERMS

switched-beam, noise figure performance, microstrip, resistive

losses, phase-shift networks

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