A high efficiency pv module integrated dc dc

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011 897

A High-Efficiency PV Module-Integrated DC/DCConverter for PV Energy Harvest

in FREEDM SystemsZhigang Liang, Student Member, IEEE, Rong Guo, Member, IEEE, Jun Li, Student Member, IEEE,

and Alex Q. Huang, Fellow, IEEE

Abstract—The future renewable electric energy delivery andmanagement (FREEDM) system provides a dc interface for alter-native energy sources. As a result, photovoltaic (PV) energy can beeasily delivered through a dc/dc converter to the FREEDM system’sdc bus. The module-integrated converter (MIC) topology is a goodcandidate for a PV converter designed to work with the FREEDMsystem. This paper compares the parallel connected dc MIC struc-ture with its counterpart, the series connected MIC architecture.From the presented analysis, the parallel connected architecturewas shown to have more advantages. In this paper, a high-efficiencydual mode resonant converter topology is proposed for parallel con-nected dc MICs. This new resonant converter topology can changeresonant modes adaptively depending on the panel operation con-ditions. The converter achieves zero-voltage switching for primary-side switches and zero-current switching for secondary-side diodesfor both resonant modes. The circulation energy is minimized par-ticularly for 5–50% of the rated power level. Thus, the convertercan maintain a high efficiency for a wide input range at differentoutput power levels. This study explains the operation principle ofthe proposed converter and presents a dc gain analysis based onthe fundamental harmonic analysis method. A 240-W prototypewith an embedded maximum power point tracking controller wasbuilt to evaluate the performance of the proposed converter. Theprototype’s maximum efficiency reaches 96.5% and an efficiencyincrease of more than 10% under light load conditions is shownwhen compared with a conventional LLC resonant converter.

Index Terms—DC-DC power converters, photovoltaic systems,smart grid, solar power generation.

I. INTRODUCTION

THE global demand for electric energy has continuouslyincreased over the last few decades. Energy and the en-

vironment have become serious concerns in today’s world [1].Alternative sources of energy generation have drawn more andmore attention in recent years. Photovoltaic (PV) sources are

Manuscript received July 1, 2010; revised January 9, 2011; accepted January10, 2011. Date of current version May 13, 2011. Recommended for publicationby Associate Editor J. M. Guerrero.

Z. Liang and A. Q. Huang are with the Future Renewable Electric EnergyDelivery and Management (FREEDM) Systems Center, Department of Electri-cal and Computer Engineering, North Carolina State University, Raleigh, NC27695 USA (e-mail: [email protected]; [email protected]).

R. Guo is with the International Rectifier Rhode Island Design Center,Warwick, RI 02818 USA (e-mail: [email protected]).

J. Li is with the ABB U.S. Corporate Research Center, Raleigh, NC 27606USA (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2011.2107581

Fig. 1. Part of the FREEDM system diagram.

predicted to become the biggest contributors to electricity gen-eration among all renewable energy generation candidates by2040 [2], [3]. In 2009, almost 7.5 GW of new PV capacity wasadded worldwide and it is expected that the global installed PVcapacity could reach 10 GW in 2010 [4].

The large-scale utilization of renewable energy depends onan advanced smart grid infrastructure where the users have theability to manage their energy consumption as well as use plug-and-generate and plug-and-store energy devices at home andin industrial applications [5], [6]. The future renewable electricenergy delivery and management (FREEDM) system is an in-telligent electric power grid integrating highly distributed andscalable alternative generating sources and storage with exist-ing power systems to facilitate a renewable energy-based soci-ety [5]. The 400-V dc bus in the FREEDM system provides analternative interface for PV converters. Fig. 1 shows part of theFREEDM system including an Intelligent Energy Management(IEM) module. As a result, PV converters in a FREEDM sys-tem only need to have a dc/dc stage to interface with the dc bus.Generally, this structure has several advantages.

1) Since the solid state transformer (SST) is the componentinterfacing with electric grid, the PV converters’ controllerdoes not require a phase locked loop, current regulator, oranti-islanding controller. Thus, the control task becomesmuch simpler.

2) The PV converter can be comprised of a single powerstage.

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898 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 3, MARCH 2011

Fig. 2. Two types of dc MIC structure: (a) parallel connection and (b) seriesconnection.

Therefore, it is very possible to reduce the system cost for endusers. At present, significant research effort has been made toimprove the performance of PV converters [7]–[9]; PV module-integrated converters (MICs) are gaining increasing amounts ofattention due to their distinctive features [10]–[20].

1) The MIC is an integrated part of the PV panel. MICs re-move losses due to the mismatch between panels and sup-port panel level maximum power point tracking (MPPT).For a string inverter or a centralized inverter, a string ormultistring of PV panels shares a single MPPT controller,but the mismatch loss is serious in partial shading condi-tions [21]. Considering the mismatch loss together withthe dc/ac conversion loss contributing to the whole PVsystem loss, string/centralized inverters may have lowersystem efficiency than MICs due to higher mismatch lossalthough they usually have higher dc/ac conversion effi-ciency than MICs.

2) Panel level hot-spot risk is removed [11] and panel life-time can be improved. Hot spot takes place when a shadedcell within a partially shaded panel becomes reverse bi-ased and dissipates power in the form of heat [22]. Forseries connected PV panels used with a string/centralizedinverter, a by-pass diode is added to each panel in practice.For the MIC solution, the by-pass diode is not necessarybecause each panel has its own MIC, leading to no directconnection between PV panels.

3) Its “plug and play” feature simplifies system installation.In summary, the MIC solution allows for more flexible PV

project planning and multifacet PV panel installation.

II. COMPARISON OF MICS IN SERIES AND

PARALLEL CONNECTIONS

Both dc MICs and ac MICs are available in the market. Onlydc MICs will be discussed in this paper, as they are suitable forthe FREEDM system. As shown in Fig. 2, dc MICs have twokinds of connection structures. Fig. 2(a) shows a type I dc MICconfiguration, consisting of multiple parallel connected MICsdirectly interfaced with a dc bus. Type II dc MICs, shown inFig. 2(b), need to form a series connection to obtain a voltagehigh enough for interfacing with the dc bus. Generally, the powerrating of both types of dc MICs is around 200 W–300 W.

The two system structures have different features. Table Isummarizes the comparison results of the two MIC structures:the parallel connection is more flexible due to its stronger an-

tipartial cloud capability, and the fact that any single failure ofan MIC will not impact any other part of the system. As a result,MICs in a parallel configuration have higher fault tolerance andreliability that make them more promising for PV application ina FREEDM system. However, the high gain requirement usuallycompromises its efficiency.

The topologies suitable for this application can be categorizedinto two groups: nonisolated topologies and isolated topologies.For nonisolated topologies, boost, buck–boost, zeta, cuk, or theirderivatives [23]–[32] are commonly used. Isolated topologiesmainly include flyback [33]–[39], current-fed push–pull [40],[41], and resonant converters [42], [43]. The typical maximumefficiency of these converters is around 80–97% [10]–[12], [19].

Among these topologies, the half-bridge LLC resonant con-verter is a good candidate due to its several unique advan-tages [44]–[46]. However, it is difficult for an LLC resonantconverter to maintain high efficiency for a wide input range un-der different load conditions. In this paper, a new resonant dc/dcconverter with dual operation modes is proposed. By chang-ing operation modes adaptively according to VPV and PPV , theconverter’s efficiency is improved.

III. OPERATION PRINCIPLE OF THE NEW

RESONANT CONVERTER

Fig. 3 shows a circuit diagram of the proposed resonant con-verter. S1 and S2 are two power MOSFETs; DS1 , CS1 andDS2 , CS2 are the body diodes and parasitic capacitances of S1and S2 , respectively. Cr is the resonant capacitor; Lr and Lm

are the magnetizing inductance of transformers Tx2 and Tx1 ,respectively. Llkg is the sum of the leakage inductance of Tx1and Tx2 . D1 , D2 and Co1 , Co2 form a voltage doubler at thesecondary side of Tx1 . A half-wave rectifier (HWR) formed byD3 , S3 , D4 , and CO3 is added to the secondary side of trans-former Tx2 . Diode D3 blocks the conductive path of the bodydiode of S3 . Thus, D3 and S3 form a unidirectional switch to en-able or disable the HWR. When the HWR is enabled, the HWRand voltage doubler will support the 400-V dc bus with theirsummed outputs. Table II summarizes the operation modes forthe proposed converter and Vth is a predefined threshold volt-age that is usually equal to the nominal voltage Vnom . For thefirst three operation conditions listed in Table II, the HWR isdisabled by turning off switch S3 . As a result, the converterbehaves like a traditional LLC resonant converter with a voltagedoubler [46]: an equivalent resonant inductor L′

r , comprised ofLr and Llkg , participates in the resonant circuit formed by Lm

and Cr . Diode D4 is conducting to provide a path for the loadcurrent. Once VPV is smaller than Vth and PPV is lower than50% of the rated power (Prated ), the PV panel is working undercondition #4 and the converter will operate in Mode II.

For one switching period, the operation of the converter inMode II can be divided into nine stages. The equivalent circuitfor each stage is shown in Fig. 4 and its key waveforms aredepicted in Fig. 5. For the description of circuit operation (andfor the subsequent dc gain derivation in the next section), thefollowing assumptions are made.

LIANG et al.: HIGH-EFFICIENCY PV MODULE-INTEGRATED DC/DC CONVERTER FOR PV ENERGY HARVEST IN FREEDM SYSTEMS 899

TABLE ICOMPARISON OF TWO TYPES OF DC MIC STRUCTURE

Fig. 3. Circuit diagram of the proposed resonant converter.

TABLE IISUMMARY OF OPERATION MODES FOR THE PROPOSED RESONANT CONVERTER

1) All the components are ideal. The body diodes and par-asitic capacitance of S1 and S2 have been taken into ac-count. The output capacitors have equal values (Co1 =Co2 = Co3).

2) Inductor Llkg includes the leakage inductance of TX 1 andTX 2 ; it also includes the wire parasitic inductance.

3) The turn ratio NT X 2 (Npri: Nsec) of transformer TX 2 isthe half of NT X 1 . Define NT X 2 = 1/2 NT X 1 = N .

The operation processes of Mode II are specified as follows.Stage 1 (t0–t1): When S2 is turned off at t = t0 , stage 1 be-

gins. Since Ipri is negative, capacitor Cs2 (Cs1) will be charged(discharged) and the switching node voltage Vsw will increaseaccordingly. Inductors Lm , Lr , and Llkg are all in resonancewith Cr . Vcr continues to decrease and no current flows throughthe secondary side of either transformer. The output capacitorsCo1 , Co2 together with Co3 supply the load current and VC o1–VC o3 all decrease in this period.

Stage 2 (t1–t2): At time t = t1 , Vsw reaches Vpv . Ds1 isforward biased and starts to conduct a current Ipri . Ipri startsto decrease. Once Ipri becomes smaller than the magnetizingcurrents ILr and ILm , the resonance of [Lm , Lr , Llkg ] and Cr


Fig. 4. Equivalent circuits for each operation stage (Mode II operation).

is stopped. Lr and Lm will be out of the resonance followingthis. The difference between Ipri and ILm will flow in the sec-ondary side of Tx1 . Similarly, the secondary side of Tx2 willconduct the current difference between Ipri and ILr . Thus, thevoltage across the primary side of Tx1 and Tx2 is clamped byVout . ILr and ILm start to decrease linearly.

Stage 3 (t2–t4): This stage begins when S1 is turned on at t =t2 . At this moment, the primary-side current Ipri is negative andflows through the body diode of S1 . Thus, ZVS turn on of S1can be achieved at t2 . The current Ipri continues to decrease and

changes its direction at t = t3 . The leakage inductor Llkg stillresonates with Cr , and Ipri keeps increasing. The magnetizingcurrents ILr and ILm continue to increase with the same slopeas in Mode 2. The rectifier diodes D1 and D3 conduct currentand power is delivered to the load. This stage ends when Ipri isequal to ILm .

Stage 4 (t4–t5): At t = t4 , Ipri and ILm are equal. The outputcurrent of the transformer Tx1 reaches zero. Transformer Tx1’ssecondary voltage is lower than the output voltage. The outputis separated from transformer Tx1 . Meanwhile, since Ipri is still


Fig. 5. Key waveforms of the proposed converter (Mode II operation).

larger than ILr , the output current of Tx2 is not zero and poweris delivered to the load through Tx2 . During this stage, Lm

participates into the resonance again and the resonance between[Llkg , Lm ] and Cr begins.

Stage 5 (t5–t6): Switch S1 is turned off at t = t5 . The currentIpri is positive and switching node voltage will decrease due tocharging (discharging) of Cs1 (Cs2).

Stage 6 (t6–t7): At time t = t6 , Vsw drops to zero that causesthe conduction of the body diode Ds2 . With the drop of Vsw , thevoltage applied to Lm (VLm ) decreases to zero and continues tobecome more negative. Once VLm is higher than a certain level,diode D2 on the secondary side of Tx1 will be forward biased.Thus, the voltage applied to Lm is clamped and ILm will droplinearly. Lm is out of resonance with Cr . Instead, only Llkgresonates with Cr and Ipri decreases steeply. This stage endswhen ILr is equal to Ipri .

Stage 7 (t7–t8): At time t = t7 , ILr is equal to Ipri; nomore current will flow in the secondary side of Tx2 . The outputis separated from Tx2 . D3 is turned off with ZCS. The voltageapplied to Lr is not clamped and Lr participates in the resonanceagain with Cr and Llkg . The current Ipri is positive and continuesto flow through Ds2 , which creates the ZVS condition for S2 ifS2 is turned on at this moment.

Stage 8 (t8–t10): At t = t8 , S2 is turned on with ZVS. Thecurrent Ipri continues to decrease due to the resonance between[Lr , Llkg ] and Cr . The transformer Tx1 delivers power to theoutput. This stage ends when current Ipri = ILm .

Stage 9 (t10–t11): At t = t9 , Ipri = ILm . No more current willflow in the secondary side of Tx1 . The voltage applied to Lm

is not clamped anymore and Lm participates in the resonanceagain with Lr , Llkg , and Cr . At t = t11 , S2 is turned off and anew switching cycle begins.

From the aforementioned analysis, the energy transferred byTx1 and Tx2 is different. The positive and negative parts of thecurrent Ipri are not symmetrical. However, the voltage-second

balance of the transformers Tx1 and Tx2 has still been preserved.Further, if a full-wave rectifier (FWR) is added instead of theHWR, Ipri will become symmetrical and the other character-istics of the converter will remain. The theoretical analysis ofthe aforementioned Mode II operation has been verified by thesimulation with Simetrix. Fig. 6 shows the simulation resultsof the proposed converter with following operation conditions:Vpv = 22 V, Vout = 400 V, Pout = 120 W (50% of Prated ), fs =83 kHz.

IV. DC GAIN ANALYSIS FOR THE PROPOSED CONVERTER

OPERATION IN MODE II

Understanding of the dc gain characteristic for a resonant con-verter has equal importance as knowing its operation principle.Since the dc gain characteristic for Mode I operation is the sameas LLC resonant converter, only Mode II operation requires anew analysis to be developed. The fundamental harmonic analy-sis (FHA) method is widely used for dc gain analysis of resonantconverters [47]–[50] and it is also valid for the analysis devel-oped in this paper. This approach is based on the assumptionthat the power transfer from the source to the load through theresonant tank is almost completely dependent on the fundamen-tal harmonic of the Fourier expansion of the currents and thevoltage involved. The voltage at the input of the two rectifiersVosq (t) can be expressed as

Vosq(t) = Vab(t) + Vcd(t) (1)

where Vab (t) and Vcd (t) are the secondary-side terminal volt-ages of transformers TX 2 and TX 1 (see Fig. 3). Like the con-ventional LLC resonant converter, the current in the secondaryside is quasi-sinusoidal and the voltage Vosq (t) reverses whenthe current becomes zero. Therefore, Vosq (t) is an alternativesquare wave in phase with the rectifier current. The Fourierexpression of Vosq (t) is

Vosq(t) =4π

Vout

∑

n=1,3,5,...

1n

sin(n2πfsw t). (2)

For convenience, the phase angle of Vosq (t) is assumed to bezero in (2). Its fundamental component Vo FHA (t) is

Vo FHA(t) =4π

Vout sin(2πfsw t). (3)

The rms amplitude of Vo FHA (t) is

Vo FHA =2√

2π

Vout . (4)

Define the fundamental part of the rectifier current to be

irect(t) =√

2Irect sin(2πfsw t). (5)

The phase angle of Irect is also zero since it is in phase withVo FHA (t). Thus, the average value of Iout can be calculated as

Iout =2

TSW

∫ T S W2

0irect(t)dt =

2√

2Irect

π. (6)

Iout can be expressed as

Iout =Vout

Rout. (7)


Fig. 6. Simulation results of the proposed converter operating in Mode II.

Fig. 7. Equivalent FHA resonant circuit model for the proposed converter operation in Mode II.

Equation (8) can be derived by combining (6) and (7) asfollows:

Irect =√

2πVout

4Rout. (8)

Insert (8) into (5)

irect(t) =√

2

(√2πVout

4Rout

)sin(2πfsw t) =

πVout

2Routsin(2πfsw t).

(9)The equivalent ac output impedance Ro ac can be derived by

combining (4) and (8) as follows:

Ro ac =Vo FHA

Irect=

8Rout

π2 . (10)

The expression for Ro ac is the same as the one for a conven-tional LLC resonant converter. With the known Ro ac , the equiv-alent FHA resonant circuit model can be obtained, as shown inFig. 7.

In this model, Vi FHA is the rms value of the fundamentalcomponent of the voltage at the switching node SW (VSW ). Thevoltage VSW is generated by the controlled switches S1 and S2 .The output current Iout is produced from Irect after the rectifiernetwork and filter capacitors. From a turn ratio perspective, theconversion gain of a transformer with turn ratio 2N followedby a voltage doubler is equal to a transformer with turn ratio N .Therefore, transformer Tx1 together with voltage doubler can be

substituted by an equivalent transformer Txe with turn ratio N .The resulting expression for the dc gain of the converter can bederived through a circuit analysis based on the model in Fig. 7.

Define the dc gain

M =NVo FHA

Vi FHA. (11)

Consider

VSW (t) =Vdc

2+

2π

Vdc

∑

n=1,3,5,...

1n

sin(n2πfsw t). (12)

vi FHA (t) is the fundamental part of VSW (t)

vi FHA(t) =2π

Vdc sin(2πfsw t). (13)

Vi FHA can be derived as follows:

Vi FHA(t) =√

2π

Vdc . (14)

Combining with (4), (11), and (14), the input-to-output volt-age conversion ratio is

Vout

Vdc=

12N

|M | . (15)


From the FHA model, Zout is the impedance seen from theprimary side of the two transformers

Zout =N 2Ro ac · Lmr · SN 2Ro ac + Lmr · S

(16)

where Lmr = Lm + Lr . The dc gain M can be derived asfollows:

M(S) =Zout

(1/S · Cr ) + S · Llkg + Zout. (17)

By substituting S = j2πfSW , the amplitude of M (S) is, asshown (18), at the bottom of this page.

For convenience, (18) can be rewritten as

M(fn ) =1√

(1 + λ − (λ/f 2n ))2 + Q2(fn − (1/fn ))2

. (19)

The parameters in (19) are defined as follows:

fr =1

2π√

Llkg · Cr

(20)

Q =Z0

N 2 · Ro ac(21)

λ =Llkg

Lm + Lr(22)

Z0 =√

Llkg

Cr(23)

fn =fSW

fr. (24)

Equations (19)–(24) reveal the dc gain characteristics forMode II operation. It is interesting that Mode II operation hassimilar dc gain expression to Mode I but with different parame-ters for the resonant tank. A series of example of dc gain curvesof Mode II operation under different load conditions (with differ-ent Q values) are plotted in Fig. 8. For very light load conditions(small Q), the gain has a large peak. On the contrary, the gainbecomes flat under heavy load conditions (large Q). Similar toan LLC converter, the dc characteristic of Mode II operationcan be divided into ZVS and ZCS regions, and the convertershould be prevented from entering the ZCS region. With properchoice of the resonant tank, Mode II operation can stay in theZVS region for Vpv and Ppv variations. The ZVS region can befurther divided into regions I and II due to slightly operationdifferences. In practical designs, the converter has unity gain atVpv = Vnom and the converter enters Mode II operation onlywhen Vpv ≤ Vnom . Therefore, it is impossible for the proposedresonant converter to work in region I after entering Mode IIoperation. Mode II operation can only be active in region II.Furthermore, the discussion about Mode II operation in the lastsection is dedicated for region II. On the contrary, Mode I op-eration can only be active in region I (see Fig. 9) because therequired dc gain should be lower than 1 in Mode I (Vpv > Vnom ).

Fig. 8. Series of example of dc gain curves of a new resonant converter withdifferent Q value (Mode II).

Fig. 9. Series of example of dc gain curves for a new resonant converter withdifferent Q value (Mode I).

V. DC GAIN VERIFICATION AND COMPARISON

To verify the dc gain expression derived in section IV, aseries of simulations have been performed for different Vpv fora given load condition. The converter’s switching frequency fs

is recorded. Equation (19) is used to calculate the dc gain resultat a given fs for the same operation condition. Through thecomparison between the dc gain from simulation (Msimulation )and the theoretical analysis result (Mcalculation), the accuracyof (19) can be evaluated. Table III shows the comparison resultsfor a 50% load condition where Msimulation is defined by

Msimulation =Vout · NVpv/2

. (25)

From Table III, Mcalculation matches with Msimulation verywell. Therefore, (19) is accurate enough for engineering designof the proposed converter. Furthermore, a comparison of the dcgain between Mode I and II operations is conducted in order toreveal the general dc gain features of the proposed converter.

M =32π2 ·Cr ·Lmr ·Rout ·f 2

SW ·N 2√

(32π2 ·Cr ·Lmr ·Rout ·f 2SW ·N 2−8·Rout ·N 2+32·π2 ·Cr ·Llkg ·Rout ·f 2

SW ·N 2)2+(−2·π3 ·Lmr ·fSW +8·π5 ·Cr ·Lmr ·Llkg ·f 3SW )2

.

(18)


TABLE IIIDC GAIN COMPARISON BETWEEN SIMULATION AND CALCULATION

The normalized frequency fn has a different base for Mode Iand II operations since they have different fr :

fn ModeI =fSW

fr ModeI,

where fr ModeI =1

2π√

(Llkg + Lr ) · Cr

(26)

fn ModeII =fSW

fr ModeII, where fr ModeII =

12π

√Llkg · Cr

.

(27)

For further analysis, fn needs to be unified using the samebase, for fn ModeI :

fn ModeI =fSW

fr ModeII· fr ModeII

fr ModeI

= fn ModeII ·fr ModeII

fr ModeI= α · fn ModeII . (28)

Both the dc gain expressions for Modes I and II can be writtenas functions of fn ModeII , as shown (29) and (30), at the bottomof this page.

Table IV gives the resonant tank parameters for example de-sign. For comparison, the equations for calculating several keyparameters are also listed in Table IV. The gain curves for thetwo operation modes can be plotted in the same figure, as shownin Fig. 10.

From Fig. 10, the two curves reach their peaks at the samefrequency fn M defined by

fn M =fM

fn ModeII=

12π

√(Lr+ Llkg+ Lm ) · Cr · fn ModeII

.

(31)Similar to the LLC resonant converter, operation in the region

where fn < fn M is forbidden. In the region fn M < fn <f0 , MModeI is always higher than MModeII . On the contrary,MModeI becomes lower than MModeII in region fn > f0 . Fora desired dc gain in the latter region, the following conclusioncan be drawn.

1) Mode II operation needs a higher switching frequencythan Mode I operation.

TABLE IVLIST OF PARAMETERS OF THE PROPOSED CONVERTER FOR GAIN ANALYSIS

Fig. 10. DC gain comparison between Modes I and II at 50% rated power.

2) The frequency difference becomes larger with higher inputvoltage. Fig. 10 takes Vpv = 22 V and Vpv = 32 V as ex-amples. It shows the switching frequency almost doublesif the converter operates in Mode II with 32-V input.

3) The gain curve of Mode II becomes much flatter at highfrequency. The gain is almost constant and stops decreas-ing. Considering that higher Vmpp requires smaller dcgain, this implies that the PV panel voltage may be out ofregulation in Mode II when Vmpp is too high. Therefore,it is reasonable to keep the converter operating in Mode Iwhen Vmpp is higher than a certain value.

MModeI(fn ModeII) =1√

(1 + λModeI − (λModeI/(α · fn M o d e I I )2))2 + Q2ModeI(α · fn ModeII − (1/α · fn M o d e I I ))2

(29)

MModeII(fn ModeII) =1√

(1 + λModeII − (λModeII/f 2n M o d e I I

))2 + Q2ModeII(fn M o d e I I − (1/fn M o d e I I ))2

. (30)


TABLE VCIRCUIT PARAMETERS FOR EXPERIMENT

Fig. 11. Efficiency improvement of the proposed converter in Mode IIoperation.

VI. DESIGN EXAMPLE AND EFFICIENCY ANALYSIS

The MIC will be operated with PV panels that normally haveVmpp of around 22–40 V. Vnom for this design is 32 V andPrated is equal to 240 W. The transformer primary side is thelow-voltage side and it has high resonant current circulating.In order to minimize the conduction loss, a 75-V MOSFETwith low Rdson is preferred and multistrand Litz wire should beused to reduce the ac resistance of the primary winding of thetransformer. There is no strict limitation on volume and size forMICs. Thus, a lower switching frequency fs (<200 kHz) canbe adopted to benefit the converter efficiency.

Table V gives component parameters for the MIC prototype.The threshold voltage Vth for operation mode decision is chosento be equal to Vnom . One can design Cr , Lr , Lm , and Tx1with a conventional design procedure for an LLC converter.Then, a secondary winding is added to Lr such that it formsthe transformer Tx2 . The devices D3 , D4 , and S3 in HWRhave the same current rating as D1 and D2 in voltage doubler.Considering that a practical transformer has a certain leakageinductance, the value of Llkg can be chosen to be 5–15% of(Lr + Lm ).

A comprehensive loss analysis has been conducted to eval-uate the efficiency of the designed converter. For comparison,the efficiency of a traditional LLC resonant converter with thesame circuit parameters is also analyzed. Their efficiency dif-ference is plotted in Fig. 11 for 5–50% of Prated . The efficiency

TABLE VILOSS BREAKDOWN OF THE PROPOSED CONVERTER IN MODE II WITH 10% OF

Prated (Vpv ≤ 32 V)

TABLE VIILOSS BREAKDOWN OF THE LLC CONVERTER WITH 10% OF Prated

(Vpv ≤ 32 V)

Fig. 12. System diagram for the experiment with a work flow chart for thedc/dc controller.

Fig. 13. Picture of a 240-W MIC prototype.


Fig. 14. Waveforms of an MIC prototype: (a) Mode I (ch1: 10 V/div; ch4: 10 A/div; t = 4 μs) and (b) Mode II (ch1: 50 V/div; ch2: 200 V/div; ch3: 1 A/div;ch4: 10 A/div).

Fig. 15. Waveforms to verify the ZVS operation in Mode II (ch1: 10 V/div; ch2: 20 V/div; ch4: 10 A/div). (a) Vin = 22 V, 20% of Prated (verify upper sideswitch ZVS) and (b) Vin = 22 V, 20% of Prated (verify lower side switch ZVS).

improvement drops when Ppv increases. When Ppv approaches50% of Prated , the efficiency improvement is reduced to almostzero. Therefore, there is no benefit to keep converter runningin Mode II when Ppv > 50% of Prated and mode change isrequired.

To get a better understanding of the efficiency improvementin Mode II operation, a loss breakdown is conducted for bothMode II operation and normal LLC operation with Vpv < 32 Vand Ppv = 10% of Prated . Tables VI and VII give the analysisresults. As discussed in the previous section, Mode II operationwill increase the switching frequency. Thus, the switching lossof MOSFET may increase due to the increase in the number ofswitching events. However, the data in Table VI show a signifi-cant decrease in the total switching loss. This is because higherfrequency operation leads to a much lower resonant currentthrough the MOSFET during its turn-off event. Due to the samereason, the MOSFET conduction loss and transformer copperloss are also greatly reduced. Moreover, the higher frequency

operation reduces the transformer core loss by causing smallervariation of the magnetic field strength in a switching period.As a result, the total loss is dramatically reduced by Mode IIoperation.

VII. EXPERIMENTAL RESULTS

An experimental prototype has been built to verify the per-formance of the proposed converter. Fig. 12 depicts the systemdiagram for experiment and Fig. 13 shows a picture of the pro-totype. An MPPT controller implemented in a microcontrollerwill provide a reference voltage Vpv ref that will be used by thedc/dc controller to determine the converter’s operation modebased on the criteria described in Table II. The dc/dc controllerwill check Vpv and Ppv every few minutes and its operationfollows the work flow chart in Fig. 12.

Fig. 14 shows the operation waveforms of MIC prototype inModes I and II. In Mode II, only the positive part of current


Fig. 16. Measured efficiency improvements with HWR (Mode II) for 5–50%of Prated (Vpv ≤ 32 V).

Fig. 17. Efficiency measurement results for the designed MIC prototype.

Isec TX1 will flow through the HWR. Since Isec TX1 returns tozero before each half cycle ends, ZCS turn off of diodes D1–D3is realized. Fig. 15 verifies the ZVS feature of the proposedconverter for Mode II operation. It clearly shows that ZVS turnon is achieved for both the high-side and the low-side MOSFETsin Mode II.

The efficiency of the proposed converter under different Vpvand Ppv is measured in the experiment. For comparison, the effi-ciency of operation without the HWR (normal LLC operation) isalso recorded. Fig. 16 shows the efficiency difference betweenMode II operation and normal LLC operation. From Fig. 16,the maximum efficiency improvement happens at 5% of Pratedfor all input conditions. For this condition, over 10% improve-ment is achieved. With an increase of the load, the efficiencyimprovement drops. Fig. 17 gives the complete efficiency datafor the MIC prototype. A high efficiency of 96.5% occurs inMode II with Vpv = 32 V and Ppv = 50% of Prated . The highestweighted efficiency is 95.8% in the experiment.

VIII. CONCLUSION AND FUTURE WORK

The PV converters can take advantage of the 400-V dc busin FREEDM systems to reduce its complexity as well as coststo the end user. The parallel connected dc MICs are good can-didates for this application. In this paper, a high-efficiency dualmode resonant converter topology is proposed for dc MICs. Thenew resonant converter can change resonant modes adaptivelydepending on the PV panel operation conditions. A detailedtheoretical analysis of the converter operation and its dc gainfeatures is presented in this paper. The analysis and the new

converter’s performance have been validated by the experimentresults from a 240-W prototype. Future work includes the com-pletion of an advanced energy controller design for the MIC thatcan receive commands from the IEM and allows for a flexiblecontrol of the power generation profile.

ACKNOWLEDGMENT

The authors would like to thank Edward Van Brunt’s helpduring the manuscript revision. This work made use of ERCshared facilities supported by the National Science Foundationunder Award Number EEC-0812121.

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Zhigang Liang (S’10) was born in Sichuan, China,in 1981. He received the B.S. and M.S. degreesin electrical engineering from Zhejiang University,Hangzhou, China, in 2003 and 2006, respectively.He is currently working toward the Ph.D. degreein the Future Renewable Electric Energy Deliveryand Management (FREEDM) Systems Center, NorthCarolina State University, Raleigh.

From 2006 to 2007, he was a System Engineer withMonolithic Power Systems (MPS), Inc., Hangzhou,China. His research interests include high-efficiency

power conversion, micro inverters and MICs for Photovoltaic applications, andenergy management in dc microgrid.

Rong Guo (M’10) was born in Hunan, China, in1982. She received the B.S. degree in electrical engi-neering and automation from Xi’an Jiaotong Univer-sity, Xi’an, China, in 2003, the M.S. degree in powerelectronics from Zhejiang University, Hangzhou,China, in 2006, and the Ph.D. degree in electricalengineering from North Carolina State University,Raleigh, in 2010.

She is currently an Application Engineer at theRhode Island IC Design Center, International Rectier,Warwick, RI, engaged on the denition and applica-

tion of multiphase dc/dc converter ICs for servers and desktop computers. Herresearch interests include high-frequency power conversion, analog IC design,and lighting technology.


Jun Li (S’07) was born in Liaoning, China, in 1981.He received the B.S. degree in automation fromTianjin University, Tianjin, China, in 2004, the M.S.degree in power electronics from Zhejiang Univer-sity, Hangzhou, China, in 2006, and the Ph.D. degreein power electronics from North Carolina State Uni-versity, Raleigh, in 2010.

He is currently a Senior R&D Engineer in ABBU.S. Corporate Research Center, Raleigh, NC. Hisresearch interests include topology and control ofhigh-power multilevel converters for MV drives and

renewable energy generation.

Alex Q. Huang (S’91–M’94–SM’96–F’05) receivedthe B.Sc. degree in electrical engineering fromZhejiang University, Hangzhou, China, in 1983, theM.Sc. degree in electrical engineering from theChengdu Institute of Radio Engineering, Chengdu,China, in 1986, and the Ph.D. degree fromCambridge University, Cambridge, U.K., in 1992.

From 1994 to 2004, he was a Professor with theCenter for Power Electronics Systems, Virginia Poly-technic Institute and State University, Blacksburg.Since 2004, he has been a Professor of Electrical

Engineering with North Carolina State University (NCSU), Raleigh, and theDirector of NCSU’s Semiconductor Power Electronics Center. He is also theProgress Energy Distinguished Professor and the Director of the new NationalScience Foundation’s Engineering Research Center for Future Renewable Elec-tric Energy Delivery and Management Systems, Department of Electrical andComputer Engineering, North Carolina State University, Raleigh. His researchareas are power management, emerging applications of power electronics, andpower semiconductor devices. He has published more than 200 papers in jour-nals and conference proceedings, and holds 14 U.S. patents.

Prof. Huang is the recipient of the NSF CAREER Award and the prestigiousR&D 100 Award.

Technology

A high efficiency pv module integrated dc dc