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Effects of conductor surface condition on signalintegrity

Martin BayesShipley Company L.L.C., Marlborough, Massachusetts, USA and Rohm and Haas

Electronic Materials L.L.C., Marlborough, Massachusetts, USA

Al HornRogers Corporation, Rogers, Connecticut, USA

Introduction

Understanding the impact of printed wiring board (PWB)material properties on electrical performance at highfrequencies has become increasingly important, assemiconductor device rise times have fallen and systemspeeds have risen (Bayes, 2003).

In order to create correct functioning systems withoutextensive prototyping, the influence of design rules andmaterial choices on system signal integrity must be assessedat the design stage. A selection of electromagnetic modelingtools [1,2] is available to PWB designers, allowing suchsimulations.

In order to use electromagnetic modeling tools, a varietyof system design attributes must be entered into the model.

These include both geometric parameters such as conductordimensions (trace widths, heights and lengths) and dielectricspacing, and also material electrical properties such asdielectric constant, dissipation factor and conductivity of theconductor material.

There are a number of methods for measurement ofdielectric electrical properties over the ranges of frequenciesrelevant to system designers. Such test methods are

necessary, since there is no simple physical model that canaccurately predict the values of dielectric constant anddissipation factor from other physical properties of thedielectric.

In contrast, it has generally been possible to adequatelypredict conductor properties using the materialconductivity and a skin depth model for high frequencyconductivity.

However, modeling of more complex situations, such asthose involving conductor roughness or the use of surfacecoating on conductors (for either adhesion promotion orsolderability enhancement), has required the use of moreempirical approaches (Byers, 2002).

Conductor dissipative losses

Any conductor with a finite conductivity presents resistanceto flow of current. The resistance to flow of direct current ofa material is characterized by its resistivity. The relativebehaviour of conductors towards DC current flow may beinferred from their respective values of resistivity.

Table I shows resistivity values for metals commonlyused in electronic applications (Handbook of Chemistry andPhysics, 2002; Karim, 2001).

At DC, several observations may be made about therelative performance of metals:

. resistivity of silver is approximately 7 percent lowerthan that of copper; and

. resistivity of nickel is approximately 4.5 times largerthan that of copper.

In contrast to the DC model, resistivity is not the onlyparameter responsible for characterizing the resistanceposed by a conductor to the flow of alternating current.

The skin-effect phenomenon introduces a morecomplex relationship involving several material propertiesand the frequency of the alternating current.

Passage of a signal along an interconnection correspondsto the presence of time-varying electric and magnetic fieldsaround the conductor. The degree to which these fieldspenetrate into a conductor is governed by two properties of

the conductor material:1 conductivity electric field, and2 permeability magnetic field.

Solution of Maxwells equations allows calculation of themagnetic and electric fields as a function of depth within theconductor (Ramoet al., 1993).

The form of the solution shows that the magnitude ofboth magnetic field and current density decreaseexponentially with penetration into the conductor. The flowof alternating current is therefore concentrated towards thesurface or skin of the conductor.

A parameter that usefully characterizes the distribution ofcurrent within the conductor is known as the skin depth (d),the depth at which current and magnetic field have decreased

to 1/eof their surface values.

d 1=Pfsm1=2

where fis the frequency (Hz), sis the conductivity (S/m),and m is the permeability H=m m0mr:

If the depth dependent current distribution is integrated,an equivalent surface resistivity to alternating current (Rs)can be calculated.Rs is found to be identical to the DCresistance of a planar conductor of thickness d:

Rs 1=ds Pfm=s1=2

At frequencies where the skin depth is much less than thethickness of the conductor, the relative behaviour ofconductors towards AC current flow may be inferred fromtheir respective values of surface resistivity.

Table II shows the calculated values of surface resistivityfor metals commonly used in electronic applications.

Since surface resistivity is proportional to the square rootof both (conductivity)21 and permeability, the relative

The Emerald Research Register for this journal is available at The current issue and full text archive of this journal is available at

w ww .emeraldinsi ght. com/researchregister w ww .emeraldinsight.com /0305-6120. htm

Keywords

Printed-circuit boards,

Surface conductivity

Abstract

The evolution of digital electronic

systems to ever-faster pulse rise

times has placed increased

demands on printed wiring board

(PWB) materials. Signal loss

associated with dielectric

materials has driven development

and commercialization of cost-

effective low loss laminate

materials. In order to provide a

better understanding of conductor

material and surface finish

choices, efforts have been made

to quantify the impacts of these

factors on loss. An alternative

test approach has been identified

which provides a measure of

conductor performance,

decoupled from both system

geometry and the influence of

laminate material.

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behaviour of metals at AC frequencies differs from thatat DC.

Two observations can be made about these values for

surface resistivity of metals1 Surface resistivities of non-magnetic metals differ by a

smaller proportion than their DC resistivities; forexample, the value for silver is about 4 percent less thanthat of copper.

2 Surface resistivities of magnetic metals differ by agreater proportion than their DC resistivities; forexample, the value for nickel is more than 50 timeslarger than that of copper.

The data shown in Table II would be expected to accuratelyreflect the relative performance for cases where theconductor consists of a single ideal material with a smoothsurface.

A number of cases can be identified where the frequencydependence of the surface resistivity would be expected todiffer from the ideal f1/2 behavior.

In the case of a conductor with a surface finish, consistingof one or more layers of different materials over a substrate,the distribution of the current between the substrate andthe surface finish will depend on the frequency. As thefrequency rises (and current becomes more and moreconcentrated towards the surface of the conductor), theeffective surface resistivity will vary between the values forthe substrate and the surface finish.

A conductor with significant surface roughness wouldalso be expected to show higher values of effective surfaceresistivity at frequencies where the skin depth is comparableto or smaller than the RMS roughness.

Just as the DC resistance of a conductor is inverselyproportional to its cross-sectional area, the AC resistance

of a conductor is inversely proportional to its perimeterlength.

For a typical PWB interconnection:

Perimeter / trace widthtrace thickness

Reduction in conductor loss can therefore be achieved bytwo obvious means, increased conductor width or reducedconductor length.

Use of increased conductor widths comes at the price ofreduced wiring density and increased layer-to-layerseparations (so as to maintain the original transmission lineimpedance).

While industry roadmaps understandably emphasize theevolving needs for finer line capabilities, many highfrequency applications ranging from backplanes to 10 Gbitoptical interface modules use relatively wide traces (Sayre

et al., 1999; Williamset al., 2003).

Conductor dissipative losses surfaceroughness effects

Additional conductor losses at high frequencies, associatedwith surface roughness, have been well known since the1930s. Morgan (1949) published the first quantitativeanalysis of the effect of surface roughness on conductorlosses.

His analysis, based on numerical solution, provides

values for relative power dissipation (versus a smoothsurface) for a number of well-defined surface geometries,including square, rectangular and triangular grooves,oriented both parallel and perpendicular to the direction of

current flow.Morgan characterized the surface roughness effects in

terms of the ratio of root mean square roughness (D) to skindepth (d).

For a surface with regular square grooves, of depth andwidth x and period 2x:

D x=2

Table III shows the values that Morgan derived for relativepower dissipation for the case of square grooves alignedtransversely to the direction of current flow.

In order for a reduction in surface roughness of a

conductor to yield a significant reduction in conductor loss,D/dmust be greater than approximately 0.5 for thefrequencies of interest.

Experimental measurement ofinterconnection loss review of recentpublications

Approaches to measurements of interconnection loss may bedivided into two groups1 measurements of total system loss, and2 measurements of fundamental parameters controlling

loss.

The testing approach adopted will depend on the intendeduse of the data.

A design engineer may wish to carry out measurements tovalidate the results of modeling experiments. As long as themeasured results are acceptably close to the predictedvalues, there is little incentive to develop a more refinedmodel or to isolate the root cause of any deviations.

In contrast, a supplier of PWB materials may wish torecord data to either ensure consistency of supplied productor to develop improved products. In these circumstances, itis desirable to be able to make measurements that enableisolation of the contributions of separate loss mechanisms.

Table I

Resistivity of metals used in electronic applications (at 208C)

Metal Resistivity (mVcm)

Silver 1.59Copper 1.72

Gold 2.44Aluminum 2.82Nickel 7.80

Tin 11.5Tin 99.3 copper 0.7 11.7

Tin 96.5 silver 3.5 12.3Tin 63 lead 37 15

Electroless Ni-P ca. 50-120

Table II

Calculated surface resistivity of metals used in electronic

applications

Metal Surface resistivity (V)

Silver 2.51 1027

( f 1/2

)Copper 2.61 10

27( f

1/2)

Gold 3.10 1027

( f 1/2

)Aluminum 3.33 1027 ( f 1/2)

Tin 6.74 1027

( f 1/2

)Tin 99.3 copper 0.7 6.80 1027 ( f 1/2)

Tin 96.5 silver 3.5 6.97 1027

( f 1/2

)Tin 63 lead 37 7.69 102

7( f

1/2)

Electroless Ni-P 1.422.21026

( f 1/2

)Nickel 1.36 10

25( f

1/2)

Table III

Effect of surface roughness on conductor loss (after Morgan)

RMS roughness/

skin depth (D/d)Relative power

dissipation

0 1.000.25 1.04

0.50 1.171.00 1.57

1.75 1.75

Martins Bayes and Al Horn

Effects of conductor surface

condition on signal integrity

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Brist et al. (2002) and Cullenet al. (2001) have carried outseveral investigations of the effects of dielectric materialsand surface finish on high frequency signal loss.

In their first paper Cullenet al. (2001), the test vehiclesused were microstrip circuits, fabricated from three differentdielectric constant materials (all 500 mm thickness). In orderto maintain a constant 50V characteristic impedance, it wasnecessary to adjust the microstrip widths for each of thelaminate materials.

For each dielectric material, samples were prepared witheach of a number of different surface finishes (OSP, ENIG,immersion silver and HASL) to allow evaluation of theireffects on total system loss.

The differences in loss between the surface finishes wererelatively small (less than 0.04 dB/cm at 10 GHz). Theauthors hypothesized that larger differences would be seenat higher frequencies.

Results of a second phase of measurements (Brist et al.,2002), carried out using a differential pair test structure,were presented at the 2002 SMTA Conference.

Under these test conditions, the effect of surface finishwas found to be more pronounced, due to the greatermagnitude of the electromagnetic field at the conductorsurfaces with the surface finish.

Immersion tin was found to exhibit an increase of

attenuation of approximately 0.2 dB/cm (over OSP andimmersion silver), while a 5.4mm thick electroless nickelwith immersion gold showed an increase of approximately0.6 dB/cm.

Overall levels of loss were found to be higher in thesecond study. The higher losses were attributed at least in

part due to differences in effective conductor cross-sectionand width between single ended microstrip and differentialpair test structures.

Analysis of these data demonstrates the difficulties ofexperimental design and interpretation when many factorsinfluencing both dielectric and conductor loss are varied.

Separation of conductor and dielectricloss

It is desirable to be able to make measurements that clearlyseparate conductor and dielectric effects.

Examples of this type of approach are the standard IPCmethods used to measure dielectric loss tangents.

IPC-TM-650 Methods, 2.5.5.5 and 2.5.5.5.1 (1998a, b)

and Traut (1997a, b) both use an approach in which theconductor losses of a well-characterized electrical structure(a resonant stripline) are subtracted from the total systemlosses to yield values for dielectric loss.

Equations earlier derived (Cohen, 1954, 1955) forconductor loss in a stripline resonator are used to calculate

the conductor correction. The most accurate measurementsof loss tangent are obtained when the use of smooth copperfoils eliminates the need to adjust conductor losses for theinfluence of surface roughness.

A modification of the 2.5.5.5.1 test method has beenevaluated which would allow the measurement of relativesurface resistivity versus a smooth copper surface.

Description of modified IPC-TM-6502.5.5.5.1 test method

The 2.5.5.5.1 method is based on the stripline resonator teststructure shown in Figure 1.

This structure resonates both at a fundamental frequencyand at a number of higher nodes (integral multiples of the

fundamental frequency). At these resonant frequencies, thereflection (or return loss, whichever you want to use) ofmicrowave energy in the structure is diminished.

The width at half power (3 dB points) of each of theseresonances can be used to derive the Q(quality) factor of thestructure, which is related to the total system loss.

The Q factor derived from the measured resonance(Qloaded) requires adjustment to eliminate the effect ofmeasurement process, so as to provide the value of

Qunloaded.The measurement consists of the determination of the

width of the resonances of the different nodes across thedesired range of frequencies (at a specified level of power).The applied power level is maintained within the specifiedrange through the use of a coupling fixture with anadjustable gap dimension.

Signal generation and measurement of the resonantfrequencies and power levels are most easily achieved usinga network analyzer.

The overall losses are related by the following equation:

Loss tangent 1=Qunloaded 2 1=Qconductor

1/Qconductoris the conductor loss factor.The (complex) equations for the conductor loss are as

follows (IPC-TM-650, Methods 2.5.5.5 and 2.5.5.5.1, and1998a, b):

1=Qconductor acc=pfr1r0:5

ac 4Rs1rZ0Y=3772B

Rs 0:00825fr0:5 surface resistivity

Z0 377=41r0:5Cf W=B 2 T

377 120p Free space impedance V

Cf 2XlogeX 1 2 X2 1 logeX221=p

Y X 2WX2=B X21 T=B

logeX 1=X21=p

X 1=1 2 T=B

where 1r is the relative permittivity of dielectric;B is theground plane spacing (mm); c is the 299.976 speed of light(mm/ns); Wis the resonator width (mm); Tis the resonatorconductor thickness (mm); and fris the node resonantfrequency (GHz).

The only parameters in these equations that are notsimple functions of the physical dimensions of the teststructure are the relative permittivity of the dielectric, thesurface resistivity of the conductor and the frequency ofthe resonance.

Since the relative permittivity can be calculated fromthe frequencies at which a resonance occurs, the onlyremaining variables are the surface resistivity and the

frequency.The equation for 1/Qconductorcan therefore be expressed

in a simplified form as follows:

1=Qconductor KRs=K0fr

An estimate of the surface resistivity of a conductor versusthat of a reference sample may be made as follows.. An initial series of measurements of (1/Qunloaded) are

made with smooth copper conductors and two sheets ofdielectric material.

. Values of (1/Qreference conductor) for the smooth copperconductors are calculated from the conductor lossequations, as a function of frequency.

. Values of loss tangent for the dielectric material arethen calculated from the values of (1/Qunloaded)and(1/Qconductor).

. The smooth copper conductors are removed andreplaced with the conductors to be tested. These testsare performed with the same sheets of dielectricmaterial.

. Values for Qunloadedare obtained as a function offrequency.




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. The values ofQunloadedand the previously measuredvalues for the loss tangent of the dielectric material areused to calculate values for (1/Qtest conductor).

. The ratio of the measured conductor loss to thereference conductor loss provides a measure of theirrelative surface resistivities:

1=Qtest conductor1=Qreference conductor

KRs test=K

0fr

KRs reference=K0fr

Rs test

Rs reference

Use of a dielectric material with as low as possible a losstangent, will increase the proportion of the loss associatedwith the conductor, and thus improve the measurementaccuracy of the conductor loss.

Experimental

Reference data were obtained from a 15.2 cm long resonatorstructure, prepared from smooth, mechanical wroughtcopper foil (52mm thick). The widths of the two ground

planes and the center conductor were 2.54 and 0.64 cm,respectively.

Test samples were either prepared by further processingsamples of the wrought foils (surface finishes/inner-layeradhesion promotion) or cutting samples of commerciallyavailable PWB foil materials to the required dimensions.

Two 1.55 mm thick sheets of RT/Duroidw 5880 PTFEcomposite laminate (Rogers Corporation, Rogers, CT) wereused as the low loss dielectric material.

Measurements were made using a Hewlett Packard8510C Network Analyzer over a range of frequencies from

600MHz to 14 GHz.The overall test matrix was as follows.Sample type/descriptionElectroless nickel 8 percent P: 1.25mmElectroless nickel 8 percent P: 2.5 mm

Electroless nickel 8 percent P: 3.75mmElectroless nickel 8 percent P: 5 mmElectroless nickel 8 percent P: 6.25mm

Immersion tin: 0.5mmImmersion tin: 1.5mm

Black oxideConverted black oxideOxide replacement: 1mm etchOxide replacement: 3mm etchElectrodeposited (ED) Foil A: vendor treat sideED Foil B: vendor treat sideED Foil B: reverse treated foil (RTF) side

Results and discussion electroless

nickel

The results of tests of electroless Ni-P (8 wt percent P)

surface finishes, varying from 1.25 to 6.25 mm, shown in

Figure 2, clearly illustrate the effects of both coatingthickness and frequency on relative surface resistivity(relative Rs).

At low frequencies, all values of relative Rstend towards1, as the behavior of the conductor approximates to that ofthe underlying copper.

As frequency rises, the relativeRs of all the electrolessnickel samples rises. The degree of increase is stronglyaffected by the deposit thickness.

At an electroless nickel thickness of 5 mm or less, valuesofRs increase progressively over the range of frequenciesevaluated. The degree of increase of the thinner coatings issmaller (1.5 for the 1.25 mm thickness and 3.3 for the2.5mm thickness).

The relativeRs for the sample with a thickness of6.25mm, reaches a stable value (approximately 8) above6 GHz, presumably as the current flow above that frequencybecomes restricted to the coating alone.

The value of the relative Rs at frequencies above 6 GHzcorresponds to a resistivity 82, i.e. 64 times larger than

copper, since the surface resistivity is proportional to thesquare root of the resistivity.

Based on a value of copper resistivity of 1.7 mm cm, thevalue of resistivity for electroless nickel (8 wt percent P) iscalculated to be approximately 110 mV cm, in goodagreement with previously published values.

Results and discussion immersion tin

The results of tests of two different thickness (0.5 and

1.5mm nominal) samples of immersion tin are shown inFigure 3. The samples show behaviour that is analogous inform to that of the electroless nickel samples, with lowfrequency relative Rs tending towards 1.

The relativeRs of the 1.5mm thick sample reaches a

limiting value (of approximately 4.4) at about 4 GHz.The observed limiting value corresponds to a relativeresistivity versus copper of (4.4)2 i.e. 19.4. Based on thisvalue, the value of resistivity for an immersion tin coating iscalculated to be approximately 33mV cm.

While this value is substantially larger than the value forpure tin (11mV cm), it is certainly not unreasonable,considering the formation mechanism and structure of theimmersion tin deposits.

The lower thickness required to reach the limitingrelativeRs(versus electroless nickel) may be a consequenceof the lower resistivity of the immersion tin deposit, sincelower resistivity yields smaller values of skin depth.

The 0.5mm thick sample shows much lower values ofrelative Rs, suggesting that the copper substrate dominates

performance over the range of frequencies examined.

Results and discussion PWB foiladhesion treatments

Results of tests carried out on PWB foil surfaces are shownin Figure 4.

Results obtained for the vendor side treatment of Foil Ashowed an increase in relative Rs from 1 at 600 MHz toapproximately 1.7 at 14 GHz.

The relativeRsbehaviour of the vendor and RTF sides ofFoil B was relatively similar (and considerably superior toFoil A). The values of relative Rs at low frequencies wereless than 1, possibly reflecting the effects of measurementerrors. At higher frequencies, the relativeRs increase withthe values reaching 1.1 at 14 GHz.

Measurements of the surface roughness of the foil

samples were taken using a Dektak (Model 200v) stylusprofilometer. Results were derived from 256 parallel scans,covering a total sampled area of 500 500mm:

The measured values of RMS deviation (variance) andRa (mean deviation) for each of the three foil samples areshown in Table IV.

Figure 1

Stripline resonator structure




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These measurements confirm that the sample exhibiting thehighest relative Rs (Foil A) was the roughest of the threesamples.

However, while the surface roughness measurementsshow a significant difference between the two sides of FoilB, the relative Rs behavior of the two sides is very similar.This may indicate that factors other than surface roughnesscontribute to the measured loss behaviour.

Results and discussion inner-layer

adhesion promotion

The behaviour of the four adhesion promotion processesevaluated was found to be indistinguishable from wrought

copper foil (within the experimental error) between600MHz and 14 GHz.

Based on these results, the surface composition andmorphology changes produced by these inner-layeradhesion promotion processes are too small to have ameasurable impact on loss.

Repetition of these experiments on ED foil substrates isplanned, to determine if differences in copper grain structuremight have an influence on performance.

Conclusions

. A simple modification of an existing dielectric testmethod IPC-TM-650 2.5.5.5.1, allows measurement ofconductor surface resistivity relative to a referenceconductor (a smooth copper foil) at frequencies up to14 GHz.

. Preliminary results using this technique demonstrate theability to quantitatively measure the effects of both

nature and thickness of surface finishes and surfaceroughness of PWB foil surfaces on conductor surfaceresistivity.

. Values of electroless nickel resistivity derived fromthese measurements are consistent with valuespreviously reported.

Figure 2

Relative surface resistivity of electroless nickel over a copper substrate

Figure 3

Relative surface resistivity of immersion tin over a copper substrate




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. Values of immersion tin resistivity derived from thesemeasurements are about three times higher thanliterature values for pure tin.

. Inner-layer adhesion promotion processes, applied tothe wrought copper foil surface, were found to havealmost no influence on relative surface resistivity.

Notes

1 Ansoft HFSSe Available at: www.ansoft.com

2 Mayo Clinic Special Purpose Processor Development Group,Available at: http//:www.mayo.edu/sppdg

ReferencesBayes, M. (2003), Printed wiring board material sets implications

of high frequency applications, IPC EXPO, March 2003.

Brist, G.et al. (2002), Reduction of high-frequency signal loss

through the control of conductor geometry and surface

metallization,SMTA Conference 2002.

Byers, A. (2002), Accounting for high frequency transmission line

effects in HFSS, Presentation at Ansoft Design Conference.

Available at Web site: http://www.ansoft.com/hfworkshop03/

andy_Byers_Tektonix.pdf, Los Angeles, CA

Cohen, S.B. (1954), Characteristic impedance of the shielded-strip

transmission line, IRE Trans. MTT, No. 2, pp. 52-7.

Cohen, S.B. (1955), Problems in strip transmission lines,IRE

Trans. MTT, No. 3, pp. 119-26.

Cullen, D.et al.(2001), Effects of surface finish on high frequency

signal loss using various substrate materials, IPC Expo, April

2001.

(1998a), Stripline test for permittivity and loss tangent at X-band,

Revision C.

(1998b), Stripline test for complex relative permittivity of circuit

board materials to 14 GHz.

Karim, Z. (2001), Lead-free solder bump technologies for flip chip

packaging applications,Proceedings of Advanced Packaging

Technology Conference, SMTA, 2001.

Morgan, S. (1949),J. Appl. Phys., Vol. 20 No. 4, pp. 352-62.

Ramo, S., Whinnery, J. and Van Duzer, T. (1993),Fields and Waves

in Communication Electronics, Wiley, New York, NY.

Sayre, E.et al. (1999), OC-48/2.5 Gbps interconnect engineering

design rules,Digital Communication System Design

Conference.

Traut, G.R. (1997a), Complex permittivity of RF circuit boardmaterials by resonances of a stripline section in the 0.2-15 GHz

range,Measurement Science Conference, January 1997.

Traut, G.R. (1997b), Complex permittivity over a wide frequency

range by adjustable air gap probing a stripline resonator, IPC

EXPO, March 1997.

Weast, R.C. (Ed.) (2002),Handbook of Chemistry and Physics,

CRC Press, Florida, FL.

Williams, L.et al.(2003), Simulating the XFP electrical interface,

CommsDesign, Available at Web site: www.commsdesign.

com/design_corner/OEG20030129S0013

Figure 4

Relative surface resistivity of PWB foil treatments

The authors would like tothank Pat Larrow of Rogers

Corporation for her

invaluable assistance with

the electricalmeasurements.

Table IV

Surface roughness measurements of PWB foil adhesion

treatments (mm)

Sample Treatment RMS Ra

Foil A Vendor side 3.05 2.42

Foil B RTF side 1.42 1.13Foil B Vendor side 1.00 0.72




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