High-Frequency Transformer Isolated Bidirectional DC DC

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 12, DECEMBER 2011 3599

High-Frequency Transformer Isolated BidirectionalDC–DC Converter Modules With High Efficiency

Over Wide Load Range for 20 kVASolid-State Transformer

Haifeng Fan, Student Member, IEEE, and Hui Li, Senior Member, IEEE

Abstract—This paper presents the design of new high-frequencytransformer isolated bidirectional dc–dc converter modules con-nected in input-series-output-parallel (ISOP) for 20-kVA-solid-state transformer. The ISOP modular structure enables the useof low-voltage MOSFETs, featuring low on-state resistance andresulted conduction losses, to address medium-voltage input.A phase-shift dual-half-bridge (DHB) converter is employed toachieve high-frequency galvanic isolation, bidirectional power flow,and zero voltage switching (ZVS) of all switching devices, whichleads to low switching losses even with high-frequency operation.Furthermore, an adaptive inductor is proposed as the main energytransfer element of a phase-shift DHB converter so that the circu-lating energy can be optimized to maintain ZVS at light load andminimize the conduction losses at heavy load as well. As a result,high efficiency over wide load range and high power density canbe achieved. In addition, current stress of switching devices canbe reduced. A planar transformer adopting printed-circuit-boardwindings arranged in an interleaved structure is designed to obtainlow core and winding loss, solid isolation, and identical parametersin multiple modules. Moreover, the modular structure along with adistributed control provides plug-and-play capability and possiblehigh-level fault tolerance. The experimental results on 1 kW DHBconverter modules switching at 50 kHz are presented to validatethe theoretical analysis.

Index Terms—Circulating energy, high efficiency, high-frequency transformer, input-series-output-parallel (ISOP), iso-lated bidirectional dc–dc converter, solid-state transformer (SST),zero voltage switching (ZVS).

I. INTRODUCTION

THE future intelligent electric energy distribution and man-agement systems are expected to integrate highly dis-

tributed and scalable renewable generating sources, whereinsolid-state transformer (SST) is one of the key enabling

Manuscript received January 10, 2011; revised April 26, 2011; accepted June6, 2011. Date of current version December 6, 2011. This work was supportedby the National Science Foundation under Award Number EEC-0812121. Rec-ommended for publication by Associate Editor J. Biela.

H. Fan is with Texas Instruments Inc., Tempe, AZ 85284 USA (e-mail:[email protected]).

H. Li is with the Department of Electrical and Computer Engineering,Florida A&M University—Florida State University College of Engineering,Tallahassee, FL 32310 USA (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2011.2160652

elements and intended to replace the conventional line-frequency (50/60 Hz) transformer based on iron/steel cores andcopper/aluminum coil [1]–[3]. Besides the ability of the conver-sion and control of electric energy, SST, a new power electronicssystem, can provide many additional advantages such as intelli-gent energy management, good power quality, low weight, lowvolume, and high power density. Various configurations for SSTwere reported in [1]–[7], of which the ac–dc–dc–ac configura-tion is the most popular one due to its capability to providepower factor correction, reactive power compensation, and anadditional dc bus [2], [4]–[7]. Fig. 1 shows the circuit diagram ofa 20-kVA-single-phase SST consisting of an ac–dc rectifier, anisolated bidirectional dc–dc converter, and a dc–ac rectifier. Thehigh-frequency transformer isolated bidirectional dc–dc con-verter not only provides the galvanic isolation function and a dcbus, but also becomes critical to determine the overall efficiencyand power density of the system, and, therefore, is one of themain challenges for the SST design.

There are two possible ways to address the demand of mediumvoltage (MV) and high power capability of the dc–dc con-verter stage of SST. One is to develop a semiconductor tech-nology and /or directly connecting devices in series to reachhigher nominal voltages while maintaining conventional con-verter topologies; the other is to develop new converter topolo-gies with traditional semiconductor technology, known as mul-tilevel converters [4], [5], [8] or modular converters [9]–[11].The former inherits the benefit of well-known circuit structuresand control methods. However, the new power semiconductordevices are more expensive. In addition, the power semicon-ductor devices with higher voltage rating usually have very highswitching losses when operated with the switching frequencyof more than 20 kHz, and, therefore, fail to achieve high ef-ficiency and high power density. Multilevel converters enablelow-voltage rating switches to be used under MV input appli-cations. However, for those over than three-level converters, thesystem reliability cannot be guaranteed for a large quantity ofdiodes or flying capacitors [12], [13]. By adopting new con-verter topologies, the modular approach is able to use the wellknown, mature, and more economic power semiconductor de-vices to handle MV power conversion. The main advantages ofthe modular approach include: 1) significant improvement inreliability by introducing desired level of redundancy; 2) stan-dardization of components leading to reduction in manufactur-ing cost and time; 3) power systems can be easily reconfigured

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3600 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 12, DECEMBER 2011

Fig. 1. Circuit diagram of 20-kVA-single-phase SST.

to support varying input-output specifications; and 4) possi-bly higher efficiency and power density of the overall system.Input-series-output-parallel (ISOP) modular configuration is themost promising candidate to realize MV to low voltage conver-sion for SST application.

Several ISOP modular dc–dc converters have been reportedin [14]–[18], and much of previous work has focused on devel-oping control techniques to achieve input voltage and output cur-rent sharing, stable operation, and better dynamic performancein terms of ISOP configuration. However, all the previous workhas little attention paid to the topology of the constituent mod-ules of ISOP modular dc–dc converter, and none of them has setMV power system as the target application. Although varioustopologies have been employed for the constituent modules ofISOP modular dc–dc converter in previous research, unfortu-nately, none of them can meet the requirements of SST. Phase-shift full-bridge converter presented in [16] and [17] is attrac-tive for high power application, and can achieve zero-voltage-switching (ZVS) operation for primary switches, but fails to pro-vide bidirectional power flow path due to the rectifying diodes onsecondary side. The forward converter in [14] and [15] requiresadditional circuitry for transformer reset and lacks bidirectionalpower flow capability. The push–pull converter proposed as theconstituent module of ISOP modular converter for informationtechnology equipment application in [18] succeeds to providebidirectional power flow, and achieve high efficiency with high-frequency operation with each module operated with less 10-Vinput. However, the voltage stress of primary side switches inthe push–pull converter twice the input voltage; hence, it is notsuitable for the MV application.

None of previous work, however, has investigated the phase-shift dual-bridge bidirectional dc–dc converter as the constituentmodule of the ISOP converter, although it appears to be themost promising candidate for MV power conversion systemdue to its capability to achieve high-frequency transformer iso-lation, bidirectional power flow, and ZVS operation for allswitching devices without auxiliary switch devices. The phase-shift dual-bridge dc–dc converters such as dual-active-bridge(DAB) and dual-half-bridge (DHB) converters have been re-ported in [19]–[22]. However, these converters can operate inthe ZVS mode only within a limited region restricted by the con-verter voltage ratio of input to output and the load condition, andsuffer additional conduction losses due to the circulating energyat heavy load. Consequently, high efficiency can be achievedonly within a limited load range. A few new control methodswere proposed in [23] and [24] to handle the loss of ZVS due

to input voltage variations. However, none of these previousattempts addressed both the loss of ZVS at light load and theadditional conduction losses due to circulating energy at heavyload at the same time.

This paper proposes an ISOP modular DHB converter as thehigh-frequency transformer isolated bidirectional dc–dc con-verter stage of SST. The proposed converter employs phase-shift DHB topology for the individual modules to achieve high-frequency galvanic isolation, bidirectional power flow, and ZVSoperation of all the switch devices. The ZVS technique com-bined with low-voltage MOSFETs leads to low switching andconduction losses even under high-frequency operation. Fur-thermore, an adaptive inductor is proposed as the main energytransfer element of phase-shift DHB converter so that the cir-culating energy can be optimized to maintain ZVS operation atlight load and minimize the conduction losses at heavy load aswell. As a result, the efficiency at both light and heavy load canbe significantly improved compared with the conventional DHBconverter with fixed commutation inductor, and, therefore, highefficiency over wide load range and high power density can beachieved. In addition, current stress of switching devices canbe reduced. Moreover, a planar transformer adopting printed-circuit-board (PCB) windings arranged in an interleaved struc-ture is designed to obtain low core loss, low winding loss, solidisolation, and identical parameters in multiple modules. Besides,the modular structure along with a distributed control schemeprovides plug-and-play capability and possible high-level faulttolerance. Finally, the experimental results on a 1-kW-DHB con-verter module switching at 50 kHz are presented to validate thetheoretical analysis.

II. HIGH-FREQUENCY TRANSFORMER ISOLATED

BIDIRECTIONAL DC–DC CONVERTER

A. System Description of the Proposed ISOP Modular DHBConverter

As shown in Fig. 2, the SST in the future renewable electricenergy delivery and management (FREEDM) system not onlydelivers energy to loads but also integrates distributed renewableenergy sources and energy storage devices. Similar to the blockdiagram shown in Fig. 1, the SST consists of an ac–dc rectifier,an isolated dc–dc converter, and a dc–ac inverter. The ac–dc rec-tifier interfacing with the 7.2-kV-electric utility grid is used toprovide power factor correction function while converting 7.2-kV ac to 12-kV dc. The dc–dc converter, the key stage of SST,provides high-frequency galvanic isolation and converts 12-kV

FAN AND LI: HIGH-FREQUENCY TRANSFORMER ISOLATED BIDIRECTIONAL DC–DC CONVERTER MODULES 3601

Fig. 2. Single-phase SST in the FREEDM system at one residential home.

Fig. 3. ISOP modular DHB converter.

dc to 400-V dc as well. The 400-V dc is used to interface dcloads, distributed renewable energy sources, and distributed en-ergy storage devices, and meanwhile it can also be converted to120/240-V ac for end-use application through a dc–ac inverter.

At FREEDM systems center, two different approaches arebeing investigated for the line side ac–dc rectifier. One is theflying-capacitor multilevel converter reported in [25], and theother is the multilevel-cascaded H-bridge converter presentedin [26]. In order to interface with 12-kV dc bus generated by anac–dc rectifier from 7.2-kV-electric utility grid, the proposedhigh-frequency transformer isolated dc–dc converter for thedc–dc conversion stage of SST, as shown in Fig. 3, is dividedinto multiple low-voltage dc–dc converter modules connectedin ISOP configuration. The input and output voltages of eachmodule are chosen as 500 and 400 V, respectively. Thus, thelow-voltage commercial silicon MOSFETs with low conduc-tion losses and high switching speed can be selected as theswitching device. The complete dc–dc converter, of which the3-D design is shown in Fig. 4, has an eight-layer structure with

Fig. 4. 3-D design of the proposed ISOP modular DHB converter.

three modules on each layer. Each module is a bidirectionaldc–dc converter, which adopts phase-shift technique to real-ize ZVS operation for all switching devices without auxiliaryswitching devices in either direction of power flow [7]–[9], and,therefore, enables the high-frequency operation while keepinglow switching losses. Although a total of 24 modules will beused to interface MV, the utilization of low-voltage device alongwith ZVS operation results in high efficiency, high frequency,good thermal performance, and eventually high power densityof the dc–dc conversion stage. As a result, the SST can achievemuch smaller size than conventional line-frequency (50/60 Hz)transformer by adopting the proposed high-frequency high-efficiency modules design.

B. Topology Selection of the Constituent Modules

DAB and DHB are two popular topologies among phase-shiftdual-bridge bidirectional dc–dc converters. Fig. 5 and Table Icompare the operational conditions of DAB and DHB convert-ers. Transformer flux swing of DHB is only half of DAB’s when


Fig. 5. Comparison between DAB and DHB.

TABLE IOPERATIONAL CONDITIONS COMPARISON

same switching frequency and effective cross-sectional area ofthe transformer core are employed. It indicates that the DHBconverter achieves smaller transformer core loss, which will beelaborated in Section IV. Moreover, a DHB’s use of half thenumber of switching devices and corresponding drivers as DABresults in a more economical implementation especially in thismultiple-module structure. The phase-shift DHB is, therefore,selected for the dc–dc converter module in this paper.

The output power of phase-shift DHB can be expressed as

Pout =n (Vin/2) (Vout/2) φ (π − |φ|)

2π2fL(1)

where L is the sum of the leakage inductance of the transformerand the external auxiliary inductance and ϕ is the phase-shiftangle. Then, the output current can be given by

Iout =Pout

Vout=

nVinφ (π − |φ|)8π2fL

. (2)

The output current expression suggests that the output currentIout is independent from the output voltage Vout . This uniquecurrent source like characteristics distinguishes DHB from thebuck-derived dc–dc converters, and enables inherent stable op-eration when they are connected in input-parallel-out-parallel(IPOP). As a result, multiple ISOP modular DHB converterscan be directly connected in IPOP to extend the power ratingwhile no additional control is needed. This is an important ad-vantage of the proposed ISOP modular DHB converter for theMV high-power application.

Fig. 6. Conventional DHB converter.

Fig. 7. Proposed DHB converter with an adaptive inductor.

Fig. 8. Key ideal waveforms of the phase-shift DHB converter.

III. NEW PHASE-SHIFT DHB CONVERTER WITH HIGH

EFFICIENCY OVER WIDE LOAD RANGE

A. Operation Principle Description

Figs. 6 and 7 show the circuit diagram of the conventionalDHB converter and that of the proposed DHB converter withan adaptive inductor, respectively. The key ideal steady-statewaveforms of the two converters are similar and are shown inFig. 8. The operation of the conventional DHB converter is fullydescribed in [19]–[22]. In the proposed new DHB converter, anauxiliary adaptive inductor L is used as the commutation in-ductor and can be controlled to adapt to the output power by


utilizing the output current Iout as the bias current IBIAS , whilethe commutation inductance of the conventional DHB converteris fixed regardless of the output power. This feature enables thenew DHB converter to be operated with optimized L, ϕ, and cor-responding circulating energy over the entire load range. Theappropriate circulating energy will help the converter reduceswitching losses by maintaining ZVS operation at light loadand minimize additional conduction losses caused by circulat-ing energy at heavy load as well. As a result, the efficiencyunder both light and heavy load conditions, compared with theconventional DHB converter, can be significantly improved.

To simplify the analysis, the converter voltage ratio of theinput voltage Vin to output voltage Vout is assumed to be 1(i.e.,Vout = NsVin /Np , where Np and Ns are the number of pri-mary and secondary turns in the transformer, respectively.). Theoutput power can be given by

Pout =1

t7 − t0

∫ t7

t0

vaeiL = V 2inϕ(π − |ϕ|)

8π2fL(3)

where iL is the instantaneous current of commutation inductorL and ϕ is the phase shift usually ranging from −π/2 to π/2.The output power of the proposed converter can be controlledby ϕ and L as well, while the output power of the conventionalDHB converter can only be controlled by ϕ.

The instantaneous current of commutation inductor at t2 canbe given by

iL(t2) = − Vinϕ

4πfL. (4)

By combing (3) and (4), the current stress of S2 can be givenby

∣∣iL(t2)∣∣ =

2πPout

Vin (π − |ϕ|) . (5)

The delivered power during (t0–t1) and (t2–t6) can be obtainedby

{Pd01 = 1

t1 −t0

∫ t1

t0vaeiL

Pd26 = 1t6 −t2

∫ t6

t2vaeiL .

(6)

Combining (6) and the boundary conditions iL (t1) = −iL (t0)and iL (t2) = −iL (t6) yields

Pd01 = Pd26 = 0 (7)

which means that the average power during both (t0–t1) and (t2–t6) are equal to zero, but the energy stored in the commutationinductor will circulate in the circuit and generate additional con-duction losses; the total circulating energy during one switchingperiod (t0–t7) can be given by

EC = 4∫ t5

t2

vaeiL = V 2inϕ2

8πfL=

ϕPout

2f(π − |ϕ|) . (8)

EC is the sum of the energy stored in the commutation in-ductor during (t0–t1) and (t2–t6), and the energy stored in thecommutation inductor during each transition period is half of

Fig. 9. Circulating energy of the conventional DHB converter with an opti-mized fixed inductor.

EC , and can be expressed as

EL =EC

2=

ϕPout

4f (π − |ϕ|) . (9)

ZVS operation of the switching devices is achieved by uti-lizing the circulating energy EC to discharge the output capac-itance Co of the switching devices, and the circulating energyEC should be no less than the total capacitor energy to maintainZVS operation, i.e.,

EC ≥ Eth (10)

where Eth is the energy of ZVS threshold and insufficient energyEC will result in the loss of ZVS of the switching devices. On theother hand, large circulating energy will cause large additionalconduction losses. In order to achieve high efficiency over awide load range, EC is expected to be large enough to maintainZVS to reduce switching losses at light load, but to be minimizedto reduce additional conduction losses at heavy load.

In the conventional phase-shift DHB converter, commutationinductance L is fixed and the output power is controlled only byϕ. According to power equation (3), phase shift ϕ, if rangingfrom −π/2 to π/2, is proportional to the output power Pout .Combined with (8), it can be seen that the circulating energyEC is proportional to Pout , and therefore, EC at heavy loadis always much larger than that at light load due to larger ϕand Pout . Consequently, L can only be optimized for a limitedload range, and so does the high efficiency. Fig. 9 shows thecirculating energy with respect to the output power of the con-ventional DHB converter with an optimized fixed inductance.It can be seen that the circulating energy is too small to main-tain ZVS under light load conditions while it is too large andwill cause large additional conduction losses under heavy loadconditions.

The proposed phase-shift DHB converter with an adaptiveinductor in this paper can overcome the aforementioned draw-backs by introducing another control variable, commutation in-ductance L. At light load, L is controlled to be relatively largeto obtain large ϕ and, therefore, sufficient EC to maintain ZVSof switching devices. On the other hand, L at heavy load is con-trolled to be much smaller than that at light load to reduce ϕ


Fig. 10. Adaptive inductor. (a) Schematic. (b) Photo.

Fig. 11. Inductance as a function of dc bias current.

and, therefore, reduce EC . As a result, the circulating energy andthe resulted additional conduction losses at heavy load can beminimized without losing ZVS at light load. Compared with theconventional phase-shift DHB converter, the efficiency of theproposed dc–dc converter can be significantly improved underboth light and heavy load conditions, and high efficiency can beachieved over wide load range.

B. Adaptive Inductor Design

The adaptive inductor can be implemented using a dou-ble rectangular modulus (RM) core, as explained in [27]. Theschematic and the photo are shown in Fig. 10. The main induc-tance L is wound around the center leg whereas the symmetricalbias winding is wound around the two side arms. The side armswindings are serially connected in opposite polarity to cancelout the ac voltages induced by the center leg. The inductance Lcan be controlled by the bias current IBIAS . The output currentIout is used as the bias current in this paper so that L can beoptimized according to the output power automatically.

C. Improvement of the Phase-Shift DHB ConverterWith an Adaptive Inductor

The numerical analysis is presented on one dc-dc convertermodule in this section. The specifications are: f = 50 kHz, ratedpower Pout rated = 1 kW, Vin = 500 V, and Vout = 400 V. Forthe comparison purpose, an optimized fixed 90-μH inductor Lis chosen for the conventional phase-shift DHB converter, whilethe adaptive inductance as shown in Fig. 11 is adopted for theproposed DHB converter.

By substituting L into (3), ϕ as a function of output powercan be obtained as shown in Fig. 12. Then according to (5)and (8), the current stress of S2 and the circulating energyEC can be obtained as shown in Figs. 13 and 14, respectively.

Fig. 12. Phase shift with respect to output power.

Fig. 13. Current stress of the DHB converters.

Fig. 14. Circulating energy of the DHB converters.

Compared with the conventional DHB converter, the proposedDHB converter with an adaptive inductor can not only achievemuch higher circulating energy at light load to maintain ZVS toreduce switching losses, but also realize much lower circulatingenergy at heavy load to reduce the additional conduction losses.Therefore, high efficiency can be achieved over wide load range.

IV. HIGH-FREQUENCY TRANSFORMER DESIGN

Planar transformer with coils encapsulated within multi-layer PCB can achieve lower profile and higher power den-sity than conventional wire-wound transformer especially forthe multiple-module system. In addition, the windings of trans-former are etched within the PCB and, thus, are completely re-peatable; this can make the windings of the transformer identicalin multiple modules and contribute to the balance among thesemodules. Furthermore, the planar transformer utilizes solid


Fig. 15. Planar transformer.

Fig. 16. Transformer core loss with respect to frequency and primary voltage.

insulation excluding air from the construction to minimizecorona and partial discharge and, therefore, enhance reliabil-ity of SST. However, it is difficult to find a planar core suitablefor this high-voltage application requiring large cross-sectionalarea. In this paper, a pair of PC40 PQ107/87/70 ferrite cores ismodified to much lower profile while keeping the desired cross-sectional area. After modification, the total window height ofthe transformer is reduced from 56 to 4.55 mm. The final trans-former prototype is shown in Fig. 15; the primary to secondaryturn ratio is 15:12 and the core loss can be calculated by thefollowing empirical formula:

Pcl = VeCm fxByac (11)

where Ve is the effective core volume of transformer; Cm , x,and y are the coefficients related to the core material; Bac is themaximum flux density and can be expressed as

Bac =VT D

2NpAef(12)

where Np is the primary number of turns and VT is the appliedvoltage on the primary side of transformer.

Fig. 17. Cross section of transformer winding and corresponding plot of Jand H distributions. (a) Noninterleaved winding arrangement. (b) Interleavedwinding arrangement.

As shown in Fig. 5, VT equals Vin for DAB and Vin /2 forDHB. Fig. 16 shows the calculated transformer core loss at 80◦C with respect to VT and f. The higher the frequency f andthe lower VT , the lower the core loss is. For 50 kHz operation,core loss of DAB with VT = 500 V is 10.56 W, while the coreloss of DHB with VT = 250 is only 1.778 W, which verifies theanalysis that DHB has much lower core loss than DAB whensame switching frequency and effective cross-sectional area ofthe transformer core are employed.

Both skin effect and proximity effect will increase high-frequency copper losses in transformer winding [28], and there-fore, these effects must be taken into account when designingthe transformer winding. PCB winding offers the flexibility toachieve the winding structure as desired. In this paper, ten-layerPCB with 2 oz copper is adopted for the transformer wind-ing. Fig. 17 shows two different winding arrangements and


TABLE IIKEY SPECIFICATIONS AND CIRCUIT PARAMETERS

Fig. 18. ac resistance of the transformer winding with respect to frequency.

Fig. 19. Photos of prototype. (a) Experimental setup in the lab. (b) One DHBconverter module with an adaptive inductor.

Fig. 20. Key waveforms of two DHB converters at 300 W. (a) New DHBconverter with an adaptive inductor. (b) Conventional DHB converter with afixed inductor.

their corresponding simulation results of current and magneticfield strength distribution in each layer. Other than the non-interleaved winding shown in Fig. 17(a), a triple interleavedwinding arrangement is utilized to optimize the magnetic fieldstrength and, therefore, reduce ac resistance in this paper. Themaximum magnetic field strength of the interleaved windingis only 1000 A/m, while that of noninterleaved winding isup to 3200 A/m. Fig. 18 shows the total ac resistance of the


Fig. 21. Key waveforms of two DHB converters at 1 kW. (a) New DHBconverter with an adaptive inductor. (b) Conventional DHB converter with afixed inductor.

transformer winding with respect to switching frequency. Com-pared with noninterleaved winding, the triple interleaved wind-ing presented in this paper can achieve much lower high-frequency ac resistance and, therefore, lower winding loss. Thewinding loss of the interleaved transformer winding is 7.48 Wout of 30.43 W overall loss of the dc–dc converter at rated outputpower.

V. EXPERIMENTAL RESULTS

The stable and accurate input-voltage and output-current shar-ing of the proposed ISOP modular DHB converter have beenverified by both simulation and experiments in [29]. In this pa-per, a 50-kHz, 1-kW new DHB converter module, as shownin Fig. 19, was built in the lab and tested to verify the high-

Fig. 22. Measured efficiency of two DHB converters.

frequency- and high-efficiency operation. The specifications andcircuit parameters of the individual converter module are shownin Table II.

The conventional DHB converter with fixed inductor was alsotested and compared with the proposed new DHB converter withan adaptive inductor to validate the extended high-efficiencyrange of the latter. The adaptive inductance shown in Fig. 11 isadopted for the new DHB converter while an optimized fixed90 μH is chosen for the conventional DHB converter. Fig. 20shows the key switching waveforms of two DHB converters un-der light load condition. ZVS of S2 of the DHB converter withan adaptive inductor can be maintained in Fig. 20(a), while S2 ofthe conventional DHB converter loses ZVS in Fig. 20(b). Fig. 21shows the switching waveforms of the two DHB converters un-der heavy load condition. The DHB converter with an adaptiveinductor can achieve lower current stress and smaller phase shiftwhich means smaller circulating energy and resulted additionalconduction losses. Fig. 22 shows comparison of the measuredefficiency of two DHB converters. Both DHB converters canrealize high efficiency during mid-range load conditions. How-ever, the DHB converter with an adaptive inductor can achievehigher efficiency under both light and heavy load conditions.

VI. CONCLUSION

High-frequency transformer isolated bidirectional dc–dc con-verter modules with high efficiency over wide load range havebeen proposed and designed for 20 kVA SST in this paper. Theproposed converter modules are connected in ISOP modularstructure to enable the use of low-voltage MOSFETs, featuringlow on-state resistance and resulted low conduction losses, toaddress MV power conversion. A phase-shift DHB converterhas been chosen for the individual modules to achieve high-frequency galvanic isolation, bidirectional power flow, and ZVSoperation of all the switching devices, which leads to low switch-ing losses even with high-frequency operation. Furthermore, anadaptive inductor has been first introduced in this paper as themain energy transfer element of a phase-shift DHB converter sothat the circulating energy can be optimized to maintain ZVSoperation at light load and minimize the conduction losses atheavy load as well. As a result, high efficiency over wide loadrange and high power density can be achieved. Besides, thecurrent stress of switch devices in the proposed DHB converterwith an adaptive inductor can be reduced when compared with


the conventional DHB converter. The adaptive inductor can beapplied to other phase-shift dual-bridge converters, and sameimprovements can be obtained. In addition, a planar transformerwith interleaved windings is designed and implemented to ob-tain low core loss, optimized high-frequency copper loss, lowprofile, and solid insulation which leads to enhanced reliability.The theoretical analysis is validated by the experimental results,which indicates that high efficiency up to 97.2% over a wideload range with 50 kHz operation can be achieved. Moreover,the proposed modular structure along with a distributed con-trol scheme can provide plug-and-play capability and possiblehigh-level fault tolerance.

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Haifeng Fan (S’09) received the B.S. degree fromthe Huazhong University of Science and Technology,Wuhan, China, in 2001, the M.S. degree from Zhe-jiang University, Hangzhou, China, in 2004, and thePh.D. degree from The Florida State University, Tal-lahassee, FL, in 2011, all in electrical engineering.

He is currently at Texas Instruments Inc., Tempe,AZ. His research interests include high-frequencyhigh-power-density dc–dc converters, topology andcontrol of bidirectional dc–dc converters, and powerelectronics for utility interface.

Hui Li (S’97–M’00–SM’01) received the B.S. andM.S. degrees in electrical engineering from theHuazhong University of Science and Technology,Wuhan, China, in 1992 and 1995, respectively. Shereceived the Ph.D. degree in electrical engineeringfrom the University of Tennessee, Knoxville, in 2000.

She is currently an Associate Professor in the De-partment of Electrical and Computer Engineering,Florida A&M University—Florida State UniversityCollege of Engineering, Tallahassee. Her researchinterests include bidirectional dc–dc converters, cas-

caded multilevel inverters, and power electronics application in hybrid electricvehicles.

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