Mitigating EMI of Powerline Communications Using Carrier

Mitigating EMI of Powerline Communications

Using

Carrier-less UWB Pulses

Der Fakultät für Ingenieurwissenschaften der

Universität Duisburg-Essen

zur Erlangung des akademischen Grades eines

Doktors der Ingenieurwissenschaften

( Dr.-Ing. )

genehmigte Dissertation

von

M.Sc. Getahun Mekuria Kuma

aus Äthiopien

Referent: Prof. Dr.-Ing. Holger Hirsch

Korreferent: Prof. Dr. rer. nat. Achim Enders

Tag der mündlichen Prüfung: 2. September 2008

Acknowledgements

I thank the Lord my God with “all my heart and with all my soul and with all my mind” for his

boundless provisions, physical and spiritual, “…wisdom and power are His” Daniel 2:21.

“Therefore, I will praise you, O LORD,, ……I will sing praises to your ame.”2. Samuel 22:50

I am very fortunate to have Prof. Dr.-Ing. Holger Hirsch as my PhD supervisor. Since the time

I applied to do my research under your supervision and during each and every single day of my

stay at the Institute of Energietransport und -speicherung (ETS), dear Prof. Hirsch, your

willingness to help me, your wide ranges of theoretical and practical knowledge and the way

you bring light to any of my difficulties and problems have helped me hugely in bringing my

studies to where it stands today. You are not only my advisor; the friendly atmosphere you

have created with all of us is what makes the working environment at ETS exceptional. I am

greatly indebted to you, I simply say: thank you so much.

I would like to greatly thank Prof. Dr. rer. nat. Achim Enders from Institute of EMC, TU

Braunschweig, for willing to be my external examiner despite his extremely tight schedules .

Thank you, Prof. Enders, for giving me this chance to be my examiner and for your important

feedbacks.

I owe a great deal of thanks to Prof. Dr.-Ing. Peter Jung, Prof. Dr.-Ing. Axel Hunger and

Prof. Dr. rer. nat. Franz J. Tegude for giving me their precious time, for the discussions I have

had with each of them and for their willingness to be member of my PhD examination

committee.

Had it not been for the full financial support I have been receiving from Deutscher

Akademischer Austausch Dienst (DAAD), it would have been unthinkable to persuade my

PhD here in Germany. I am very indebted to DAAD and personally to Dr. Ronald Weiß, Mrs.

Dagmae Eckert and Mrs. Jennifer Schenk from Referat 413 (Afrika/Sub Sahara) for extending

the generosity of DAAD to me. Thank you so much.

My many thanks are also to each staff member of ETS. Each one of them has helped me

throughout my stay at ETS.

Dr.-Ing. Fekadu Shewarega and his family, thank you for encouraging me during my studies,

thank you and I owe you too much.

My many relatives and friends back home, my father and my mother, my brothers and my

sisters, all my friends whose wish and prayers are to see and to hear that I remain healthy, and

that I be successful in my studies, I say: thank you so much.

Last but by no means the least, I say thank you to my wife Dr. Solomie Jebessa. The days were

tough, but thank God they are gone.

Getahun Mekuria Kuma

September 2008,

Duisburg, Germany

- iv -

Table of Contents

Acknowledgements........................................................................................................ ii

Table of Contents ..........................................................................................................iv

Figures.........................................................................................................................vi

Tables ........................................................................................................................vii

Glossary and Acronyms .............................................................................................. viii

1. Introduction ....................................................................................................11

1.1. Introduction: Powerline Communication ............................................................................................ 11

1.2. Why Interference is an Important Subject in PLC .............................................................................. 13

1.3. The Ultra-Wideband............................................................................................................................ 14

1.4. Governing EMC Standards for PLC ................................................................................................... 16

1.5. Current Status of PLC Technology ..................................................................................................... 17

2. Characterization of a Powerline Channel .........................................................19

2.1. Modelling ............................................................................................................................................ 19

2.1.1. Non-branched Channel ............................................................................................................. 20

2.1.2. One-branched Channel ............................................................................................................. 21

2.1.3. General n-branched Channel .................................................................................................... 22

2.2. Simulation Results .............................................................................................................................. 23

2.2.1. Non-branched Channel ............................................................................................................. 24

2.2.2. One-branched Channel ............................................................................................................. 24

2.2.3. Two-Branched Channel............................................................................................................ 26

2.2.4. Three-Branched Channel.......................................................................................................... 28

2.2.5. Effect of Position and length of Branches ................................................................................ 29

2.3. Impulse Echo Characterization ........................................................................................................... 32

2.3.1. Modelling Reflection Types ..................................................................................................... 32

2.3.2. Localization of Strong Reflection Points.................................................................................. 35

2.3.3. Line Attenuations on Symmetrical and Asymmetrical Signals ................................................ 37

2.3.4. Effect of Distribution Board on Received Signal Amplitudes.................................................. 38

2.3.5. TCTL and LCTL ...................................................................................................................... 40

3. Transmission of UWB Pulses over Powerline Channel ......................................46

3.1. Formulation of the UWB Signal Pulses .............................................................................................. 46

3.1.1. Gaussian and its Derivative Pulses ........................................................................................... 46

3.1.2. Power Spectral Density (PSD) ................................................................................................. 49

3.2. Transmission of UWB Pulse Signals .................................................................................................. 50

3.2.1. Pulse Parameters....................................................................................................................... 50

3.2.2. Improving Reception ................................................................................................................ 51

- v -

3.2.3. Simulations............................................................................................................................... 53

3.2.4. Transmission Setup .................................................................................................................. 55

3.2.4.1. Reference Transmission.................................................................................. 55

3.2.4.2. Transmission on Test Bench ............................................................................ 56

4. Theoretical Analysis of Interferences from UWB Signals ..................................59

4.1. Radiated Power Loss from 2PWN ...................................................................................................... 59

4.2. Power Spectral Density of UWB Signals............................................................................................ 61

4.3. Effect of Modulation in Minimizing Spectral Lines ........................................................................... 63

4.3.1. Un-modulated pulses ................................................................................................................ 64

4.3.2. Modulated pulses...................................................................................................................... 65

4.3.2.1. Binary Phase Shift Keying (BPSK) ................................................................... 65

4.3.2.2. Amplitude Shift Keying (ASK) ........................................................................ 65

4.3.2.3. Pulse Position Modulation (PPM)..................................................................... 68

4.3.2.4. ON-OFF Keying (OOK) ................................................................................. 69

4.3.2.5. Other Alternatives ......................................................................................... 69

4.3.3. Carrier-based Transmissions .................................................................................................... 70

4.4. UWB Signals on a Narrow-band Receiver.......................................................................................... 71

4.5. Pulse width and Measurement frequency............................................................................................ 72

4.6. Low/High PRF Region, PDCF and Effective Duty Cycle .................................................................. 74

5. EMI Measurement Setups and Results .............................................................77

5.1. Measured Signal Spectrum.................................................................................................................. 77

5.2. EMI Measurement Setups ................................................................................................................... 83

5.2.1. Disturbance Voltage Measurement Setup ................................................................................ 83

5.2.2. Radiation Measurement Setup.................................................................................................. 85

5.3. Measurement Results .......................................................................................................................... 86

5.3.1. Disturbance Voltage Measurement Result ............................................................................... 86

5.3.2. Radiated Field Measurement Results ....................................................................................... 87

5.3.2.1. Measurement Points....................................................................................... 87

5.3.2.2. The Test-Bench ............................................................................................ 87

5.3.2.3. The Measurement Results ............................................................................... 87

6. Discussions and Conclusions ...........................................................................91

6.1. Discussions and Conclusions based on Results................................................................................... 91

6.2. Topic Proposals for related Future Researches ................................................................................... 93

References and Bibliography .........................................................................................95

- vi -

Figures

Figure 1 ABCD representation of a 2PWN ................................................................................................. 20

Figure 2 Non-branched Powerline channel ................................................................................................. 20

Figure 3 One-branched Powerline channel.................................................................................................. 21

Figure 4 General n -branched Powerline channel ....................................................................................... 23

Figure 5 Transfer Function and Impulse Response of a non-branched channel .......................................... 24

Figure 6 Transfer Function and Impulse Response of a channel with one 2 m branch ............................... 25

Figure 7 Transfer Function and Impulse Response of a channel with one 5 m opened branch................... 25

Figure 8 Transfer Function and Impulse Response of a channel with two 2 m branches............................ 27

Figure 9 Transfer Function and Impulse Response of a channel with two branches, 1 m and 3 m ............. 28

Figure 10 Transfer Function and Impulse Response of a channel with three branches ................................. 29

Figure 11 Impulse Response of three-branched channel for different branch length parameters.................. 30

Figure 12 Impulse Response of three-branched channel for different branch position parameters ............... 31

Figure 13 Measurement setup for (a) Group-1 and (b) Group-2 injections ................................................... 33

Figure 14 Modelling of echoes from Group-1 and Group-2 injections ......................................................... 33

Figure 15 Impulsive Input signals (a) Symmetrical and (b) Asymmetrical................................................... 34

Figure 16 Measurement Results from (a) Group-1 and (b) Group-2 injection types..................................... 35

Figure 17 Effect of Distribution Board on (a) Group-1 and (b) Group-2 injections...................................... 39

Figure 18 Measurement Setup for (a) TCTL and (b) LCTL as defined in ITU-T G.117 .............................. 41

Figure 19 Impulsive echo measurement results of (a) TCTL and (b) LCTL................................................. 42

Figure 20 Gaussian Pulse and its first four derivative pulses ........................................................................ 47

Figure 21 Fourier Transform of the pulses in Fig.20..................................................................................... 48

Figure 22 Plots of Autocorrelation functions of 1DGP and 2DGP................................................................ 49

Figure 23 Parameters for Pulse Design ......................................................................................................... 50

Figure 24 Representation of signals a PLC channel ...................................................................................... 51

Figure 25 Convolution of received and template pulses................................................................................ 51

Figure 26 Multiplication of received and template pulses............................................................................. 52

Figure 27 Possible circuitry to minimize effects of branches from received pulses. ..................................... 52

Figure 28 Received Pulses with and without the correction circuitry on the four channels .......................... 54

Figure 29 Measurement Configuration of UWB transmission over 2PWN .................................................. 55

Figure 30 Received 2DGP over a 1 m coax................................................................................................... 56

Figure 31 Received 2DGP over the different channels and effect of the proposed correction circuitry........ 57

Figure 32 Wave propagation to and from a 2PWN ....................................................................................... 60

Figure 33 Representation of the discrete signal pulse ................................................................................... 62

Figure 34 Format of M-Array PAM UWB modulation................................................................................. 64

Figure 35 Amplitude distribution example for 4-ASK .................................................................................. 66

Figure 36 Format of M-Array PPM UWB modulation ................................................................................. 68

Figure 37 Representation of OOK Modulation ............................................................................................. 69

Figure 38 Simulated plots of Continuous and Discrete PSDs of UWB pulses .............................................. 70

Figure 39 UWB pulses as seen over NB Receiver ........................................................................................ 71

- vii -

Figure 40 Representation of "Effective Time" .............................................................................................. 75

Figure 41 Measured spectrum comparisons at 4 MHz .................................................................................. 79

Figure 42 Measured spectrum comparisons at 8 MHz .................................................................................. 80

Figure 43 Measured spectrum comparisons at 12 MHz ................................................................................ 81

Figure 44 Re-plot of Fig.41(c), Fig.42(c), and Fig.43(c) with mathematically averaged curve .................... 82

Figure 45 Setup for Disturbance Voltage Measurement................................................................................ 84

Figure 46 Setup for Radiation Measurement ................................................................................................. 85

Figure 47 Measured Results of Disturbance Voltages from the different signals ......................................... 86

Figure 48 Electric Field from un-modulated sinusoidal and un-modulated UWB ........................................ 88

Figure 49 Electric Field from un-modulated sinusoidal and BPSK modulated UWB................................... 88

Figure 50 Electric Field from un-modulated UWB and BPSK modulated UWB ......................................... 89

Figure 51 Electric Field from BPSK modulated sinusoidal and BPSK modulated UWB............................. 89

Figure 52 Electric Fields from all the four transmissions.............................................................................. 90

Tables

Table 1 Parameter values for curves in Fig.16 .......................................................................................... 36

Table 2 Average Line Attenuations of Symmetrical and Asymmetrical signals....................................... 37

Table 3 Comparing the effect of DistBrd on received signals for the two Groups of injections............... 39

Table 4 Amplitude and Timing parameters for Symmetrical inputs and Asymmetrical outputs .............. 43

Table 5 Amplitude and Timing parameters for Asymmetrical inputs and Symmetrical outputs .............. 43

Table 6 TCTL Values in Time and Frequency domains ........................................................................... 44

Table 7 LCTL Values in Time and Frequency domains ........................................................................... 44

Table 8 Summary of the maximum spectral components of the different signals..................................... 83

- viii -

Glossary and Acronyms

1DGP 1st Derivative Gaussian Pulse

2DGP 2nd Derivative Gaussian Pulse

2PW- Two-port Wired Network

AFG Arbitrary Function Generator

ASK Amplitude Shift Keying

AV Average value of the electrical quantity measured (Voltage, Power)

AWG- Additive White Gaussian Noise

BALU- BALanced to Unbalanced signal convertor

BER Bit Error Rate

BPL Broadband Power Line

BPSK Binary Phase Shift Keying

BW Band Width

Cat-5 Category-5 UTP cable

CE-ELEC European Committee for Electrotechnical Standardization, abbreviated from its French name:

Comité Europée de Normalisation Electrotechnique

CISPR The Special International Committee on Radio Interferences, abbreviated from its French

name: Comité International Spécial des Perturbations Radioélectriques

CW Continuous Wave

DC Direct Current

DistBrd Distribution Board

DPO Digital Phosphor Oscilloscope

DS-UWB Direct Sequence UWB

EM Electromagnetic

EMC Electromagnetic Compatibility

EMI Electromagnetic Interference

ESD Energy Spectral Density

FCC Federal Communications Commissions of the USA

GHz Giga Hertz

ICT Information and Communication Technology

IF Intermediate Frequency

ITE Information Technology Equipment

kHz Kilo Hertz

LCL Longitudinal Conversion Loss

LCTL Longitudinal Conversion Transmission Loss

- ix -

MAC Media Access Layer Protocol

MB-OFDM Multi-band OFDM

MHz Mega Hertz

-B Narrow Band

-RZ Non-return-to-Zero

OFDM Orthogonal Frequency Division Multiplexing

OOK ON-OFF Keying

OPM Orthogonal Pulse Modulation

PAM Pulse Amplitude Modulation

PA- Personal Area Network

PDCF Pulse Desensitization Correction Factor

PHY Physical Layer Protocol

PK Peak value of the electrical quantity measured (Voltage, Power)

PLC Powerline Communication

PPM Pulse Position Modulation

PRF Pulse Repetition Frequency

PSD Power Spectral Density

PSK Phase Shift Keying

PVC Polyvinyl Chloride

QAM Quadrature Amplitude Modulation

QoS Quality of Service

QP Quai Peak value of the electrical quantity measured (Voltage, Power)

RBW Resolution Bandwidth

SA Spectrum Analyzer

S-R Signal-to-Noise Ratio

TCL Transverse Conversion Loss

TCTL Transverse Conversion Transmission Loss

T-IS- Telecommunications-port Impedance Stabilization Network

USB Universal Serial Bus

UTP Unshielded Twisted Pair

UWB Ultra Wide Band

WUSB Wireless USB

- x -

- 11 -

1. Introduction

Contents

• Introduction: Powerline Communication

• Why Interference is an Important Subject in PLC

• The Ultra-Wideband

• Governing EMC Standards for PLC

• Current Status of PLC Technology

1.1. Introduction: Powerline Communication

The Powerline Communication (PLC) is a technology which utilizes the existing electric

networks, both the distribution network outside and the installations inside buildings, for

broadband data communication over the band of frequencies up to 30 MHz without requiring

to install additional data cables. For many decades now electrical power utilities have been

using their power transmission and distribution infrastructure for data communications aimed

exclusively at monitoring and controlling transmission lines, optimizing power generations

distributed over an interconnected power grid, remote load connecting and disconnecting,

online kilo-watt-hour-meter (kWHM) reading, online tariff setting, and other related services

aimed at improving efficiency of the power grid and customer services. Utilization of the

electric network infrastructure as a medium for commercial broadband data communications,

however, is one of the recently evolving broadband technologies and is aimed at transforming

the electrical network to serve additionally as a communication network. The main idea behind

PLC is the reduction of cost and expenditure in the realization of new telecommunication

networks [HHAH04].

This double-faceted use of the electric network is, however, not without challenges and these

challenges are related to the different features of the power line network itself:

• It is a shared channel. The number of outlets on a single socket line inside a building,

for example, explains how much a given PLC channel is shared among the consumer

loads. These consumer loads can be switched ON or OFF any time while data

communications is taking place between nodes.

• The loading condition of the channel is highly fluctuating. The different consumer

devices connected or disconnected have wide range of load sizes and hence the

Chapter.1- Introduction

- 12 -

impedance of the network fluctuates continuously as these loads are switched ON or

OFF.

• The network is characterized by unsymmetries of wide ranges: mismatches between

characteristic impedance of the line and the connected loads, mismatches caused due to

branches, wires or cables of different sizes on the same channel, unsymmetries related

to un-patterned proximity to grounding points, presence of switch gears (Distribution

Boards), presence of noises that are impulsive in nature, etc.

These challenges, however, have never prevented utilizing electric network for broadband data

transmission within a building as a Local Area Network (LAN) channel and outside a building

as an access network.

Broad-band PLC technologies are generally implemented in the following different categories:

Access PLC

The electric supply system consists of High-Voltage (HV) networks, Medium-Voltage (MV)

networks and Low-Voltage (LV) networks. In most countries the voltage levels of the HV, the

MV, and the LV networks are 110-380 kV, 10-30 kV, and 230/240 V, respectively. These

networks are within the domain of responsibility of the utility companies owning and/or

managing the respective networks. It is additional utilization of these transmission and

distribution electric network as a communication infrastructure that is commonly categorized

as Access PLC.

Access PLC is to be used to bridge long distances to avoid building extra communication links

and is considered as one of the so called “last-mile” technologies comprising of Fiber optical

cable, Digital Subscriber’s Line (DSL), broadband cable and Fixed/Mobile broadband wireless

access networks. The telecom backbone is to be coupled at the MV front end (at the HV/MV

substation) or at the LV front end (at the distribution transformer) using either Optical fibers or

Satellite communications depending on the accessibility, cost, and Quality of Service (QoS)

[HHAH04].

In-House PLC

In-House PLC is the utilization of the electric network inside a building as a Local Area

Network (LAN) channel. This enables the configuration of LAN in a building without

requiring or installing any additional unshielded Twisted Pair (UTP) cables thereby the cost of

installation and all sorts of inconveniencies related to installing new cabling inside already


- 13 -

existing buildings are hugely reduced. With its currently emerging 200 Mbps modems,

simplicity of establishing communications between nodes, and its option for a wireless

extension of the PLC service through its PLC wireless Access Point (AP), In-House PLC is

undergoing promising transformations making it optimum solution to some of the limitations

of its competing technologies.

E-energy

This is an initiative for an Information and Communication Technology (ICT)-based energy

system of the future. This is expected and aimed at providing an efficient electric power

generations, transmissions, distributions and consumptions using the ICT. The current PLC

technology is, therefore, the basis for providing an optimized service in the energy sector.

1.2. Why Interference is an Important Subject in PLC

What makes the subject of Electromagnetic Interference (EMI) an important issue in the PLC

technology can be explained very easily through discussing what is not part of the PLC channel

but part of the other communication channels for minimizing EMI and making them highly

immune to disturbances from other external sources.

Twisted-Pair Channel:

The two wires of a pair inside a standard telecommunication cable are twisted across the whole

cable length. The radiated field at a given radial distance due to an electric current +I inside

one wire of a pair is fully compensated by an equal and opposite field due to a current −I in the

other wire of that same pair. Additionally, the channel is not shared among multiple users and

hence the channel experiences relatively uniform impedance characterization during the period

of broadband data transmission. Disturbance voltage from external sources induced on one of

the wires is also fully compensated by an equal but opposite voltage induced on the other wire

of the same pair.

Broadband Cable:

The coaxial cable of Cable-TV channel used for broadband Cable is primarily intended to carry

TV signals in the range of hundreds of MHz frequencies, which is much higher than the current

broadband PLC frequency of operation. Therefore, these cables had been primarily intended to

carry signals of much higher frequencies without major degradation in providing minimum


- 14 -

interference to the environment. In addition to carrying the returning current, the major purpose

of the braided copper sheath between the inner dielectric insulator and the outer plastic sheath

is also to provide electromagnetic “blanket” to protect the radiation of the EM field from the

inner conductor to the environment and also to protect disturbance voltages from external

sources to the signal carrying inner conductor. The characteristic impedance of the cable is also

designed to provide a reflection-free transmission as much as possible.

PLC Channel:

The parallel-wire transmission channel of the electric network has only a slight twist (for

mechanical reasons) and typically has no protection sheath one finds in other channels that are

primarily intended for minimizing interferences and maximizing immunity at broad-band

frequency of operations. This is, therefore, one of the challenges facing the PLC technology: to

minimize EMI and to maximize its immunity to external disturbances. Additionally, the

existing governing standards for the EM emissions and susceptibility limit lines have been set

for those technologies and channels equipped with the features discussed earlier for minimized

EMI and maximized immunity and as a “new-comer” the PLC is also faced with the

requirement of meeting those limit lines. Even though currently there are discussions and

deliberations taking place to straighten these EMC related unfair treatment facing the PLC the

technology the PLC has also proved itself to be robust enough to win the challenges and has

since long already become marketable in countries with strict EMC regulations.

1.3. The Ultra-Wideband

Definition

The FCC 15.503 (d) defines Ultra Wideband (UWB) signal as a signal that satisfies either of

the following two conditions:

• Signal band width (BW) of 500 MHz, or

• Fractional BW of 20%

Fractional BW, here represented as bwf , is in turn defined in FCC 15.503 (c) as:

c

lh

bwf

fff

−= (1.1)

Where:


- 15 -

cf is the center frequency at which the signal has the maximum power

emission

hf and

lf are the higher and lower frequencies, respectively, at which the power

emission of the signal is 10 dB below the maximum emission.

These two requirements are not related to each other and signals can fulfill either of the two

conditions based on areas of applications of the signals.

Since the introduction of the UWB technology, however, it has been widely implemented for

wireless applications with the 500 MHz requirement due to the assignment of the 3.1 GHz to

10.6 GHz band for wireless applications without requiring any license.

The basics behind the UWB technology, which is thought to have evolved from classical high-

power pulse transmissions for radar applications is based on exploiting the advantage of a

wider BW that comes from transmission of a narrow pulse and both approaches of UWB

realizations have now evolved in to two competing technologies in the wireless applications

and each has its own advantages and disadvantages. The two major industry alliances

promoting these different approaches of wireless UWB implementations are the WiMedia

Industry Alliance promoting the Multi-band OFDM (MB-OFDM) [WiMedia] and the UWB-

Forum promoting the carrier-less and pulse-based UWB implementation based on Direct

Sequence technology (DS-UWB) [UWBFor]. Each technology has established its own PHY

and MAC layer protocols and it is yet early to be sure which technology dominates future

wireless UWB implementations. Despite the differences, however, the central point of UWB

transmission is a minimized interference between co-existing transmissions at very high data

rate.

Improved Channel Capacity

Shannon’s equation shows that the Channel Capacity (C ) increases as a function of BW faster

than as a function of Signal-to-Noise ratio ( SR )

)1(log. 2 SRBWC += (1.2)

Where the SR can again be expressed in terms of the transmitted signal power ( P ) and the

noise power spectral density (o

) as:

oBW

PSR

.= (1.3)


- 16 -

Therefore, (1.2) and (1.3) show that a given amount of increase in the channel capacity requires

either a near-linear increase in the bandwidth or an exponential increase in the transmitted

signal power.

It is also possible to re-write (1.3) in terms of the signal Power Spectral Density, PSD , as

follows:

o

PSDSR = (1.4)

This shows that, for a constant transmitted total power an increase in the BW of a signal

linearly decreases the PSD. As will be shown in Chapter 4.1, the electromagnetic radiation

from a channel in the case of radio protection (which is addressed in standards) is directly

proportional to the signal PSD rather than the transmitted total signal power.

Therefore, here is the advantage of UWB: increasing the BW linearly increases the channel

capacity and linearly decreases the radiation interference without changing the transmitted total

power.

1.4. Governing EMC Standards for PLC

The major EMC related standards for the PLC implementations are the following two:

• CISPR-22: Information Technology Equipment (ITE), Radio Disturbance

Characteristics, limits and methods of measurement. Like any other ITE, a PLC device falls

in the scope of this standard. CISPR 22 distinguishes the frequency range below 30 MHz,

where emissions are measured as conducted signals and the frequency range above 30 MHz

where measurements of the disturbance field strength are performed with antennas. Since

current PLC technologies are limited to frequencies below 30 MHz conducted

measurements and related limits should be performed according to the CISPR 22

philosophy. However, the methods described in CISPR 22 for the conducted emissions are

not yet suitable for PLC, because they either focus on mains terminals (use of measurement

networks that blocks high frequencies) or on telecommunication terminals (use of

measurement networks that are not designed for higher voltages).

• FCC-Part 15: The PLC is commonly known as Broadband Powerline (BPL) in the US

and the FCC 15 conduction and radiation limits have been amended to incorporate the BPL

interference issue. The Public Notice by FCC on August 3, 2006 states the possibility of


- 17 -

co-existence between BPL and other communication systems by imposing a reduction of

emissions by 20 dB below the normal Part 15 emission limits to provide adequate

interference protection for radio systems and also authorizing deployment of BPL

[FCCA06].

1.5. Current Status of PLC Technology

Currently there are different international and national Working Groups (WG) mandated to

undertake designing and specifying the different Layers involved in the PLC communication

system and the EMC related matters of the PLC. Some of the major working groups are:

• IEEE: The following WGs under the IEEE take care of their particular areas of

activities:

o IEEE-P1675: is responsible to the specifications and standards related to the

PLC Hardware installations and Safety issues.

o IEEE-P1775: is the IEEE WG responsible to the EMC requirements-Testing

and Measurement methods of the PLC.

o IEEE-P1901: is the IEEE WG working on the Media Access Control (MAC)

and the PHYsical (PHY) layer protocols of the PLC.

• OPERA: the “Open PLC European Research Alliance”, is a research alliance

funded from the European Union (EU) dedicated to the improving the existing, and

developing new, PLC specifications and standards. The different sub-groups of OPERA

deal with the whole spectrum of the PLC technologies: MAC and PHY Layers, EMC

related matters, Routing Protocols, Hardware, Safety, and a wide range of subjects.

OPERA works on developing new business models that integrate the PLC technology to

other existing technologies [Opera].

• HomePlug: is an Alliance of trade groups consisting of companies involved

in the development, specifications and manufacturing of PLC products and services. The

different HomePlug product specifications include the 14 Mbps HomePlug 1.0 PLC

modem and the HomePlug AV PLC modems intended for HDTV streaming over the PLC

channel.


- 18 -

• ETSI PLT is the PLC working Group under the European

Telecommunications Standards Institute (ETSI). ETSI is an independent European based

institute widely known fore its successful standardization of the Global System for Mobile

communication (GSM). ETSI-PLT is working on harmonized standards taking care of

conformity of the PLC technology with other EU Directives. This is done by studying the

technical requirements to avoid interferences with users of the radio spectrum [Etsi].

• CISPR: CISPR/I/PLT CISPR-22-PLT, Limits and methods of measurement of

broadband telecommunication equipment over power lines, revises the available CISPR 22

measurement methods intended for ITE to accommodate the special condition of the PLC

scenario.

• CE-ELEC: The sub-committee SC205A, Mains Communicating Systems,

covers the 3 kHz-148.5 kHz for home and building control applications and for utility

remote metering [SC205A], and the technical committee TC205A, Home and Building

Electronic System (HBES), covers the PLC frequency band of up to 30 MHz for various

electronic devices that are used in homes, buildings and similar environments [TC205A].

The current PLC PHYsical Layer specifications that are widely used for PLC modem design

are the following [HPAV05]:

• Frequency Band: 2 - 28 MHz

• PHY Layer data rate 200 Mbps

• Modulation OFDM

• Number of usable carriers 917

• Sub-carrier modulations varies from BPSK to 1024 QAM

- 19 -

2. Characterization of a Powerline Channel

Contents

• Modelling

o Non-branched Channel o One-branched Channel o General n-branched Channel

• Simulation Results

o Non-branched Channel o One-branched Channel o Two-branched Channel o Three-branched Channel

• Impulse Echo Characterization

o Modelling Reflection Types o Localization of Strong Reflection Points o Line Attenuation of Symmetrical and Asymmetrical Signals o TCTL and LCTL

2.1. Modelling

Modelling and characterizing the different properties of a given communication channel is one

of the primary and important properties to be done before transmitting information over the

channel. Such investigations help to understand the possible domains of the transfer function,

the impulse response, symmetry, etc, of the channel under different conceivable scenarios to

which the channel may be subjected during the actual data transmission. The PLC channel can

be modeled as a multi-path channel and can be expressed in terms of the weighting

coefficients, attenuation coefficients and delay parameters as shown in [HHAH04],

[MZKD99]. The PLC channel can also be modeled as any two-port channel through ABCD

parameters taking care of branching parameters in the modelling, and it is this approach that is

applied here.

Let us consider a 2-Port-Wired-Network (2PWN) channel represented by its ABCD matrix

parameters as shown in Fig.1. The network parameters can be expressed as:

221 BIAVV += (2.1a)

221 DICVI += (2.1b)

=

2

2

1

1

I

V

DC

BA

I

V (2.1c)

Chapter.2- Characterization of a Powerline Channel

- 20 -

Figure 1 ABCD representation of a 2PWN

For a given general network, the ABCD parameters have the following special relations:

DA = (2.2a)

1=− BCAD (2.2b)

Consider the case where Port-1 is connected to a source of internal resistance of s

R and Port-2

is connected to a load of L

R . Then, the matrix relation of (2.1c) can be re-written as :

=

−

L

ss

RV

V

DC

BA

I

RIV

/2

2

1

1 (2.2c)

Rearranging and solving for the voltage Transfer Function, one gets:

SSLL

L

RfDRRfCfBRfA

RfH

)()()()()(

+++= (2.3)

And then the impulse response of the line can be solved from:

∫= dftfjfHth )2exp()()( π (2.4)

Equation (2.3) and (2.4) show that the transfer function and the impulse response can be easily

computed if the ABCD matrix parameters of the network are known. The advantage of the

ABCD chain matrix lies in its convenience in handling computations of multiple and cascaded

sections of a given channel. The Powerline channel is one of such channels in which each

branch and each segment along the channel length is to be treated as a separate channel unit

and then cascaded together to compute the total channel matrix coefficients.

2.1.1. -on-branched Channel

Fig.2 shows the representation of a non-branched channel of length l with the source, source

and load impedance as shown.

Figure 2 Non-branched Powerline channel


- 21 -

Its ABCD chain matrix can be expressed as [TEFK03], [GMHH07]:

( ) ( )( ) ( )

=

ll

ll

γγ

γγ

coshsinh1

sinhcosh

C

C

Z

Z

DC

BA (2.5)

Where: c

Z is the Characteristic Impedance of the line, and

γ is the Propagation constant of the line:

The Characteristic Impedance and the Propagation Constant are both frequency dependent

parameters and can be solved from the following relations:

CfjG

LfjRfZ

C ππ

2

2)(

+

+= (2.6)

( ) ( )CfjGLfjRf ππγ 22)( +⋅+= (2.7)

Where: LGR ,, and C are the per unit resistance, conductance, inductance and capacitance of

the line, respectively.

Then, the transfer function of (2.3) for a non-branched channel can be written as:

( )( ) ( )( ) ( ) ( )( )SSL

C

CL

L

RlRRlZ

lZRl

RfH

γγγγ coshsinh1

sinhcosh

)(

+

++

= 2.8)

And the impulse response can then be computed from (2.4):

Therefore, (2.8) shows that the transfer function (and also the impulse response) of a non-

branched Powerline channel is influenced by the Characteristic Impedance, the Propagation

Constant, the length of the line; the source and the load impedances.

2.1.2. One-branched Channel

Next, consider the same channel of length l but has a branch of length 1brl at a distance x from

the source. The branch is connected to a load 1brR as shown in Fig.3 below:

Figure 3 One-branched Powerline channel


- 22 -

For the sake of computation of the ABCD matrix, the channel is segmented into the following

three Segments:

Segment-1: The segment of length x between the source and the branch

Segment-2: The branch of length 1brl

Segment-3: The segment of length xl − between the branch and the load

The ABCD matrices of the three Segments are:

( ) ( )( ) ( )

=

xx

Z

xZx

DC

BA

C

C

chch

chch

γγ

γγ

coshsinh1

sinhcosh

1_1_

1_1_ , for Segment-1 (2.9a)

=

1

101

1_1_1_

1_1_

eqbrbr

brbr

ZDC

BA,for Segment-2 (2.9b)

( )( ) ( )( )( )( ) ( )( )

−−

−−=

xlxl

Z

xlZxl

DC

BA

C

C

chch

chch

γγ

γγ

coshsinh1

sinhcosh

2_2_

2_2_ , for Segment-3 (2.9c)

Where the term 1_eq

Z can be computed from the following relation [TEFK03]:

)tanh(

)tanh(

11

111_

brbrC

brCbr

CeqlRZ

lZRZZ

γγ

++

⋅= (2.10)

And the ABCD matrix of the whole channel is then the cascaded matrix multiplications of the

three matrices:

=

2_2_

2_2_

1_1_

1_1_

1_1_

1_1_

chch

chch

brbr

brbr

chch

chch

DC

BA

DC

BA

DC

BA

DC

BA (2.11)

Then, the transfer function and the impulse response of the channel are computed from (2.3)

and (2.4).

2.1.3. General n-branched Channel

Consider a general n branched channel as shown in Fig.4. For a general n -branched channel:

i. the ABCD matrix of segment i of the channel, ]1,1[ +∈ ni , between the thi )1( − branch

and the thi branch is given by:

( ) ( )

( ) ( )

−−

−−

=

−−

−−

)(cosh)(sinh1

)(sinh)(cosh

11

11

__

__

iiii

c

iicii

ichich

ichich

xxxxZ

xxZxx

DC

BA

γγ

γγ

(2.12)

Where:


- 23 -

ix is the position of branch i from the source

00=x and lx

n=

+1

ii. The ABCD matrix of branch j , ],1[ nj ∈ , can also be computed from:

=

01

01

_

__

__

jeq

jbrjbr

jbrjbr

Z

DC

BA

(2.13)

Where

)tanh(

)tanh(

__

__

_

jbrjbrC

jbrCjbr

CjeqlRZ

lZRZZ

γ

γ

+

+⋅=

jbr

R _ is the load connected to branch j

jbr

l _ is the length of branch j

lbr_1

lbr_2

lbr_n

Figure 4 General n -branched Powerline channel

Hence, the ABCD matrix of the complete channel can be computed from:

=

++

++

=∏

1_1_

1_1_

1

__

__

__

__

nchnch

nchnchn

i

ibribr

ibribr

ichich

ichich

DC

BA

DC

BA

DC

BA

DC

BA

(2.14)

2.2. Simulation Results

The different mathematical formulations in the previous Section have been analyzed using

Matlab to simulate the Transfer Function and the Impulse Response of the different channel

configurations. The conductor used in the simulation is the standard 2.5 sq.mm PVC insulated

copper conductor.


- 24 -

2.2.1. -on-branched Channel

The simulated non-branched channel is a 20 m cable connected to a load matched to the

characteristic impedance of the line. Fig.5 shows the Transfer Function and magnitude of the

impulse response.

0 20 40 60 80 100-5

-4.8

-4.6

-4.4

-4.2

-4

-3.8

20.log(|H(f)|)

Frequency [ MHz ]

Transfer Function of the Channel

0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

0.5

0.6

0.7

Voltage [ V ]

Time [ ns ]

Magnitude of the Impulse Response

Figure 5 Transfer Function and Impulse Response of a non-branched channel

As expected, the Impulse Response does not have any echo depicting the ideal condition where

there are no branches and that the load connected to the channel is a perfectly matched.

2.2.2. One-branched Channel

This is the same channel as that of Section 2.2.1 but with a 2 m branch located at the middle of

the 20 m channel. The physical and electrical characteristics of the branch remain the same and

the results for the simulation are shown in Fig.6. The simulation was repeated for a 5 m branch

and the result is as shown in Fig.7 except that only the simulation result for the opened branch

is given in Fig.7 to avoid repetition.

0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

0.5

Voltage [ V ]

Time [ ns ]



- 25 -

0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

Voltage [ V ]

Time [ ns ]


0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

0.5

Voltage [ V ]

Time [ ns ]


Figure 6 Transfer Function and Impulse Response of a channel with one 2 m branch (a) opened (b) shorted and (c) matched

0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

Voltage [ V ]

Time [ ns ]


Figure 7 Transfer Function and Impulse Response of a channel with one 5 m opened branch

The following properties can be seen from Fig.6 and Fig.7:

i. The time spacing between the direct signal and the echo of the non-matched channels

is directly related to the length of the branch despite whether the branch is open

circuited or short-circuited.


- 26 -

ii. In all the three cases the magnitude of the direct signals are equal, implying that it does

not depend on the loading condition of the branch.

iii. Comparing the magnitude of the impulse response of Fig.6(a) and Fig.7, it can be seen

that the magnitude of the impulse response is independent of the length of the branch.

iv. As expected, frequency of oscillation in the transfer function is related to twice the

branch length and the dielectric property of the cable.

2.2.3. Two-Branched Channel

The same 20 m channel is investigated again for a two-branched case. The first branch is 2 m

long and is located at 25% of channel length from the source. The second branch is also 2 m

long but located at 75% of channel length from the source. Fig.8 shows the result of the

simulation. The simulation was repeated for branches of different lengths (1 m and 3 m) and

results are shown in Fig.9.

0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

Voltage [ V ]

Time [ ns ]


(a) Open

0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.1

0.2

0.3

0.4

Voltage [ V ]

Time [ ns ]



- 27 -

0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

0.25

0.3

0.35

Voltage [ V ]

Time [ ns ]


Figure 8 Transfer Function and Impulse Response of a channel with two 2 m branches (a) opened (b) shorted and (c) matched

0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

0.25

0.3

0.35Voltage [ V ]

Time [ ns ]


0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

0.25

0.3

0.35

Voltage [ V ]

Time [ ns ]



- 28 -

0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

0.25

0.3

0.35

Voltage [ V ]

Time [ ns ]


Figure 9 Transfer Function and Impulse Response of a channel with two branches, 1 m and 3 m, (a) opened (b) shorted and (c) matched

The following conclusions can be made based on Fig.8 and Fig.9

i. The echoes from equal-length branches are superimposed and hence produce an echo

stronger than the direct signal. This may lead to a wrong conclusion in making a

distinction between the two.

ii. Similar to what was said previously for a one-branched channel, varying the branch

lengths does not affect the magnitude of the direct signal response.

2.2.4. Three-Branched Channel

The number of branches is further increased to three. This time simulation of equal-length

branches are not shown here to avoid the problem related to equal-length branches as shown in

Fig.8 (a, b). The first branch is 1 m long and is located at 25% of channel length from source,

the second is 4 m long and located at 50%, and the third is 2 m long and is located at 75%.

0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

Voltage [ V ]

Time [ ns ]



- 29 -

0 20 40 60 80 100-100

-80

-60

-40

-20

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.2

Voltage [ V ]

Time [ ns ]


0 20 40 60 80 100-50

-40

-30

-20

-10

0

20.log(|H(f)|)

Frequency [ MHz ]


0 50 100 150 200 250 3000

0.05

0.1

0.15

0.18

Voltage [ V ]

Time [ ns ]


Figure 10 Transfer Function and Impulse Response of a channel with three branches, 1 m, 4 m and 2 m, (a) opened (b) shorted and (c) Matched

2.2.5. Effect of Position and length of Branches

Effects of the position and length of branches on the impulse response of the channel are

further analyzed by varying the branch position and the branch length parameters for the three-

branched channel. Fig.11 shows simulated results of the impulse response of the channel under

different combinations of branch lengths. Fig.12 also shows results when the position of the

branches are varied across the channel length. The indicated branch positions in Fig.12 are the

respective positions of the three branches from the input port expressed as a percentage of the

channel length.

Based on results of Fig.10 above and Fig.11 and Fig.12, the following conclusions can be made

for the three-branched PLC channels:

i. As discussed in Section 2.2.2 and Section 2.2.3 the amplitude of the direct pulse is

independent of both branch length and branch position.

ii. Amplitude of the direct pulse is independent of the loading conditions of the branches.

iii. Branches of equal lengths can possibly produce echoes much stronger than the

magnitude of the direct pulse.


- 30 -

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

Voltage [ V ]

(a) Shorted branches

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

Voltage [ V ]

2m, 6m, 3m

5m,16m, 3m

2m, 2m, 2m

3m,46m, 2m

Time [ sec ]

(b) Opened Branches

(b) Opened branches

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

Voltage [ V ]

(c) Matched branches

Figure 11 Impulse Response of three-branched channel for different branch length parameters


- 31 -

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

Voltage [ V ]

(a) Shorted branches

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

Voltage [ V ]

(b) Opened branches

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007

(c) Matched branches

Figure 12 Impulse Response of three-branched channel for different branch position parameters


- 32 -

2.3. Impulse Echo Characterization

In a real Powerline channel, there are other properties that need to be known in order to

characterize the unsymmetries with in the channel. There are always properties that can not be

measured or analyzed in the commonly done frequency-domain characterization of the

Powerline channel and therefore they need to be characterized in the time domain. Among

these properties are location of strong reflections, magnitude of these reflected waves, locations

of strong symmetrical-to-asymmetrical (and vice versa) signal conversions (or from

Differential Mode signal to Common Mode signal and vice versa). In addition to the transfer

function and impulse response discussed earlier, these additional characterizations of the

Powerline channel provide a better view into the channel to identify locations of strong

unsymmetries inside the network.

2.3.1. Modelling Reflection Types

To make this investigation, impulsive signals were injected to the Powerline channels at

different injection points across the channel. The magnitude of the direct pulse, the magnitude

of the echoes, the position in time of the echoes and the way these echoes from multiple

injections are aligned with respect to each other give important information in localizing the

source of these reflections with the network. For the purpose of analysis, the signal injection

points are classified into two groups, Group-1 and Group-2 as shown in Fig.13 (a, b).

• Group-1: Three signal injection points (P-1, P-2, P-3) are identified as injection

points on the same circuit line with the measurement point inside a

complex electrical network. As modelled in Fig.13 (a) such circuits are

assumed to have strong reflections coming from locations outside the

transmission path. Such reflection points are modeled as Type-A in this

Thesis.

• Group-2: Similarly, three points (P-4, P-5, P-6) are identified as injection points

on a different circuit line from the circuit line of measurement point.

These injections are assumed to be characterized by strong reflections of

Type-B or Type-C as shown in Fig.13 (b). Type-B reflection is that

which comes from a strong un-symmetry between the injection and

measurement points, whereas Type-C reflection is that which comes

from the other side of the measurement point as shown in Fig.13(b).


- 33 -

Measurements were performed in an office building according to the measurement setup of

Fig.13. The expected reflection Types for the two injection Groups are also shown [GMHH08]:

Figure 13 Measurement setup for (a) Group-1 and (b) Group-2 injections

Additionally, the types of echoes these two Groups of injections produce are assumed to have

the forms shown in Fig.14, staggered echoes from Group-1 and overlapping echoes from

Group-2 injections.

Figure 14 Modelling of echoes from Group-1 and Group-2 injections


- 34 -

It is also important to note that even though reflections from both points B and C of Fig.13 (b)

similarly produce overlapping echoes, a reverse measurement (interchanging the injection and

measurement points) helps to identify if the reflection is of Type-B or Type-C. If in the reverse

measurement the echoes are staggered similar to that of Group-1 injections, then it is of Type-

C and if the echoes still remain overlapped, then they are of Type-B and therefore the network

is assumed to have strong reflections of either Type-A or Type-B or both. The input impulsive

signals used for the measurements are shown in Fig.15 (a and b) [GMHH08].

-20

0

20

40

60

80

100

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

Voltage [ V ]

-15

-10

-5

0

5

10

15

20

25

30

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

Voltage [ V ]

Figure 15 Impulsive Input signals (a) Symmetrical and (b) Asymmetrical


- 35 -

2.3.2. Localization of Strong Reflection Points

Measurements were performed as shown in the setup of Fig.13 and the results from the two

Groups of injections are shown in Fig.16 (a and b). The parameters in the Fig.16 are to be

understood as follows:

• x

V : is the received direct signal output at the measurement point.

• ss

V : is the measured echoes after part of the signal is reflected by a nearby strong

reflection point.

• sst : is the time delay between

xV and

ssV .

-10

-5

0

5

10

15

20

25

30

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

vss1

vss2

vss3

vx1

vx2

vx3

Voltage [ V ]

vx4

vx5

vx6

-3

-2

-1

0

1

2

3

4

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

vss5

vss4

vss6

Voltage [ V ]

Figure 16 Measurement Results from (a) Group-1 and (b) Group-2 injection types


- 36 -

The delay time sst between the direct signal pulse and the different echoes of each curve are

further translated into a length ss

l of a polyethylene insulated conductor (rε = 6) to localize the

sources of the echoes as shown in Table 1. The measured values of the other parameters of the

curves are also tabulated [GMHH08].

Table 1 Parameter values for curves in Fig.16

Points x

V [ V ] ss

V [ V ] sst [µs]

ssl [m]

P-1 27.32 -8.60 0.326 19.96

P-2 29.64 -9.00 0.288 17.63

Gro

up-1

P-3 26.60 -9.80 0.254 15.55

P-4 2.36 -1.78 0.382 23.39

P-5 2.33 -2.22 0.408 24.98

Gro

up-2

P-6 3.53 -2.14 0.386 23.64

The different values of the length ss

l in the Table are to be interpreted as follows:

i. The short delay time from Point P-3 (0.254 µs) compared to that of P-1 and P-2 shows

that P-3 is nearer to the strong reflection point than P-1 or P-2 (Type-A reflection) and

therefore, point A is found at

o 15.55 m from P-3

o 17.63 m from P-2 and at

o 19.96 m from P-1.

ii. The soundness of this localization is further substantiated by the physical spacing

between the three points:

o Points P-1 and P-2 are physically spaced 2.32 m apart,

o Point P-2 and P-3 are physically spaced 2.09 m apart.

iii. From the Table,

o Points P-1 and P-2 19.96 – 17.63 = 2.33 m

o Points P-2 and P-3 17.63 – 15.55 = 2.08 m are in a good

harmony with the physical spacing between the three points.

iv. For Group-2 injections, strong reflections are coming from 23.39 m, 24.98 m, and

23.64 m, respectively, from the measuring point. Apart from a minor deviation from P-

5, which can be attributed to measurement uncertainties, these three points show that

the source of strong reflections is seems to be located at about 23.5 m from the

measurement point.


- 37 -

v. By physically inspecting these locations obtained from the echoes (15.55 m from P-3

and about 23.5 m from the measurement point), it is the electrical Distribution Board

of the office building that is located there. This has given the strong impression that

even though the electrical circuit lines inside the office building were connected to

many consumer loads such as office computers, Printers, Copier, and other electrical

devices during the measurement, it is the Distribution Board that was found to have

been the source of strong unsymmetries in the network causing strong reflections.

2.3.3. Line Attenuations on Symmetrical and Asymmetrical Signals

In this experiment the computation of the line attenuations of Symmetrical and Asymmetrical

signals in the network was analyzed as follows:

i. by computing the ratio of the received pulse to the transmitted pulse in time domain,

and

ii. by converting the time domain inputs and outputs to the frequency domain and taking

statistical average of the ratios over the entire frequency spectrum.

Table 2 shows the summary of these results.

Table 2 Average Line Attenuations of Symmetrical and Asymmetrical signals

Line Attenuation [ dB ]

Symmetrical signals Asymmetrical signals

Inject.

Points

Time Domain Freq. Domain Time Domain Freq. Domain

P-1 10 5-10 12 10-15

P-2 10 5-10 14 10-15

Gro

up-1

P-3 10 5-10 14 10-15

P-4 32 25-30 42 35-40

P-5 32 25-30 44 40-45

Gro

up-2

P-6 28 25-30 40 35-40

The following conclusions can be made based on Table 2 [GMHH08]:

i. Asymmetrical signals are attenuated much stronger than Symmetrical signals, both for

Group-1 and Group-2 injections.

ii. Even though the values in Table 2 are typical to the condition at the measurement site,

Group-2 injection points generally experience much stronger attenuation of signals


- 38 -

than Group-1 injections both for Symmetrical and the Asymmetrical signals. This fact

was re-analyzed by repeating the measurements on a different phase with different

branching topology, and the result remained the same.

iii. There is a good harmony between the transfer function values computed in the time-

domain and in the frequency domain, even though the time-domain values fall at the

lower part of the range specified by the frequency domain.

2.3.4. Effect of Distribution Board on Received Signal Amplitudes

As indicated in the previous section, the difference between the line attenuations of signals

from Group-1 and Group-2 is found to have been very considerable and hence the effect of the

Distribution Board (DistBrd), which is found out to be the source of strong reflections, is

further investigated as per the following two scenarios:

i. Scenario-1 The DB is fully operational, the injection and the measurement circuit

lines and all other circuit lines powered from the DistBrd are also operational with all

their consumer loads. The measured output under this scenario is represented as

Amplitude-I in Fig.16 for both Group-1 and Group-2 injections.

ii. Scenario-2 The feeder line to the DistBrd is disconnected, all circuits powered from

the DistBrd are disconnected and connection is established only between the injection

and measurement points. This requires no connection to the DistBrd for Group-1

injections but requires connection of the two circuit lines (the injection circuit line and

the measurement circuit line) for the case of Group-2 injections. Measurement results

of this scenario are represented as Amplitude-II in Fig.17.

Fig.17 shows only sample measurement outputs from both Groups (P-1 from Group-1 and P-4

from Group-2) since same-Group points showed almost similar results. The values of

Amplitude-I and Amplitude-II for both cases are tabulated in Table 3.


- 39 -

-15

-10

-5

0

5

10

15

20

25

30

35

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

-10

-5

0

5

10

15

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

Voltage [ V ]

Figure 17 Effect of Distribution Board on (a) Group-1 and (b) Group-2 injections

Table 3 Comparing the effect of DistBrd on received signals for the two Groups of injections

Point

Voltage

Amplitude I

[ V ]

Voltage

Amplitude II

[ V ]

Difference

[ dB ]

P-1 27.3 29.4 0.64

P-4 2.36 13.32 15.0


- 40 -

Based on Fig.17 and Table 3 the following conclusions can be made:

i. The effect caused by the DistBrd on transmission types characterized by Group-1

injections is in the range of only 0.64 dB or 7% difference.

ii. The effect caused by the DistBrd on transmission types characterized by Group-2

injections is in the range of 15 dB or 462% difference.

iii. Therefore, it is obvious that the DistBrd causes a very strong attenuation with in the

Powerline network and its influence on the different transmission-reception pairs is

also different. Due to this reason the location of the transmission-reception pairs

should be taken in to account when analyzing the Powerline network to get a better

impression of the level of attenuations the signals experience as they propagate along

the channel.

2.3.5. TCTL and LCTL

One common way of quantifying the level of unsymmetries inside a complex network is

through estimating by what level in the network are Symmetrical signals converted to

Asymmetrical signals, or vice versa, during signal transmissions. ITU-T G.117 defines

terminologies and measurement setups related to these conversion parameters meant to

quantify levels of unsymmetries inside complex networks [GMHH08].

The following Terminologies are defined in ITU-T G.117:

i. Transverse Conversion Loss (TCL) for One-port Networks, or Transverse Conversion

Transmission Loss (TCTL) for Two-port networks, defined as the level by which

Symmetrical signals are converted to Asymmetrical signals with in the network.

ii. Longitudinal Conversion Loss (LCL) for One-port networks, or Longitudinal

Conversion Transmission Loss (LCTL) for Two-port networks, defined as the level by

which Asymmetrical signals are converted to Symmetrical signals with in the network.

Fig.18 (a) and (b) show the diagrammatical representation for the computation of TCTL and

LCTL as defined in ITU-T G.117


- 41 -

(a)

(b)

Figure 18 Measurement Setup for (a) TCTL and (b) LCTL as defined in ITU-T G.117

And the expressions for these parameters are also given in ITU-T G.117 as:

2

110log20

L

T

V

VTCTL = (2.15a)

2

110log20

T

L

V

ELCTL = (2.15b)

For the purpose of characterizing the PLC channel in the time domain, measurements were

performed in accordance with the setup of Fig.18 to further analyze the network. Fig.19 (a and

b) show the impulsive echo measurement on a PLC channel for characterizing the level of

unsymmetries inside the network. These figures and their different voltage and timing

information parameters are to be understood as follows:

i. Fig.19 (a):- Symmetrical signal pulse is injected and Asymmetrical signal is

measured at the output port of the network.

• y

V is what is thought to be the original signal pulse attenuated by the internal

Symmetrical/Asymmetrical conversion factor of the Macfarlane Probe used in

the measurement.


- 42 -

• sa

V : is the Asymmetrical signal output at the measurement point due to the injected

Symmetrical signal

• sat : is the time delay between

yV and

saV indicating the location where Symmetrical

signals are strongly converted to Asymmetrical signals.

Vsa1

-1.2

-1

-0.8

-0.6

-0.4

-0.2

0

0.2

0.4

0.6

0.8

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

Vy1

Vas1

Vz1

-1.5

-1

-0.5

0

0.5

1

1.5

-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006

Figure 19 Impulsive echo measurement results of (a) TCTL and (b) LCTL

ii. Fig.19 (b):- Asymmetrical signal pulse is injected and Symmetrical signal is

measured at the output port of the network.

• z

V : is what is assumed to be the original signal received after being attenuated by

the internal Asymmetrical/Symmetrical conversion factor of the Macfarlane

probe used in the measurement.


- 43 -

• as

V : is the symmetrical signal converted from asymmetrical injection in the network.

• ast : is the time delay between

zV and

asV indicating the location where Asymmetrical

signals are strongly converted to Symmetrical signals.

These voltage amplitude and timing parameters for the different points are summarized in the

following Tables.

Table 4 Amplitude and Timing parameters for Symmetrical inputs and Asymmetrical outputs

Point sa

V [ V ] sat [µs]

P-1 -1.164 0.307

P-2 -1.227 0.293

Gro

up-1

P-3 -1.164 0.264

P-4 -0.244 --

P-5 -0.227 --

Gro

up-2

P-6 -0.333 --

Table 5 Amplitude and Timing parameters for Asymmetrical inputs and Symmetrical outputs

Point as

V [ V ] ast [µs]

P-1 -1.2 0.302

P-2 -1.2 0.281

Gro

up1

P-3 -1.55 0.240

P-4 0.216 --

P-5 0.289 --

Gro

up2

P-6 0.295 --

Both Fig.19 (a) and (b) are measurement results for injections made at point P-1 of Group-1.

Results from measurements done from injections at P-2 and P-3 also show comparatively

similar to that of P-1. Measurement results of Group-2 injections, however, exhibit very strong

and un-patterned oscillations and therefore retrieving timing parameters is found to have been

very difficult. It is for this reason that timing parameters for Group-2 injections are missing in

Table 4 and Table 5.

Comparing the time delay parameters sat of Table 4 and

ast of Table 5 with the timing

parameter sst of Table 1, it is strongly convincing that locations of strong reflections are also

locations where Symmetrical signals are converted to Asymmetrical signals and Asymmetrical


- 44 -

signals are also converted to Symmetrical signals. Or said differently, locations of strong

reflections are also locations where Modal conversions take place. This is as per the common

practice of referring the Symmetrical signal as Differential Mode (DM) signal and the

Asymmetrical signal as Common-Mode (CM) among the EMC community.

Similar to what was discussed in Section 2.3.3 for Line Attenuations, both the TCTL and

LCTL parameters are computed in Time domain as well as in Frequency domain and results

are summarized as shown in Table 6 and Table 7.

Table 6 TCTL Values in Time and Frequency domains

Points

Time Domain

(Signal ratios)

TCTL [ dB ]

Freq. Domain

transformed

TCTL [ dB ]

P-1 38 35-40

P-2 37 35-40

Gro

up-1

P-3 38 35-40

P-4 51 45-55

P-5 52 45-55

Gro

up-2

P-6 49 45-55

Table 7 LCTL Values in Time and Frequency domains

Points

Time Domain

(Signal ratios)

LCTL [ dB ]

Freq. Domain

transformed

LCTL [ dB ]

P-1 27 15-20

P-2 27 15-20

Gro

up-1

P-3 25 15-20

P-4 42 40-45

P-5 39 40-45

Gro

up-2

P-6 39 40-45

Comparing the differences between the TCTL values of Group-1 injections and Group-2

injections in Table 6, and also comparing the differences between the LCTL values of Group-1

and Group-2 injections in Table 7 it is obvious to see that these differences are related to the

source of the unsymmetries which contributed for the same range of difference in line


- 45 -

attenuations between Group-1 and Group-2 injections. This can be understood from what is

already discussed in Section 2.3.3 and Section 2.3.4.

In all of the impulsive echoes measurements being discussed in Section 2.3 the Macfarlane

probe is used for injections of both symmetrical and asymmetrical signals. This probe,

however, is calibrated for measurement resolution BW for frequencies below 30 MHz (i.e.

9 kHz) and its characteristics for measurement BW for frequencies above 30 MHz (i.e.

120 kHz) is not calibrated or standardized. Despite that, the probe has its own characteristic

parameters beyond 30 MHz range and that remains to be the property of the probe even if it is

not standardized. Additionally, any change in the resolution BW has the same effect on both

the symmetrical and the asymmetrical signals.

- 46 -

3. Transmission of UWB Pulses over Powerline

Channel

Contents

• Formulation of the UWB Signal Pulses

o Gaussian and its Derivative Pulses o Power Spectral Density (PSD)

• Transmission of UWB Pulse Signals

o Pulse Parameters o Improving Reception o Simulations o Transmission Setup

3.1. Formulation of the UWB Signal Pulses

In the wireless applications, different forms of pulses including different forms of Gaussian

pulses and Hermitian pulses are employed as UWB pulses [MGLM05], [XSMG06],

[Miller03], . Among these different families of pulses the second derivative Gaussian pulse is

much commonly used for many of the carrier-less UWB wireless transmissions. Therefore, this

same waveform of 2nd order Gaussian pulse is used in all the analysis made in this Dissertation.

3.1.1. Gaussian and its Derivative Pulses

The mathematical expressions of the Gaussian Pulse and its first four derivative pulses are

shown below [ASPO03], [BHNB04], [LZAH01], [RepJH01].

• Zero-order Gaussian pulse given by

−=

2

2

02

exp)(τt

tg (3.1)

• First Derivative Gaussian,

−==

2

2

210

1 2exp

)(

)()(

ττtt

Atd

tdgtg (3.2)

Chapter.3- Transmission of UWB Pulses over Powerline Channel

- 47 -

• Second Derivative Gaussian,

−

−==

2

2

2

2

2

2

2

2

02 2

exp1)(

)()(

τττttA

td

tdgtg (3.3)

• Third Derivative Gaussian

−

−==

2

2

2

2

433

3

03 2

exp3)(

)()(

τττttt

Atd

tdgtg (3.4)

• Fourth Derivative Gaussian,

−

+−==

2

2

4

4

2

2

4

4

4

4

04 2

exp6

3)(

)()(

ττττtttA

td

tdgtg (3.5)

Where the factors 1A , 2A , 3A and 4A are amplitude coefficients of each pulse intended to scale

its respective amplitude to some intended value, as shown in Fig.20 for a 1 V peak pulse.

1

0

1

g1 t( )

t0.7

0.15

1

g0 t( )

t0.7

0.15

1

g2 t( )

t

1

0

1

g3 t( )

t0.7

0.15

1

g4 t( )

t

Figure 20 Gaussian Pulse and its first four derivative pulses

The Fourier Transform for the zero-order pulse is given by:

( )222

0 2exp2)( ffG τππτ −= (3.6)

From the Fourier Transform property, the Fourier Transform of a higher order derivative

pulse, say an thn derivative pulse, can be computed from:

( ) )(2)(

)( 00 fGfjA

dt

tgdfG

n

nn

n

nπ=

ℑ= (3.7)

Where

[ ].ℑ represents the Fourier Transform and

nA is the amplitude coefficient


- 48 -

The plot of the Fourier Transform of the pulses in Fig.20 are shown in Fig.21

0 7.5 .107

1.5 .108

G0 f( )

G1 f( )

G2 f( )

G3 f( )

G4 f( )

f

Figure 21 Fourier Transform of the pulses in Fig.20

In this Thesis the first two derivatives of the Gaussian pulse are referred to as 1DGP (or

interchangeably )(1 tg ), and 2DGP (or interchangeably )(2 tg ), respectively, and attention is given

to this two.

The Autocorrelation functions of 1DGP and 2DGP are computed from (3.2) and (3.3) as

follows:

• For 1DGP

∫∞

∞−

−= λλλ dtggtR )().()( 1111

∫∞

∞−

−−

−

−

= λ

τλ

τλ

τλ

τλ

dtt

tR2

2

2

2

22112

)(exp

2exp

)()(

Solving the integral gives the autocorrelation of 1DGP as:

−

−=

2

2

2

2

114

exp2

12

)(τττ

π tttR (3.8)

• Similarly, for 2DGP

∫∞

∞−

−= λλλ dtggtR )().()( 2222

∫∞

∞−

−−

−

−−

−= λ

τλ

τλ

τλ

τλ

dtt

tR2

2

2

2

2

2

2

2

222

)(exp

2exp

)(11)(

Solving the integral, here too, gives:

−

+−=

2

22

3

4

224

exp4

3

4

3

16)(

ττ

ττπ

ttttR (3.9)


- 49 -

Plots of the two autocorrelation functions have been shown in the following figures:

R11 t( )

t

R22 t( )

t

Figure 22 Plots of Autocorrelation functions of 1DGP and 2DGP

It is also interesting to see that the autocorrelation functions (3.8) of 1DGP and autocorrelation

function (3.9) of 2DGP are themselves derivative pulses and are scaled version of the second

derivative and the fourth derivative Gaussian pulses, respectively.

3.1.2. Power Spectral Density (PSD)

The PSD of a signal is defined as the average power delivered to a 1 Ω load per unit frequency.

The classical approach for computing the PSD of a signal are:

i. Correlogram method: The Autocorrelation function of the signal is to be computed and the

signal PSD is the Fourier Transform of the Autocorrelation Function. For a signal )(tx with

Autocorrelation function )(tRxx

, the power spectral density function S can be expressed as:

∫∞

∞−

−= dtftjtRfSxx

)2exp()()( π

Therefore, computing the Fourier transforms of (3.8) and (3.9) gives the PSD of the 1DGP

and that of the 2DGP, respectively.

ii. Periodogram method: The signal PSD can also be computed as a statistical average of the

square of the Fourier Transform of the signal over a time period. Mathematically:

2)(

1)( fX

TfS =

Where )( fX is the Fourier Transform of the signal )(tx and T is the period of the signal.


- 50 -

3.2. Transmission of UWB Pulse Signals

Before dealing with the issue of minimized radiation from UWB signals over conductive

media, it is necessary to verify if it is really possible to transmit such narrow-pulses over

branched conductors. To prove this, 2DGP signals were transmitted over PLC channels of

different branching configurations in Laboratory environment (Test-Benches).

3.2.1. Pulse Parameters

Consider Fig.23 for the design of the appropriate pulse parameters for the 2DGP. Both

modulated and un-modulated streams of repetitive pulses of this shape were transmitted on the

PLC channel.

Figure 23 Parameters for Pulse Design

• Where:

o p

T Pulse width

o idT Idle time (also called Guard time)

Then, the pulse period T is therefore the sum of the pulse width and the guard time, leading to

the pulse repetition frequency (or rate) PRF to be given as:

idp

PRFTTT

f+

==11 (3.10)

The Idle time idT can be considered as sliding time, which means that its value is to be varied

on real time to proportionally decrease or increase the PRF by keeping the pulse width p

T

constant. Varying the PRF in-turn proportionally affects the PSD of the signal. There should be

some sort of synchronizations between the transmission and reception nodes when real-time

adjustment of the guard time is part of the transmission protocol. This helps to reliably track

the data streams.


- 51 -

3.2.2. Improving Reception

Transmission of pulse signals across Powerline channel needs to be treated differently from

wireless pulse transmissions so that the decision making circuitry inside the receiver makes

optimum decision based on the inherent characteristics of the Powerline channel [GMHH07].

Consider Fig.24 in which the transmitted pulse signal propagates across a Powerline channel of

impulse response )(th . An Additive White Gaussian Noise (AWGN) )(tn with a constant PSD

0 is assumed to have been added into the signal in the channel.

Figure 24 Representation of signals a PLC channel

Therefore, the received signal )(ty can be written as:

)()(*)()(2

tnthtgty += (3.11)

Where * indicates convolution in the time domain.

In the standard matched filtering of pulse signals, the received signal )(ty is passed through a

filter which has an impulse response of the template of the transmitted pulse )(2 tg . Let us

consider two types of filters to explore additional possibility for the impulse response of the

matched filter, if it is possible to optimize the decision making process. One filter would have

the impulse response of the transmitted pulse )(2 tg , while the other filter has an impulse

response of )(0 tg as shown below:

Figure 25 Convolution of received and template pulses

Since filtering is equivalent to convolution in the time domain the representations of Fig.25 is

meant to simplify the explanation. This solution however has the following challenges in

delivering the intended optimized decision making.


- 52 -

• The concept of matched filtering is based on the idea of optimally receiving signals in a

channel which is corrupted by additive and white noises in nature. The PLC channel,

however, has an impulse response )(th in addition to an AWGN response )(tn .

• The impulse response )(th from branched Powerline channels has multiple echoes with

time spacing and amplitude coefficients strongly dependent on the location, the

number, the length and the loading conditions of the branches. This has been shown in

the simulation results of Section 2.2

Now, consider a different approach in which the convolution in the time domain in Fig.26 is

replaced by a multiplication to time-delayed local templates of )(0 tg and )(2 tg as shown below:

Figure 26 Multiplication of received and template pulses

Simulations show that the template for the delayed version of the zero-order Gaussian pulse

produces a better result even if the transmitted pulse is a 2DGP. This is also true when the

1DGP is transmitted. Based on this understanding the Model of Fig.27 is thought to minimize

the effect of branches of the PLC channel if included as part of the Receiver circuit so that the

Receiver is able to make a better decision about the transmitted pulses [GMHH07].

Figure 27 Possible circuitry to minimize effects of branches from received pulses.

From Fig.27 and equation (3.3), one can write:

∫ ∫ −=− dtdtTtgATtgdelycdely

)()( 20 (3.12)

Where:

cA is amplitude coefficient to be set based on preference


- 53 -

delyT is the propagation delay which depends on the length and the dielectric property.

It is to be determined from the timing information before the actual data

transmission takes place.

Therefore, the received signal can be optimally computed from:

[ ][ ] )(.)()(*)()( 02 delayTtgtnthtgty −+= (3.13)

This helps to maximize the performance of the Receiver circuitry, as shown in the simulation

and the measurement results.

3.2.3. Simulations

The simulation for the received pulses with and without the effect of the correction circuitry

was done for the following four channels differing in the number of branches:

• CH-1: 2x2.5 sq. mm PVC insulated cable, 20m length, 1 m above perfect

ground surface, no branch along the channel

• CH-2: same as CH-1 but with one 2 m long branch at 10 m from signal

injection point

• CH-3: same as CH-1 but with two branches each of which are 2 m long at 5 m

and 15 m away from signal injection point

• CH-4: same as CH-1 but three branches each of which are 2 m long at 5 m,

10 m and 15 m away from signal injection point.

The results of the simulation are shown in Fig.28 for the different Channel configurations

shown above.


- 54 -

050 100 150 200

-0.5

0

0.5

1

1.5

Time [ ns ]

(b) CH-2 one Branch

without..

with correction

0 50 100 150 200

-0.6

-0.4

-0.2

0

0.2

0.4

0.6

0.8

Time [ ns ]

(c) CH-3 two Branches

without..

with correction

0 50 100 150 200

-0.4

-0.2

0

0.2

0.4

0.6

Time [ ns ]

(d) CH-4 three Branches

without..

with correction

Figure 28 Received Pulses with and without the correction circuitry on the four channels


- 55 -

3.2.4. Transmission Setup

The measurement setup for this procedure consists of the channels described in the previous

Section (CH-1, CH-2, CH-3 and CH-4) and the following equipment:

Figure 29 Measurement Configuration of UWB transmission over 2PWN

• The transmitting end consists of Tektronix AFG3101 Arbitrary Function Generator

(AFG3101) as its main component for generating the UWB pulses. The AFG3101 has

the following basic specifications:

o Sampling rate: 2.5G Giga samples per second (Gs/s)

o Frequency: 100 MHz for sinusoidal, 50MHz for Arbitrary Functions

• The receiving end consists of Tektronix DPO3104 Oscilloscope as its main component

with the following basic specifications:

o Sampling rate: 2.5 Gs/s

o Analog BW: 350 MHz

3.2.4.1. Reference Transmission

Before transmitting the pulse over the Laboratory test bench channels of different

configurations (CH-1, CH-2, CH-3 and CH-4), a reference test setup was analyzed. This

reference setup consists of the same transmitting and receiving setups similar to that intended

for the different PLC channels except that the channel is a 1 m coaxial cable with 50 Ω

characteristic impedance.


- 56 -

Figure 30 Received 2DGP over a 1 m coax

Setting the internal impedances of both the AFG3101 and the DPO3104 to 50 Ω makes this

reference transmission a perfectly matched one. Pulses to be received over branched PLC

channels are to be compared to this reference pulse received under such an ideal channel

condition. Fig.30 shows this received reference pulse over the 1 m coaxial cable together with

a “soft”-implementation of the correction circuitry of Fig.27.

The different curves in Fig.30 and Fig.31 are to be understood as follows:

• RED: This is the received pulse after the original pulse propagates across its

corresponding channel

• GREE-: This is the correction pulse to be multiplied at the multiplier circuitry of

Fig.27.

• BLUE: This is the resulting pulse which would help the decision making

circuitry to decide whether a pulse had been transmitted or not.

3.2.4.2. Transmission on Test Bench

Then the 2DGP signal is transmitted over the different channels with configurations previously

stated (Section 3.2.3).

(a) Received 2DGP over CH-1


- 57 -

(b) Received 2DGP over CH-2

(c) Received 2DGP over CH-3

(d) Received 2DGP over CH-4

Figure 31 Received 2DGP over the different channels and effect of the proposed correction circuitry

The following reasonable conclusions can be made based on the results shown in Fig.30 and Fig.31:

i. The PLC channel is not as dispersive as commonly thought for nano-second range base

band pulse transmissions. The major difference between the received pulses over the

reference channel and over the different Test bench channels is not distortion of pulse

shape (dispersion) but rather it is mainly amplitude difference.


- 58 -

ii. The improvement achieved by the proposed correction circuitry on the received pulses

can be easily seen. This is a “soft” implementation of the circuitry, which means that

template of the zero-order Gaussian pulse is multiplied to the received streams of pulses

within the digital oscilloscope on real-time. The BLUE pulses in Fig.30 and Fig.31 are

better decoded to data bits than the RED pulses during actual data transmissions.

- 59 -

4. Theoretical Analysis of Interferences from UWB

Signals

Contents

• Radiated Power Loss from 2PW-

• Power Spectral Density of UWB Signals

• Effect of Modulation in minimizing spectral lines

o Un-modulated pulses o Modulated pulses o Carrier-based transmissions

• UWB signals on a -arrow-band Receiver

• Pulse width and Measurement Frequency

• Low/High PRF Region, PDCF and Effective Duty Cycle

4.1. Radiated Power Loss from 2PW-

For a general 2PWN the radiated power loss from the network can be expressed in terms of the

different power loss terms as [GMHH06]:

othersdialcculossradPPPPP −−−= (4.1)

Where:

radP radiated power from the network

lossP total power loss inside the network, measured as a difference of input

and output power levels.

cuP copper loss inside the network

dialcP dielectric loss inside the network

othersP Other losses not accounted for by

radP ,

cuP and

dialcP

For a worst case radiation analysis the copper losses, the dielectric losses and all sorts of losses

can be neglected and all losses inside the network can be attributed to radiation. Considering

Fig.32, the total loss, and by implication the radiation loss, can then be expressed in terms of

the incident and reflected waves as:

2

2

2

1

2

1 bbaPrad

−−= (4.2)

Chapter.4- Theoretical Analysis of Interferences from UWB Signals

- 60 -

Where:

2

1a incident power at port-1

2

1b reflected power at port 1 due to mismatch at that port

2

2b power delivered to the load at port-2.

2PWNPort-1 Port-2

b1, reflected

a2=0

b2, to load

Power losses of all sorts

a1, from source

Figure 32 Wave propagation to and from a 2PWN

Re-writing (4.2):

−−=

2

1

2

2

2

1

2

12

1 1a

b

a

baP

rad (4.3a)

( )2

21

2

111 SSPPinrad

−−= (4.3b)

Where:

inP transmitted signal power

11S reflection coefficient at port-1, and

21S forward transmission coefficient

Therefore, equation (4.3b) shows that the radiated power from a wired network is characterized

by the level of the transmitted signal power and the characteristic parameters 11S and 21S of the

network itself.

The transmitted power inP is the total power injected at the input port of the network and hence

(4.3b) does not indicate the distribution of the radiated power across a given frequency band.

Therefore, it is possible to re-write (4.3b) in a way that would better represent the radiated

power as:

( )2

21

2

11 )()(1)()( fSfSfPSDfPSDsigrad

−−= (4.4)

Where:

sig

PSD PSD of the transmitted signal

rad

PSD radiated power spectral density


- 61 -

Let us see how and when the two equations (4.3b) and (4.4) are to be used:

• For a narrow-band signal the radiated power measured by a measuring receiver is the

average power over the measuring BW of the receiver and hence equation (4.3b)

applies.

• For a signal having a BW much higher than the BW of the measuring receiver equation

(4.3b) represents a fraction of the total radiated power. Therefore, it would be

reasonable to express in terms of the radiated PSD instead of the radiated power, and

hence equation (4.4) is more suitable to UWB signals than (4.3b).

Additionally, different national and international limits on radiated fields put maximum

allowable values to the radiated power over a given frequency band (which is the radiated

PSD) or to the radiated electric or magnetic field over a given frequency band. Due to this

reason it is equation (4.4) which correctly signifies the subject of this Thesis better than

equation (4.3b).

The classical approach to minimizing EMI or disturbances from signal transmissions over a

given channel characterized by 11S and 21S is either to condition the channel to optimize 11

S and

21S or to reduce the level of the transmitted power inP . Impedance matching, balancing of points

of unsymmetries, filtering, and other different techniques are some of the conditioning

techniques to optimize the values of 11S and 21S . However, as to what level can a channel be

conditioned to minimize interference depends on the nature and the complexity of the channel

itself. Some channels are relatively easier to condition than others. Powerline channels,

unfortunately, are not among those that are easy to condition as they are not primarily intended

to be communication channels. Therefore, considering a different alternative is necessary for

such cases. Equation (4.4) explains the basic reason why implementation of lower PSD signals,

as defined bysig

PSD , can be of help in minimizing EMI from a 2PWN channel.

4.2. Power Spectral Density of UWB Signals

One way to compare the radiations from two different signals across the same channel is by

comparing the distribution of the radiated power from each signal as they are transmitted

across that same channel.

Additionally, the PSD of a signal is of great importance in analyzing the interference of the

signal on a given victim circuit. This is because of the fact that the magnitude and the


- 62 -

distribution of the power spectral lines of the signal across the BW of the victim circuit depend

not only on the shape of the transmitted pulse, but also on how the pulses are modulated in

amplitude and in position at the transmitter.

Consider the following model for a general UWB transmission:

∑−

=

−=1

0

)()(f

n

nnTtatd δ ∑

−

=

−=1

0

)()(f

n

nnTtgats

Figure 33 Representation of the discrete signal pulse

Where )(tδ is the Dirac delta function. Therefore, it is possible to re-write the UWB pulse

trains as a convolution of a single pulse with a sequence of Dirac delta function, as follows:

)(*)()( tgtdts = (4.5a)

−= ∑

∞

∞−

nTtatgtsn

(*)()( δ (4.5b)

Where * indicates the convolution operation. Then the Fourier Transform of (4.5a) is:

( )∑ −== fnTjafGfDfGfSk

π2exp).()().()( (4.6)

Where:

)( fG is the Fourier Transform of a single transmitted UWB pulse )(tg

)( fD is the Fourier Transform of )(td

Note that )( fS has spikes every T/1 due to the parameters inside the summation, and its

magnitude depends on the coefficient term k

a of the modulation.

The PSD, which is the average power per Hertz, can then be expressed as:

)()(1

)(2

fSfGT

fSddss

= (4.7)

Where

2)( fG is the energy spectral density (ESD) of a single pulse

)( fS dd is the average PSD of the Dirac delta function with amplitude

coefficients k

a .

Therefore, the PSD of the UWB signal is determined by the following two factors:

i. The ESD of a single pulse, which determines the range of frequency over which the

energy of the signal is available, and


- 63 -

ii. The PSD of the Dirac delta function )( fSdd

which is determined by how the pulses are

modulated in time and in amplitude, and in turn determines the discrete spectral

components of the PSD of the UWB signal over that spectrum.

And the total energy in a single pulse is given by:

∫∞

∞−

= dttgEg

2)( in the time domain, or (4.8a)

∫∞

∞−

= dffGEg

2)( in the frequency domain (4.8b)

And, the total average power of an UWB pulse signal of period T is then given by:

TEPgg/= (4.9)

4.3. Effect of Modulation in Minimizing Spectral Lines

Consider a general M-array PAM modulated transmission format as shown in Fig.34 with a

block of data consisting of N Symbols. The symbol time s

T and the Frame duration f

T are

equal, and hence the signal can be represented as:

∑−

=

−=1

0

)()(M

k

fkkTtgats

The parameter k

a is the kth coefficient of the amplitude random variable n

a with mean a

µ and

variance 2

aσ represented as:

1210 ,......,.....,,, −=Mkn

aaaaaa

The different timing parameters of )(ts are as shown in Fig.34 and the PSD of this data

representation is given in the Literature including [Proak01], [ASPO03] [YNAM06], [Win02],

[YNAM03] as shown in equation (4.10) below.

4444 34444 2143421discrete

f

M

k ff

a

coninuous

f

a

ssT

kf

T

kG

TfG

TfS

−

+= ∑

−

=

δµσ

21

0

2

22

2

)()( (4.10)


- 64 -

1

. . .

. . .

Figure 34 Format of M-Array PAM UWB modulation

Therefore, (4.10) shows that the PSD of the transmission has both a continuous part and a

discrete part. Selection of a particular modulation technique determines whether the continuous

or the discrete part dictates the magnitude of the PSD. The performance of different

modulations commonly implemented in UWB pulse transmission on the PSD of (4.10) are

further investigated in the following sub Sections.

4.3.1. Un-modulated pulses

First, consider an un-modulated signal characterized by a constant amplitude coefficient

random variable n

a given by:

[ ]1,01 −∈∀= nan

The mean and variance of this random process is therefore:

[ ] 11 1

0

=== ∑−

=

n

nnaa

aEµ

( )[ ] 0)(1 1

0

222 =−=−= ∑−

=

n

ananaa

aE µµσ

Where

[ ].E represents the statistical expected value of a random process.

And hence from (4.10) the PSD of an un-modulated signal transmission becomes:


- 65 -

4444 34444 21discrete

f

k ff

ssT

kf

T

kG

TfS

−

= ∑

−

=

δ

21

0

2

1)( (4.11)

Therefore, the PSD of an un-modulated UWB signal consists of spectral spikes that are

separated by frequency of f

T/1 , with the envelope of these spikes following the curve of the

energy spectrum of a single pulse. The magnitudes of these spikes are proportional to the

square of the pulse repetition frequency, fPRF

Tf /1= . From the coefficients of both the discrete

part and the continuous part in (4.10) one can see that the discrete spectrum is dBfPRF

)log(10

higher than the magnitude of the continuous part. This makes un-modulated transmissions

among the maximum radiating UWB implementations [Wentz07].

4.3.2. Modulated pulses

4.3.2.1. Binary Phase Shift Keying (BPSK)

Consider again a random coefficient variable n

a representing BPSK modulation with an equal

probability of each bit and characterized by

[ ]1,01,1 −∈∀−= nan

For this random process, the mean and the variance values are:

0=a

µ

12 =a

σ

And hence the PSD of (4.10) becomes:

43421continuous

f

ssfG

TfS

2)(

1)( = (4.12)

With the spectral spikes disappearing and the PSD of the transmission equals the PSD of a

single UWB pulse. Therefore, during measurement of EMI from UWB signals, unlike carrier-

based transmissions, it is possible that interference measurements from un-modulated pulses

may be higher than that from modulated pulses.

4.3.2.2. Amplitude Shift Keying (ASK)

Similar to the BPSK, a general M-array ASK is considered in this Section. Let the UWB

transmission with M-Array ASK with its number of bits in each symbol given by M 2log=


- 66 -

has an amplitude random variable coefficient n

a . Assuming that each symbol has an equal

probability Mp /1= and distributed symmetrically on the negative and positive halfs of the

amplitude plane as shown in Fig.35 for 4-array ASK.

Figure 35 Amplitude distribution example for 4-ASK

Consider a n amplitude coefficient random variable:

[ ]1,0,,......,,,, 1210 −∈∀= − MnaaaaaaMkn

Where each discrete value k

a has a value of

12 −= kak

for all integer

−−∈

2,1

2

MMk

The mean and the variance of the random variable n

a for an M-array ASK are given by:

01 1

0

== ∑−

=

M

n

naa

Mµ (4.13)

3

)1( 22 −=

Ma

σ (4.14)

Therefore, the PSD of a UWB transmission of a general M-array ASK with each symbol

having equal probabilities and with zero mean amplitude coefficient random variable consists

of only the continuous part of (4.10) and is given by:

44 344 21continuous

f

ssfG

TM

MfS

2

2

2

)(3

)1()(

−= (4.15)

With equation (4.4) in mind, this implies that the level of EM interference from an M-array

ASK modulated UWB transmission is a product of the EM interference from a single pulse and

the variance of the amplitude coefficient random variable n

a defined by (4.14). Equations

(4.13) and (4.14) are to be understood as follows:


- 67 -

• Equation (4.13) is ensured by a symmetrical distribution of the amplitude coefficient in

the positive and negative amplitude plane, similar to the case of 4-ASK shown in

Fig.35. Additionally, each symbol should have an equal probability Mp /1= in the

signal space.

• Derivation of equation (4.14) is given below:

( )[ ] ∑−

=

−=−=1

0

222 )(1 M

n

ananaa

MaE µµσ

( )[ ] ( )∑−=

−=−=2/

2/1

222 121 M

Mk

anak

MaE µσ (4.16)

Due to the symmetry of the distribution of the amplitude over the positive and the

negative amplitude plane, (4.16) can be re-written as:

( )∑=

−=2/

1

22 122 M

k

ak

Mσ

+−= ∑∑∑

===

2/

1

2/

1

2/

1

22 1442 M

k

M

k

M

k

akk

Mσ (4.17)

The following summation properties are applied to the three sections of (4.17) as

follows:

623

23

1

2 nnnx

n

x

++=∑=

(4.18a)

2

)1(

1

+=∑

=

nnx

n

x

(4.18b)

nn

x

=∑=1

1 (4.18c)

Inserting the appropriate parameters, we get:

24

)2)(1(

12824

232/

1

2 ++=++=∑

=

MMMMMMk

M

k

(4.19a)

8

)2(2/

1

+=∑

=

MMk

M

k

(4.19b)

21

2/

1

MM

k

=∑=

(4.19c)

Combining (4.17) and the three equations of (4.19)

+

+−

++=

22

)2(

6

)2)(1(22 MMMMMM

Ma

σ and then

3

)1( 22 −=

Ma

σ (4.20)


- 68 -

4.3.2.3. Pulse Position Modulation (PPM)

Consider the case of an M-array PPM modulated UWB transmission. Fig.35 has been re-drawn

as shown in Fig.36 to accommodate parameters corresponding to PPM modulation. As one of

the commonly used modulation in the UWB applications PPM also needs to be discussed from

the EMI point of view [YSIT05].

Now, consider a PPM transmission characterized by M equally spaced positions within the

symbol time s

T . The frame time MTTsf/= occupies only one of these M possible frames and

the placement of the pulse at one of these equally spaced frame positions can be considered a

random process with mean p

µ and variance 2

pσ . Then, the signal wave form )(ts can be

expressed as:

∑∞

−∞=

+−=

n

fT

M

nktgts )( (4.21)

1

Figure 36 Format of M-Array PPM UWB modulation

In different Literatures [YNAM06], [Wentz07], [Eisen07], and for a binary PPM in

[MBGG04] the PSD of this signal for a general M-array PPM modulation with equally

probable symbols and randomly positioned in the time frame has been expressed as:

−+

−

= ∑∑∑ ∑

−

=

−

=

∞

−∞=

−

=

21

0

1

0

2

21

0

22)(

1)(

111)(

M

n

n

M

n

n

fn f

M

n f

n

f

ssfG

MfG

MTT

nf

T

nG

TMfS δ (4.22)

As can be seen from the first term of (4.22) PPM is also characterized by spectral lines spaced

at fPRF

Tf /1= . Similar to an un-modulated signal transmission discussed earlier in Section 4.3.1


- 69 -

these spectral lines follow the envelope of the spectrum of a single pulse f

TfG /|)(| 2 and are

higher than the smooth trace of the continuous spectrum by )log(10PRFf .

4.3.2.4. O--OFF Keying (OOK)

The OOK modulation is mapping of binary digits by the presence (bit 1) and absence (bit 0) of

the transmitted pulses, as shown below:

1 1 1

Figure 37 Representation of OOK Modulation

The equation for the PSD of an OOK modulated signal is correctly given in [JHMC01] as

shown in equation (4.23). Similar expressions are also shown in other sources including

[HAZC06] but these expressions lack the factor of ¼ and cannot be substantiated using (4.10)

why this parameter is missing. Computing the mean value a

µ and the variance 2

aσ for the

random variable of the amplitude distribution of an OOK modulated wave form and inserting

these values in equation (4.10) gives the same result as that of [JHMC01] and is given by.

44444 344444 2143421discrete

f

M

k ff

coninuous

f

ssT

kf

T

kG

TfG

TfS

−

+= ∑

−

=

δ

21

0

2

2

4

1)(

4

1)( (4.23)

And it has both the discrete and the continuous part. But, as discussed earlier, the amplitude of

the discrete part dominates the magnitude of the PSD.

4.3.2.5. Other Alternatives

It is also a common practice in the wireless UWB applications to modulate signals using two

different modulations to exploit the advantages both modulations provide. For example a pulse

modulated by both ASK and PPM has the advantage of a relatively smooth spectrum due to the

ASK and a better BW efficiency due to the PPM. Other techniques, such as Differential PPM

[DSJK99], delay-based BPSK [DWAC07], Pulse-based Polarity randomization [YNAM03],

are thoroughly discussed in the literature.


- 70 -

0 2.5.107

5 .107

0

0.01

0.02

0.03

0.04

UWB un-modulated Maximum at 12 MHz

Pow

er S

pect

ral

Den

sity

, PS

D [

W/ H

z ]

0 2.5 .107

5 .107

0

0.001

0.002

UWB BPSK Maximum at 12 MHz

Pow

er S

pect

ral

Den

sity

, PSD

[ W

/ H

z ]

0 2.5.107

5 .107

0

0.01

0.02

0.03

0.04


Pow

er S

pect

ral

Den

sity

, PS

D [

W/ H

z ]

0 2.5 .107

5 .107

0

0.001

0.002


Pow

er S

pect

ral

Den

sity

, PSD

[ W

/ H

z ]

0 2.5.107

5 .107

0

0.01

0.02

0.03

0.04


Pow

er S

pect

ral

Den

sity

, PS

D [

W/ H

z ]

0 2.5 .107

5 .107

0

0.001

0.002


Pow

er S

pect

ral

Den

sity

, PSD

[ W

/ H

z ]

Figure 38 Simulated plots of Continuous and Discrete PSDs of UWB pulses for maximum emissions at (a) 4 MHz, (b) 8 MHz, and (c) 12 MHz

4.3.3. Carrier-based Transmissions

For the analysis of interferences from carrier-based transmissions, it is a common practice in

the EMC measurements to transmit sinusoidal signals at a given frequency and then measure

the interference level (both conducted and radiated) at that particular frequency. Therefore, this

same approach is considered here and no further analytical expressions are required other than

what is commonly practiced [Perez95], [Paul06], [Schwab96], [FKPR05].


- 71 -

4.4. UWB Signals on a -arrow-band Receiver

It is also very important to analyze how the standard EMI receivers or any other narrow-band

receivers perform or measure the level of interferences that is coming from signals having

much larger bandwidth than their own [XSMG06]. The signal frequency spectrum and the

filter response of the measuring instrument can be represented as follows:

f

Figure 39 UWB pulses as seen over NB Receiver

Where in the above figure:

)( fHIF

is the receiver IF filter IF band-pass transfer function

)( fS is the spectrum of the received UWB signal

Therefore, the filter out put can be computed from:

)().()( fSfHfYIF

= (4.24)

In this case, in which the filter pass-band )( fHIF

has a much narrower bandwidth compared to

the spectrum:

• The spectrum )( fS is relatively constant across the IF pass band frequency range of the

filter centered at 0f .

• Therefore, the exact shape of the pulse is not important and the response of the filter to

a single pulse is the same as the impulse response of the IF filter itself, )( fhIF

, which is

the inverse Fourier Transform of )( fHIF

.

Now, let us further analyze the receiver’s output signal. Using (4.6) and (4.37)

)().().()( fDfGfHfYIF

= (4.25)

And considering )( fG to be constant across the IF bandwidth of the filter with magnitude of

)( 0fG :


- 72 -

<−

≥=

0);().().(

0);().().()(

0

0

ffDfGfH

ffDfGfHfY

IF

IF (4.26)

The IF filter impulse response )( fhIF

and the Gaussian Pulse )(tg are both real-valued time

functions, and hence both )( fHIF

and )( fG are conjugate symmetric. Additionally, for the sake

of interference evaluation it is possible to take )()( 00 fGfG −= even if the phase of the later

lags behind by π radians.

Therefore, without any loss of generality:

)(.|)(|).()( 0 fDfGfHfYIF

≅ (4.27)

And the PSD of the IF output is then:

)()()()(2

0

2fSfGfHfS

ddIFyy≅ (4.28)

Therefore, one can clearly see that the PSD of an UWB pulse received by a narrow-band EMI

measurement receiver (or a general narrow-band victim circuit) is a function of the ESD of a

single pulse at the center frequency 0f of the receiver IF filter and the pulse modulation

described by the PSD )( fSdd

. The detail analysis for the computation of )( fSdd

has been made

in Section 4.3.2 based on the implemented modulation technique.

4.5. Pulse width and Measurement frequency

In the standard measurement procedures for interferences (both conduction and radiation) from

a device operating at a given frequency 0f the Receiver’s IF center frequency should be shifted

to that same 0f to synchronize both the operation and the measurement frequencies. Also in

transmissions involving multiple carriers like the OFDM, the measurement center frequencies

should be synchronized to the center frequency of each subcarrier to measure the maximum

emission contributed by the transmission at that subcarrier.

This issue, however, is different when it comes to the measurement procedures involving UWB

signal transmission. In the following section a relation between the pulse width and the

measurement frequency is derived.


- 73 -

Consider the two commonly used UWB waveforms )(1 tg of (3.2) and )(2 tg of (3.3) for the

subject being considered. The corresponding frequency spectrums of these two waveforms are

obtained from equation (3.7) by setting 1=n for )(1 fG and 2=n for )(2 fG and also inserting the

corresponding amplitude coefficients 1A and 2A from the pulses.

( ) ( )( )2222

1 22)( feeffG τππττπ −= for )(1 tg and (4.29)

( ) ( ) ( )2222222

2 24)( feffG τππττπ −= for )(2 tg (4.30)

These spectrums have magnitude maxima at frequencies 1f and 2f , respectively, which can be

computed from:

0)(

1

1 == ff

df

fdG for )(1 tg , and (4.31)

0)(

2

2 == ff

df

fdG for )(2 tg (4.32)

Solving the above equations gives the following results:

τπ2

11 =f (4.33)

τπ2

12 =f (4.34)

Additionally, referring to )(2 tg of Fig.23 for the applications on conductive media, the energy of

the pulse is localized over the [ ]ττ 44.4,44.4− range of the time axis. A similar, but smaller pulse

range is indicated in [ASPO03]. Therefore, the pulse width p

T and the pulse width factor τ can

then be reasonably approximated as:

τ88.8≅p

T (4.35)

Combining (4.34) and (4.35) for )(2 tg gives the frequency of the maximum emission as:

pT

f2

2 ≅ (4.36a)

This is true whether a guard time g

T , as shown in Fig.23, is introduced or not; if introduced it

will only affect the total pulse period T , and hence affects the pulse repetition frequency, but

does not affect or shift the point of maximum emission 2f from what is defined by (4.36a).

This point of maximum emission 2f is a function of only the pulse factor τ . Similar analysis

for )(1 tg gives the maximum emission frequency point 1f as:


- 74 -

pT

f41.1

1 ≅ (4.36b)

For GHz range frequencies for wireless applications the values of 1f and 2f may slightly shift

from the values obtained from (4.36) due to the approximations used which would affect pulses

with a sub-nano range width p

T .

4.6. Low/High PRF Region, PDCF and Effective Duty Cycle

A-must-to-know fact during measurement of emissions from pulse based transmissions is what

is commonly known as Low- and High- PRF Region. For measurements to be done using

Spectrum Analyzer (SA), or EMI Test Receiver, the transmission is said to be “Low PRF

Region” or “High PRF Region” depending on the ratio of the Resolution Bandwidth (RBW) of

the measurement instrument to the PRF of the pulse as follows [Wentz07]:

• Low PRF Region: when RBW > 1.7 PRF, and

• High PRF Region: when RBW < 0.3 PRF (preferably RBW < 0.1 PRF)

The investigation in [AppN-150] shows that in the transition region where RBW are in the

same value range with the PRF ( )PRFRBW ≅ it is difficult to predict the response of the SA

since it is fully dependent on the modulation implemented for transmission of the pulses. For

the Low PRF region, the time spacing between pulses is large enough to allow the output of the

IF Filter inside the SA to return to zero between two consecutive pulses. Due to this, the Peak

Power (PK) and the Average Power (AV) measurements of the SA are independent of the

presence or absence of modulation and therefore both PK and AV Power measured by the SA

are the same for both modulated and un-modulated pulses, and the PK and AV Power

measurements are independent of the PRF as long as RBW > 1.7 PRF is maintained [AppN-

150].

For the High PRF Region, however, the RBW is sufficiently narrow such that the “line

spectrum” of impulses spaced at PRF is visible on the SA for a repetitive pulse train. Measured

PK and AV power levels are determined by the modulation implemented.

Another important and commonly considered during measurement of emissions from UWB

pulses is the effective duty cycle, effτ , which is defined as the width of a rectangular pulse

which has the same amplitude and the same area as the transmitted UWB pulse. This is defined

by (4.37) and shown in Fig.40 [Wentz06], [Wentz07].


- 75 -

∫=PRF

o pk

effV

dttp/1

)(τ (4.37)

Where:

)(tp is the transmitted UWB pulse and

pkV is the peak voltage of the pulse as shown in Fig.40 below. For the

remaining analysis 1=pk

V is assumed.

Equal area

eff

Vpk

time

Figure 40 Representation of "Effective Time"

The computation of this effective time effτ for a 2DGP is to be done in consistence with the

relation and the expression in equations (3.2) and (4.37) and knowing that the pulse centered at

0=t crosses the time axis at points of τ− and τ in terms of the pulse width constant:

∫∫∫−

−==PRFPRF

o

effdttpdttpdttp

2/1/1

)(2)()(τ

τ

τ

τ (4.38)

This takes in to account the whole area of the 2DGP, the positive main signal part (in the time

domain) and the two negative parts of the pulse. Inserting (4.35) in (4.38) and solving the

integral, the effective pulse width effτ in terms of the pulse width factor τ can be reasonably

expressed as:

ττ 43.2=eff

(4.39)

Which means that a 2DGP expressed by (3.3) with a period of 8.88τ has the same amplitude

and equal energy to a rectangular pulse of width 2.43τ . Hence, the transmission has an

effective duty cycle of 27%. In other words, the emission from a 100% duty cycle UWB pulse

is the same as the emission from a rectangular pulse of 27% duty cycle having same amplitude.


- 76 -

Pulse Desensitization Correction Factor (PDCF) is a controversial factor which is thought to be

used to adjust SA measured power levels characterized by the effective duty cycle to the

corresponding 100% duty cycle value. The strong understanding is however, particularly for

applications less than 1GHz frequency, not to consider the PDCF. This is an argument based

on the fact that the interference level from an UWB transmission should be considered as only

that amount received by the NB victim receiver and should not be extrapolated to what the

victim would have received had it had the same BW as the transmitting system.

Therefore, even if knowing the effective duty cycle gives a comparative impression between

the transmitted UWB pulse and an equivalent rectangular pulse of equal energy the PDCF has

no direct relevance to the subject of this work.

- 77 -

5. EMI Measurement Setups and Results

Contents

• Measured Signal Spectrum

• EMI Measurement Setups

o Disturbance Voltage Measurement Setup o Radiation Measurement setup

• Measurement Results

o Disturbance Voltage Measurement Result o Radiated Field Measurement Results

5.1. Measured Signal Spectrum

One way of comparing interferences (both conducted interferences and radiated interferences)

from a given network due to two or more different transmission protocols is done by analyzing

the voltage spectra of these different transmission protocols using a Spectrum Analyzer (SA).

This is because of the fact that level of interferences and voltage spectral components are

directly proportional [JTSI05], [Loyka00].

Therefore, continuous streams of:

• Sinusoidal waves

• Un-modulated UWB pulses, and

• BPSK modulated UWB pulses

were generated and the voltage spectra of each of these signals were measured. All of the

signals waveforms generated and measured during this investigation have the following

parameters:

• Amplitude 0 dBm (or 107dBµV on a signal generator of 50 Ω internal impedance)

• For safety of the SA, 20 dB Attenuator was connected at the input port of the SA

The generation and measurement of these signals were performed as follows:

a. Sinusoidal signal: Continuous Wave (CW) is generated using AFG3101

Function Generator. For the sake of observing the signals at few selected

frequencies 4 MHz, 8 MHz and 12 MHz points were taken as the signal

Chapter.5- EMI Measurement Setups and Results

- 78 -

frequencies. The measured voltage spectra of the CW at those three frequencies are

shown in Fig.41(a), Fig. 42(a) and Fig.43(a).

b. Un-modulated UWB: Continuous stream of pulses with mathematical equation

of (3.3) and waveform of Fig.23 (with zero idle time idT ) were generated using the

AFG3101 Function generator. The relation between the pulse width and the

frequency of maximum spectrum (4 MHz, 8 MHz and 12 MHz) were adjusted

according to the relation in (4.36a). The amplitude of the streams of pulses is the

same as that used for CW. The measured discrete spectral lines of the un-modulated

pulses at those frequencies are shown in Fig.41(b), Fig. 42(b) and Fig.43(b).

c. BPSK-modulated UWB: Continuous streams of the same pulses for the un-

modulated transmission were used except that the pulse trains were alternate

positive and inverted pulses of a pre-determined pattern. This is due to the fact that

the AFG3101 used for generating the UWB pulses is not meant to and hence does

not have the necessary speed to switch between alternate pulses for a randomized

data streams. This approach leads to a periodic sequence which contributes for the

presence of spectral lines of its own related to the measurement BW of the

Spectrum Analyzer. This can be easily noticed from Fig.41(c), Fig. 42(c) and

Fig.43(c). These spectra are re-plotted as shown in Fig.44 showing the average of

the continuous spectrum which would have been the case had the UWB pulses been

modulated by random sequences of alternate +1 and -1. During the EMI

measurements to be discussed in Section 5.2, therefore, the EMI test Receiver

measured the rather higher spectra components resulted from the pre-determined

alternate patters of stream of pulses instead of the continuous spectrum represented

by the average trace shown in Fig.44. Additionally, these discrete spectra are

relatively constant despite changes in the pulse width. The actual continuous spectra

as shown in Fig.44 (a, b, c), however, linearly decreases with a decreasing pulse

width. Therefore, the results to be shown in Section 5.2 for the BPSK modulation is

the worst case scenario and the actual emission from BPSK modulated pulses are

much lower than the results presented in that Section [Loyka98], [NLAH03]. A

summary of the maximum values of these different wave forms are shown in

Table 8.


- 79 -

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

Figure 41 Measured spectrum comparisons at 4 MHz (a) sinusoidal, (b) un-mod UWB (c) bpsk UWB


- 80 -

(a) Sinusoidal Frequency [ Hz]

(b) unmod-UWB Frequency [ Hz]

(c) bpsk-UWB Frequency [ Hz]

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

Figure 42 Measured spectrum comparisons at 8 MHz (a) sinusoidal, (b) un-mod UWB (c) bpsk UWB


- 81 -

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

Figure 43 Measured spectrum comparisons at 12 MHz (a) sinusoidal (b) un-mod UWB (c) bpsk UWB


- 82 -

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

10

20

30

40

50

60

70

80

90

0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008

(b) BPSK-UWB at 8MHz Frequency [ Hz]

(a) BPSK-UWB at 4MHz Frequency [ Hz]

(c) BPSK-UWB at 12MHz Frequency [ Hz]

averaged curve

averaged curve

averaged curve

Figure 44 Re-plot of Fig.41(c), Fig.42(c), and Fig.43(c) with mathematically averaged curve


- 83 -

The difference in the voltage Spectra of these waves at those sample frequencies are

summarized in the following Table.

Table 8 Summary of the maximum spectral components of the different signals

Measured Voltages at the three selected measurement frequencies

[ dBµV ]

4 MHz

8 MHz

12 MHz

CW 86 86 86

Un-modulated UWB 81 81 81

known +1 and -1 pattern 71 71 71 BPSK-

modulated Average of the spectrum 68 65 62

The differences in the measured data summarized in Table 8 above are indications of the

differences expected in the level of interferences when these different forms of signals are

transmitted over the same network.

5.2. EMI Measurement Setups

The amplitudes of all the signals generated and transmitted for the measurement of conducted

and radiated Interferences are the same 0 dBm or 107 dBµV (for a generator of 50 Ω output

impedance) except that the 20 dB Attenuator connected to the input of the SA during the

spectral measurements stated in Section 5.1 is no more in use during the interference

measurements.

Additionally, a signal format of alternate positive and negative sinusoidal waves were

generated and its interference also investigated together with the other three transmission

formats explained in Section 5.1. This signal has the same pattern similar to that of the BPSK-

UWB wave streams and hence it is considered as “sinusoidal BPSK” in this section.

5.2.1. Disturbance Voltage Measurement Setup

The measurement of the conducted disturbance voltage is performed according to the setup

shown in Fig.45. The Telecom-port Impedance Stabilization Network (T-ISN) is defined in

Committee Draft CISPR/I/89/CD for coupling of symmetrical and asymmetrical signals from a


- 84 -

given network. This Committee Draft defines a common-mode impedance of 150 Ω and an

LCL value of 30 dB for a T-ISN and hence the one used in the measurement setup of Fig.45

fulfills these requirements.

Figure 45 Setup for Disturbance Voltage Measurement

The different components (equipment) shown in the setup of Fig.45 are explained as follows:

• EMI-Receiver This is an EMI Test Receiver from Rhode & Schwarz (R&S). It is

compliant to the CISPR 16 standards for measurement equipment.

Measurements were performed using all its three detectors: Peak

Detector (PK), Average Detector (AV) and Quasi-Peak Detector (QP).

• AFG3101: This is an Arbitrary Function generator used to generate waveforms

loaded in to its memory. Different standard and pre-loaded wave forms

can be generated.

• T-IS-: as defined in CISPR/I/89/CD and explained earlier.

• BALU-: Macfarlane balun was used for injecting a purely balanced signal in to

the T-ISN similar to a PLC Modem. The Asymmetrical port of the balun

was terminated with a 50 Ω load.

• Laptop: Both for controlling the frequency of the AFG3101 Function generator

and for retrieving the measured data from the R&S EMI Test Receiver.

• Software: This software (Mess4) is for the synchronization of the AFG3101

Function generator and the EMI Test Receiver, for controlling the

frequency of the Generator, for setting the measurement point by the

EMI Test Receiver and also for retrieving the measured data to the

laptop.

• Saftey-box This is a High Pass Filter for the protection of the AFG3101 Function

Generator from being damaged by the 50 Hz, 220 V line.


- 85 -

5.2.2. Radiation Measurement Setup

The measurement setup for the radiated field measurements is as shown in Fig.46 below.

AFG3101

DPO4034

Oscilloscope for

Observing Waveform

The 2-port wired

Channel

antenna point

2PWNS11

S12

S21

S22

GPIB

Cables

AFG3101

Arbitrary Function Generator

EMI-Receiver Laptop for Measurement

and controlling signal

generation

BALUN

50Ω

termination

Sym

Figure 46 Setup for Radiation Measurement

The following equipment and devices are used for the measurement:

• EMI-Receiver: This is the same R&S EMI Test Receiver described in the

previous Section. Radiated field measurements were performed using all

its three detectors: PK, AV and QP.

• AFG3101: Same as explained in the previous Section.

• DPO4034: A Digital Phosphorous Oscilloscope for observing the transmitted

signals. This is not mandatory but only helps to observe transmissions.

• Antenna: The Magnetic Loop antenna is calibrated for measurements below

30 MHz. Even if it is not placed in a strictly far-field region for all the

frequencies of measurements, such radiated field measurements are

commonly practiced.

• BALU-: The same Macfarlane probe as previously stated. But here the Sym port

of the balun was terminated by 50 Ω and signal injections were

Asymmetrical.

• Laptop Both for controlling the frequency of the AFG3101 Function generator

and for retrieving the measured data from the R&S EMI Test Receiver.

• Software Same as explained in the previous Section.


- 86 -

5.3. Measurement Results

The methodology followed in the measurements is the comparison of the different

interferences (Disturbance Voltage, and radiated Field) between the following four types of

wave forms:

• Un-modulated sinusoidal wave

• BPSK modulated sinusoidal wave

• Un-modulated UWB pulse

• BPSK modulated UWB pulse

Since the carrier-based transmissions employ the sinusoidal signals as carrier signal the

difference in the level of interference between the sinusoidal and the UWB signals is a proof of

an achievable reduction in interference by employing a carrier-less UWB transmission instead

of the carrier-based transmission. Therefore, much attention is not to be given to the magnitude

of interference from each waveform since that depends on the measurement scenario and

network configuration.

5.3.1. Disturbance Voltage Measurement Result

Fig.47 shows the disturbance voltage measurements of the four different waveforms using the

QP detector. The PK and AV detectors also show relatively similar results and hence are not

plotted. Discussions on results are made in Chapter 6.

40

45

50

55

60

65

70

75

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Conducted Disturbance Voltage [ dB

µV

]

Figure 47 Measured Results of Disturbance Voltages from the different signals


- 87 -

5.3.2. Radiated Field Measurement Results

In addition to the disturbance voltage measurement described in the previous Section,

measurements of radiated fields were made on a Laboratory Test Bench Powerline channel.

The radiated magnetic field measurement was done using Magnetic Loop Antenna and the

equivalent electric field is computed based on the Antenna Factor and the 51.5 dB Ω

conversion parameter between measured magnetic field and computed electric field.

5.3.2.1. Measurement Points

For the measurement of radiated field, different measurement points along the length of the

channel have been set as antenna points. The results obtained from measurements at these

different points are relatively the same and hence only results from one of these points (P1) are

presented here.

5.3.2.2. The Test-Bench

The channel selected for the measurement of the radiated filed is CH-4 of Section 3.2.3 at a

height of 1.5 m above perfect ground and with asymmetrical signal injection.

5.3.2.3. The Measurement Results

The measured results are displayed in the following figures, discussions and conclusions based

on the figures follow in Chapter 6.

The results shown in Fig.48-Fig.52 show graphs of electric field strengths converted from the

measured radiated magnetic fields using the 51.5 dB Ω as said earlier.


- 88 -

55

60

65

70

75

80

85

90

95

100

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Figure 48 Electric Field from un-modulated sinusoidal and un-modulated UWB

45

50

55

60

65

70

75

80

85

90

95

100

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Figure 49 Electric Field from un-modulated sinusoidal and BPSK modulated UWB


- 89 -

45

50

55

60

65

70

75

80

85

90

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Figure 50 Electric Field from un-modulated UWB and BPSK modulated UWB

45

50

55

60

65

70

75

80

85

90

95

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Figure 51 Electric Field from BPSK modulated sinusoidal and BPSK modulated UWB


- 90 -

Electric Field Strength [ dB

µV

/m ]

50

55

60

65

70

75

80

85

90

95

100

0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007

Figure 52 Electric Fields from all the four transmissions

- 91 -

6. Discussions and Conclusions

Contents

• Discussions and Conclusions based on results

• Topic proposals for related future researches

6.1. Discussions and Conclusions based on Results

Different sorts of findings from the different mathematical and experimental investigations

have been illustrated in the different Sections and sub-Sections of the Chapters of the

Dissertation about the certainty of the case-in-point: Mitigating EM Interferences of

Powerline Communications using Carrier-less UWB pulses. Full texts of those findings have

been placed where they would maintain the smooth flow of the particular subject discussed at

those sub-sections.

In this Section, therefore, a review of those findings and a discussion of measurement results of

Chapter 5 are summarized. Other points worth mentioning to cement the different components

are also part of the summary made here.

1. It is necessary that the technology of UWB communication system is to be understood

as one based on the definition outlined in FCC 15.502 where either of the 500 MHz or

the 20% fractional BW requirements is fulfilled. The choice between the two

requirements depends on the application area and the limitations related to band of

operations.

2. Signal BW can be increased in either of the following two ways:

a. In carrier-based transmissions: by increasing the number of carriers

b. In carrier-less transmissions: by transmitting very-narrow pulses

3. The main advantage of increased BW is a reduced interference to a co-existing

transmission without compromising the total transmitted power. This is very necessary

in high data rate communications such as broadband PLC.

Chapter.6- Discussions and Conclusions

- 92 -

4. The source of strong un-symmetry inside the Powerline channel is the Distribution

Board of the network. Strong reflections, attenuations and modal conversions take place

due to the strong un-symmetry introduced by the Distribution Board. Therefore, care

should be taken to consider the position of the two communicating ends relative to the

Distribution Board when characterizing Powerline channels.

5. The radius of coverage in wireless UWB transmission is very short (approx. 3 m) due to

the sub-nano range pulse width. Transmission of the pulses over a conductive media

with pulse width compatible to the frequency band of PLC operation, however, is not

characterized by such shortcomings. The plots of received pulses discussed in Section

3.2.4 are for pulses received over 20 m PLC channels.

6. Modulation formats resulting into smooth spectral components are to be used for a

highly reduced level of interferences from UWB transmissions.

7. Results summarized in Table 8 and Fig.41 – Fig.44 and Fig.47 - Fig.52 show that the

reduction in interference is huge if a carrier-less UWB pulse transmission is

implemented with wisely selected modulation formats instead of carrier-based

transmissions.

8. Therefore, the findings of this Dissertation can be further converged in two sentences:

• It is possible to transmit nano-range UWB pulses over PLC channels.

• Reduction of Interferences by more than 15 dB can be achieved over the

PLC band of operations below 30 MHz.

Chapter.6- Discussions and Conclusions

- 93 -

6.2. Topic Proposals for related Future Researches

UWB technology is still undergoing important phase of transformations even for the wireless

applications. The two approaches of achieving that “ultra-wide” band are either increasing the

number of sub-carriers or transmitting sub-nano range pulses in baseband. These are competing

and seemingly un-reconciling approaches towards the common goal: a huge reduction in

interferences between co-existing transmissions of high data rates. To what extent the number

of sub-carriers can be increased also depends on the available band of operation. The PLC

technology is limited to a maximum of 30 MHz range and the need for a reduced interference

level, therefore, requires looking for a solution from any sort of UWB implementations.

Therefore, for the sake of exploiting the potential of UWB pulses for wired channel

applications in general, and for PLC in particular, researches in the following directions are

believed to contribute for the maturity of the approach:

1. Performance measurement of UWB signals on wired channels

2. Methods of Interference Measurement for UWB signals over PLC band of operation

3. Conducted Interference measurements of UWB signals

4. Injections of UWB signals to wired channels

5. Looking for a possibility of expanding the PLC band of operations

6. Algorithmic and Hardware realization of wired UWB

7. Integration of wireless UWB and PLC protocols both for the wireless extension of

existing PLC transmissions and for a wired extension of existing UWB protocols. The

emerging wireless USB (WUSB) in a digital-home requires researches aimed at

integrating UWB technologies and PLC technologies.

- 94 -

References and Bibliography

- 95 -


[App--150] Application ote 150-2, Agilent Spectrum Analyzer Series

[ASPO03] H. Sheng, P. Orlik, A.M. Haimovich, L.J. Cimini, “On the Spectral and Power Requirements for Ultra-Wideband Transmission,” IEEE International Conference on Communications, Alaska, USA, May 2003.

[BH-B04] Bo Hu, Norman C. Beaulieu, “Accurate Evaluation of Multiple-Access Performance in TH-PPM and TH-BPSK UWB Systems,” IEEE Transactions on Communications, Vol.52, No.10, October 2004.

[DMT-07] Daniel Mänson, Tony Nilsson, Rajeev Thottappillil, “Propagation of UWB Transients in Low-Voltage Installation Power Cables,” IEEE Transaction on Electromagnetic

Compatibility, Vol.49, No.3, August 2007. [DSJK99] Da-shan Shiu, Joseph M. Kahn, “Differential Pulse Position Modulation,” IEEE

Transaction on Communications, Vol.47, No.8, August 1999. [DWAC07] David D. Wentzloff, Anantha P. Chandrakasan, “Delayed-based BPSK for Pulsed UWB

Communication,” IEEE International Conference on Acoustics, Speech, and Signal Processing (ICASSP), Hawaii, USA, April 2007.

[Eisen04] Michael Eisenacher, Optimierung von Ultra-Wideband Signalen, (German) PhD Dissertation, Institut für Nachrichttechnik der Universität Karlsruhe (TH), Karlsruhe, Germany, 2004.

[Etsi] www.etsi.org/website/technologies/powerline.aspx [FCCA06] Press Release, Federal Communications Commission, “FCC Adopts Memorandum

Opinion and Order on Broadband Over Powerlines to Promote Broadband Services to

All Americans,” Washington, DC, US, August 3, 2006. [FKPR05] Florian Krug, Peter Russel, “Quasi-Peak Detector Model for a Time-Domain

Measurement System,” IEEE Transaction on Electromagnetic Compatibility, Vol.47. No.2, May 2005.

[GMHH06] Getahun Mekuria, Holger Hirsch, “EMC Analysis of Pulse Transmission on a Wired Network,” 18th International Wroclaw Symposium and Exhibition on Electromagnetic

Compatibility, Wroclaw, Poland, June 2006. [GMHH07] Getahun Mekuria, Holger Hirsch, “UWB Pulse Transmission over Powerline Channel,”

IEEE-International Symposium on Powerline Communications and its Applications

(ISPLC), Pisa, Italy, March 2007. [GMHH07B] Getahun Mekuria, Holger Hirsch, “Powerline Communication: Untapped Broadband

Infrastructure in Developing Countries,” Invited Paper, World IT Forum (WITFOR), Addis Ababa, Ethiopia, August 2007.

[GMHH08] Getahun Mekuria, Holger Hirsch, “Charakterisierung der elektrischen Unsymmetrie von komplexen Leitungsnetzen im Zeitbereich,” (German), Elektromagnetische

Verträgleichkeit-2008 (EMV2008), Düsseldorf, Germany, February 2008. [HAZC06] Hüseyin Arslan, Zhi Ning Chen, Marai-Gabriella Di Bendetto, Ultra Wideband Wireless

Communication, pp.92, Wiley-Interscience publication, USA, 2006. [HHAH04] Halid Hrasnica, Abdelfatteh Haidine, Ralf Lehnert, Broadband Powerline

Communications etwork Design, John Wiley & Sons, Ltd, 2004. [HPAV05] “HomePlug AV White Paper” [JHMC01] J. R. Hoffman, M.G. Cotton, R.J. Achatz, R.N. Statz, R.A. Dalke, Measurements to

Determine Potential Interferences to GPS Receivers from Ultra-wideband Transmission

Systems, NTIA Report 01-384, US Department of Commerce, February 2001. [JSSM06] John Santhoff, Steven A. Moore, Ultra-Wideband Communications Through Local Power

Lines, US Patent No. US 7027483 B2, April 11, 2006 [JSSM06B] John Santhoff, Steve Moore, Ultra-Wideband Communication Through a Wire Medium,

US Patent No. US 7099368 B2, August 29, 2006. [JTSI05] J. I. Takada, S. Ishigami, J. Nakada, E. Nakagawa, T. Yasui, “Measurement Techniques

of Emissions from Ultra-Wideband Devices,” Institute of Electronics, Information and Communication Engineers (IEICE) Transactions, Fundamentals, Vol.E88-A No.9, September 2005.

[Kisser01] William A. Kisser, The Temporal and Spectral Characteristics of Ultra-wideband Signals, NTIA Report 01-383, US Department of Commerce, USA, January 2001.


- 96 -

[KPYK06] Keun-joo Park, Hyun-chin Kim, Young-kwang Seo, Yeong-bae Yeo, System of

Transmitting Wire-line Single Band Orthogonal Division Multiplexing based Ultra-

wideband Signal over Pipeline Carrying CATV Broadcasting Signal, US Patent No. US 2006/0262770 A1, November 23, 2006.

[KSDM04] K. Siwiak, D. McKeown, Ultra-Wideband Radio Technology, John Wiley & Sons, Ltd, 2004.

[Loyka00] Sergey Loyka, “Nonlinear EMI Simulation of an AM Detector at the System Level”, IEEE Transaction on Electromagnetic Compatibility, Vol.42, No.1, February 2000.

[Loyka98] Sergey Loyka, “On Accuracy of Numerical EMC/EMI Modeling over a Wide Frequency Range,” International Symposium on EMC, Rome, Italy, September 1999.

[LZAH01] Li Zhao, Alexander M. Haimovich, “Interference Suppression in Ultra-Wideband Communications,” Conference on Information Sciences and Systems, the John Hopkins University, Mrach 2001.

[MBGG04] Maria-Gabriella Di Benedetto, Guerino Giancola, Understanding Ultra Wide Band Radio

Fundamentals, pp.107, Prentice Hall Publications, USA, 2004. [MBTK06] Maria-Gabriella Di Bendetto, Thomas Kaiser, Andreas F. Molisch, Ian Oppermann,

Chrstian Politano, Domenico Porcino, UWB Communication Systems a Comprehensive

Overview, Hindawi Publications Corporation, 2006. [MGLM05] M. Ghavami, L.B. Michael, R. Kohno, Ultra Wideband Signals and Systems in

Communication Engineering, John Wiley & Sons, Ltd, April 2005. [Miller03] Leonard E. Miller, Why UWB? A Review of Ultrawideband Technology, Report to

NETEX Project Office, DARPA, USA, April 2003 [MZKD99] Manfred Zimmermann, Klaus Dostert, “A Multi-path Signal Propagation Model for the

Powerline Channel in the High Frequency Range,” International Symposium on

Powerline Communications and its Applications (ISPLC), Lancaster, UK, April 1999. [-LAH03] Nikolaus H. Lehmann, Alexander M. Haimovich, “The Power Spectral Density of a Time

Hopping UWB Signal: A Survey,” IEEE Conference on Ultra Wideband Systems and

Technologies (UWBST), VA, USA, November 2003. [Opera] www.ist.opera.org [Paul06] Clyton R. Paul, Introduction to Electromagnetic Compatibility, 2nd. Ed. John Wiley &

Sons, Inc, 2006. [Perez95] Reinaldo Perez, Handbook of Electromagnetic Compatibility, Academic Press, 1995 [Proaki01] John G. Proakis, Digital Communications, 4th ed., pp.205, McGraw-Hill Publications,

2001. [Proph06] Graham Prophet, Powerline’s Other data channel, EDN Europe, May 2006 website:

http://www.edn.com/article/CA6294227.html [Rappa02] Theodore Rappaport, Wireless Communications, 2nd edition, Printice Hall, USA, 2002. [RepJH01] Report: UWB-GPS Compatibility Analysis Project, Strategic Systems Department, The

Johns Hopkins University, March 2001. [RJD-03] Robert T. Johnk, David R. Novotny, Chriss A. Grosvenor, Nino Canales, Jason G.

Veneman, “Time-Domain Measurements of Radiated and Conducted Ultrawideband Emissions,” IEEE Conference on Ultra Wideband Systems and Technologies (UWBST), VA, USA, Nov. 2003.

[SC205A] http://www.cenelec.eu/Cenelec/CENELEC+in+action/Horizontal+areas/ICT/SC205A.htm [Schaef05] Werner Schaefer, “Measurement of Impulsive Signals with Spectrum Analyzer or EMI

Receiver,” IEEE-International Symposium on Electromagnetic Compatibility, EMC2005, August 2005, Chicago, USA

[Schwab96] Adolf J. Schwab, Elektromagnetische Verträglichkeit, Springer-Verlag, 1996 [TC205] http://www.cenelec.eu/Cenelec/CENELEC+in+action/Horizontal+areas/ICT/TC205.htm [TEFK03] Tooraj Esmailian, Frank R. Kschischang, P.Glenn Gulak, “In-building Power lines as

high-speed communication channels: channel characterization and a test channel ensemble,” International Journal of Communication Systems, 16:381-400, 2003.

[UWBFor] DS-UWB Alliance website: http://www.uwbforum.org [Wentz06] David D. Wentzloff, “Gaussian Pulse Generators for Subbanded Ultra-Wideband

Transmitters,” IEEE Transactions on Microwave Theory and Techniques, Vol.54, No.4, April 2006

[Wentz07] David D. Wentzloff, Pulse-Based Ultra-Wideband Transmitters for Digital

Communication, PhD Dissertation, Massachusetts Institute of Technology (MIT), MA, USA, June 2007.

[WiMedia] MB-OFDM Alliance website: http://www.wimedia.org [Win02] Moe Z. Win, “Spectral Density of Random UWB Signals,” IEEE Communications

Letters, Vol.6, No.12, December 2002.


- 97 -

[XSMG06] X. Shen, M. Guizani, R.C. Qiu, T. Le-Ngoc, Ultra-Wideband wireless communications

and networks, John Wiles & Sons, Ltd, July 2006. [Y-AM03] Yves-Paul Nakache, Andreas F. Molisch, Spectral Shape of UWB Signals Influence of

Modulation Format, Multiple Access Scheme and Pulse Shape, Mitsubishi Electric Research Laboratory (MERL), TR-2003-40, May 2003.

[Y-AM06] Yves-Paul Nakache, Andreas F. Molisch, “Spectral Shaping of UWB Signals for Time-Hopping Impulse Radio,” IEEE Journal on Selected Areas in Communications, Vol.24, No.4, April 2006.

[YSIT05] Yasuo Suzuki, Ichihiko Toyoda, Masahiro Umehira, “Interference Analysis from Impulse Radio UWB Systems Using Simple Signal Models,” Institute of Electronics, Information and Communication Engineers (IEICE) Transactions, Fundamentals, Vol.E88-A No.11, November 2005.

[YWXD07] Yue Wang, Xiaodai Dong, Ivan J. Fair, “Spectral Shaping and NBI Suppression in UWB Communications,” IEEE on Wireless Communications, Vol.6, No.5, May 2007.

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CURRICULUM VITAE Full Name Getahun Mekuria Kuma Date of Birth 01.12.1969 Place of Birth Illubabor, Ethiopia Marital Status Married Permanent Email: [email protected]

Education April 2005 – Sept. 2008: Dr.-Ing. (PhD) in Electrical Engineering

Energietransport und –speicherung, Universität Duisburg-Essen, Duisburg, Germany Title: “Mitigating EMI of Powerline Communication using

Carrier-less UWB pulses” Supervisor: Prof. Dr.-Ing. Holger Hirsch Sept.1997 – Jul.1999 M.Sc.in Electrical Engineering Major in Communication Engineering Addis Ababa University, Addis Ababa, Ethiopia Sept.1987 – Jul.1992 B.Sc.in Electrical Engineering Addis Ababa University, Addis Ababa, Ethiopia

Additional Certifications

CCAI: Cisco Certified Academy Instructor (No. 3266948CCNA) Port Elizabeth Technikon University, Port Elizabeth, South Africa

CCNA: Cisco Certified Network Associate (No. CSO10787298)

Employment History:

Oct.2002 -- Sept.2004 Lecturer Dept. of Electrical and Computer Eng., Faculty of Technology, Addis Ababa University, Addis Ababa, Ethiopia

Mar.2000 -- Aug.2002 Head of Engineering Department Siemens Ethiopia Ltd., Addis Ababa, Ethiopia Oct.1996 -- Nov.1997 Hardware Engineer GATE Private Limited Company, Addis Ababa, Ethiopia Jul.1992 -- Sept.1996: Electrical Distribution Network Design Engineer

Ethiopian Electric Light and Power Authority, Jimma, Ethiopia.

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Publications 1. Getahun Mekuria, Holger Hirsch, “Measurement of Radiated Field from UWB signal over

Powerline Channel, IEEE-EMC Symposium, Detroit, USA, 18-22 August 2008. 2. Getahun Mekuria, Holger Hirsch, “Charakterisierung der elektrischen Unsymmetrie von

komplexen Leitungsnetzen im Zeitbereich,” (German), Elektromagnetische Verträgleichkeit-2008 (EMV2008), Düsseldorf, Germany, February 2008.

3. Getahun Mekuria, Holger Hirsch, “Powerline Communication: Untapped ICT Infrastructure in the

Developing Countries,” World IT Forum (WITFOR 2007), Addis Ababa, Ethiopia, 22 – 24 August 2007 (Invited Paper).

4. Getahun Mekuria, Holger Hirsch, “UWB Pulse Transmission over Powerline Channel”, IEEE-

International Symposium on Powerline Communication and its Applications (ISPLC), Pisa, Italy, 26-28 March 2007.

5. Getahun Mekuria, Holger Hirsch, “EMC Analysis of Pulse Transmission on a Wired Network”,

18th International Wroclaw Symposium and Exhibition on Electromagnetic Compatibility, Wroclaw, Poland, 28-30 June 2006.

6. Getahun Mekuria, Eneyew Adugna, Deva Rajan, “Information Secrecy and Public-Key

Cryptography.” Zede-Journal of Ethiopian Engineers and Architects, Addis Ababa, Ethiopia, Dec 2001.

7. Getahun Mekuria, Eneyew Adugna, Deva Rajan, “Cryptography-an Ideal Solution to Privacy,

Data Integrity & Non-Repudiation.” Zede-Journal of Ethiopian Engineers and Architects, Addis Ababa, Ethiopia, Dec 1999

Documents

Mitigating EMI of Powerline Communications Using Carrier