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KING SAUD UNIVERSITY

COLLEGE OF ENGINEERING

RESEARCH CENTER

Final Research Report No. 425/4

Electronically tuned antenna for third generation mobile

communication

By

Dr. Abdel Fattah Sheta

RabiII 1426 H

May 2005G


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Table of Contents

Page

Acknowledgment 3

List of Figures 4Abstract (English) 5

Abstract (Arabic) 6

Chapter 1: Compact Microstrip Antennas: Characteristics and Limitations 7

1.1 Introduction 7

1.2 Challenges and Fundamental Limitations 8

1.3 Tuning Concept Solution 10

Chapter 2: Active Devices Used in RF Tuning

122.1 Introduction 12

2.2 Characteristics of Varactor Diodes 12

2.3 Characteristics of PIN Diodes 16

Chapter 3: Tunable Antenna Techniques and Mobile Phone RF SystemArchitecture

20

3.1 Introduction 20

3.2 Varactor Based Tunable Microstrip Antennas 21

3.3 Switching Based Tunable Microstrip Antennas 233.4 The proposed RF System Architecture 25

Chapter 4: Dual Band Tunable Antenna For Cellular Phone 27

4.1 Introduction 27

4.2 Effect of Varactor Diodes in Microstrip Circuits 28

4.3 Compact Tunable Microstrip Antenna 31

4.4 Experimental Results 39

Chapter 5: Conclusions and Recommendation 41

References 43

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Acknowledgment

The authors would like to acknowledge the assistance and the financial support provided

by the Research Center in the College of Engineering at King Saud University for this

project under grant number 4/425.

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LIST OF FIGURES

Page

Figure 1.1 Antenna within a sphere of radius r. 9

Figure 1.2 Fundamental limit of Q versus antenna size kr < 1 , k = 2/ = /c. 9

Figure 2.1 Varactor characteristics. 18Figure 2.2 Varactor diode equivalent circuit. 18

Figure 2.3 Physical structure of a PIN diode. 18

Figure 2.4 PIN diode equivalent circuit. 19

Figure 2.5 I-V characteristics of a PIN diode. 19

Figure 3.1 Electronic tuning of half-wavelength microstrip antenna. 22

Figure 3.2 Electronic tuning of quarter-wavelength shorted microstrip antenna. 22

Figure 3.3 Diode tunable PIFA. 22

Figure 3.4 Frequency tuning by switching techniques. 24

Figure 3.5 The conventional dual-band full duplex RF front end. 26

Figure 3.6 The proposed dual-band full-duplex RF front end based on a tunable antenna

pair.

26

Figure 4.1 Capacitance-Voltage relation of SMTD3001. 29

Figure 4.2 Electrical length equivalence of a varactor located at the end of the line. 29

Figure 4.3 The variation of effective electrical length of the varactor diode SMTD3001

against microstrip line width on Duroid substrate with r = 2.2 and 1.57 mmthickness.

30

Figure 4.4 L and inverted L shape antenna. 32

Figure 4.5 The proposed dual-band tunable microstrip antenna. 32

Figure 4.6 1 versus (2 + eff) for different values of K. 35

Figure 4.7 Layout of the dual-band tunable antenna designed for GSM applications. 35

Figure 4.8 Simulation results (S11) of the dual band proposed antenna at the lower

frequency band (GSM-900 MHz) for various values of reverse bias voltage

VR.

36

Figure 4.9 Simulation results (S11) of the dual band proposed antenna at the higherfrequency band (DCS-1800 MHz) for various values of reverse bias voltage

VR.

36

Figure 4.10 Resonance frequencies of the dual band proposed antenna against reverse biasvoltage VR.37

Figure 4.11 Far field simulated radiation patterns at different bias conditions. 38

Figure 4.12 Measured return loss of the tunable antenna for reveres bias of 0, 1, and 3 V. 40

Figure 4.13 Measured return loss of the tunable antenna for reveres bias of 0, 1, and 3 V

for a modified lower band element.

40

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ABSTRACT

Small size antennas usually suffer from bandwidth limitations. Bandwidth can be

increased by adding lossy elements but, they significantly affect the efficiency of the

antenna. One method to solve the efficiency problem without increasing the antenna size

is to use the tunable antenna concept. Recently, tunable antennas attract much attention

in mobile communications. The objective of this project is to develop compact dual-band

electronically tunable microstrip antennas for operation in GSM/DCS-1800 system.

Tunable antenna, is a small size antenna that would not cover all bands simultaneously,

but provides narrower instantaneous bandwidths that are dynamically selectable at higher

efficiency than conventional antennas. Bandwidth selection can be achieved by

electronically change the reactive loading of the resonator by means of PIN diode or

varactor diode. The main characteristics of the PIN and varactor diodes that are useful

for tunable microstrip antenna design are described. The loading effect of varactor diode

on microstrip circuits is investigated and design curves that relate the biasing voltage to

the effective electrical line length for different line widths are presented. The analysis of

a compact dual-band microstrip antenna suitable for this application is, then, described.

A dual-band antenna is designed and implemented to operate at the GSM/DCS-1800

bands when connected to the varactor diode SMTD3100. The design and

implementation is carried out on Duroid dielectric substrate with r = 2.2 and thickness

1.57 mm. The simulations are performed using IE3D simulator. Simulation results show

that the required bandwidth can be easily covered with voltage changes from 0 V to 3 V,

which is suitable for mobile hand phones. Frequency shift between simulations andmeasurements due to the limited accuracy of the varactor diode capacitance is observed.

This frequency shift is compensated by adding a small piece of copper foil on the element

that resonates at the lower band. However, the circuit layout is difficult to support such

modification at the higher band. Computed radiation patterns show consistency for

different bias conditions and show omni-directional shape.

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. .

.

.

GSM/DCS-1800

.

.

PIN

.

.

.

GSM/DCS-1800

SMTD31002.2

1.57. IE3D

0 3 .

.

.

.

.

.

.

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CHAPTER 1

COMPACT MICROSTRIP ANTENNAS: CHARACTERISTICS AND

LIMITATIONS

1.1 INTRODUCTIONMicrostrip antennas (MSAs) in general have a conducting patch printed on a grounded

dielectric substrate (usually r 10). MSAs have some attractive features such as, low

profile, light weight, easy fabrication, and conformability to mounting hosts. However,

MSAs inherently have narrow bandwidth characteristics. The patch conductors, normally

of copper or gold, can assume virtually any shape, but regular shapes such as rectangular,

square, circular, elliptical, triangular, and annular ring, are generally used to simplify

analysis and performance prediction. Ideally, the dielectric constant, rof the substrate

should be low (r < 2.5), to enhance the fringe fields that account for the radiation.

However, other performance requirements may dictate the use of substrate materials

whose dielectric constants can be greater than, say, four.

In general, MSAs are half-wavelength structures. At the lower microwave frequency,

especially below 2 GHz, the size of conventional MSAs, becomes too large to be

integrated in mobile handset. However, with some modifications, any of the basic

microstrip structures can be optimized for:

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(1) Implementation in a very small area at the expense of bandwidth.

(2) Reasonable small area with acceptable bandwidth, for mobile applications.

(3) Large area with associated wide bandwidth

The possibility to employ antennas that fit in smaller volumes, but still have an efficient

behavior is certainly a challenge. The challenges and fundamental limitations for

designing small antennas will be discussed in the next section.

1.2 CHALLENGES AND FUNDAMENTAL LIMITATIONS

The term electrical small antenna has become understood to include any antenna which

fits inside a sphere of radius r 1/k, as shown in Fig. 1.1, where k is the wave number

associated with the electromagnetic field. It has been noted that as the antenna size

decreases the bandwidth decreases. The bandwidth is derived from the quality factor (Q)

by assuming that the antenna equivalent is a resonant circuit with fixed values. The

fractional bandwidth is defined as the normalized spread between the half-power

frequencies as:

Q

1

f

ff(BW)Bandwidth

center

lowerupper=

= (1.1)

If the antenna is lossy, a series resistance will be added to the radiation resistance that

results in a significant decrease of Q and so increase in bandwidth. The Q-size relation is

illustrated in Fig. 1.2 for various efficiencies [1]. These curves represent the minimum

values of Q (or the highest bandwidth) that can be obtained from an antenna whose

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structure can be enclosed within a sphere of radius r and whose radiated field outside the

sphere can be represented by a single spherical wave mode.

r

Fig. 1.1 Antenna within a sphere of radius r.

Fig. 1.2 Fundamental limit of Q versus antenna size kr < 1 , k = 2/ = /c [1].

0.1 0.3 0.5 0.7 0.9 1.1 1.3 1.5

100

80

40

20

10

8

4

2 = 100%10%

5%

r

Antenna within a

sphere of radius r

radiation efficiencyQualityfactorQ

50%

kr

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In conclusion, according to the theory developed in [1]-[3], there exists a fundamental

law that restricts the performance of the antennas enclosed in a given volume. The law

states that the Q of any linearly polarized antenna cannot be smaller than what is

obtained, as shown in Fig. 1.2, if only the lowest spherical mode is allowed to act outside

the smallest sphere that encloses the antenna. Thus, in order to approach the theoretical

limit, the antenna structure should utilize as efficiently as possible the enclosing sphere.

To do so, various wide band microstrip structures have been developed by increasing the

substrate thickness, while compactness has been achieved by decreasing the surface.

Planar inverted F antennas (PIFAs) are the most important structures developed for this

purpose. Various PIFAs have been proposed for single, dual, and triple band [4]-[12].

More size reduction is possible, if a tunable frequency band operation can be obtained

from the antenna structure instead of single wide band operation.

1.3 TUNING CONCEPT SOLUTION

The narrow bandwidth of conventional microstrip antennas has many restrictions in real-

time applications. The tunable antenna concept offers solutions to this problem. Recently,

tunable antennas attract much attention for their applications in wireless communications,

electronic surveillance, and countermeasures by adapting their properties to achieve

selectivity in frequency, bandwidth, polarization and gain.

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Microstrip antenna is a resonant element, and so its resonance frequency can be

determined by its lumped element equivalence. Therefore, any reactive loading of the

patch leads to a change in its resonance frequency. Such loading can be performed either

mechanically or electronically. Shorting pins or posts [13], stubs [14], variable dielectric

layer thickness [15], varactor diodes [16][19], switching diodes [20]- [23], and optical

control have been used to tune microstrip antennas. Pins, posts, stubs, and variable

dielectric layer thickness give rise to mechanical tuning. Whereas varactor and switching

diodes embedded in the patch and optical control of PIN diode impedance can be used for

electronic tuning of the patch antenna.

The study of dual-band electronically tunable microstrip antenna is the subject of this

project. Varactor and PIN diodes are the main electronically frequency control

components that are used for such applications. The basic theory and principle of

operation of these devices are presented in the next chapter. In chapter three, the studies

of the main microstrip antennas structures that can use electronic tuning are covered. The

design and implementation of a new dual-band structure which is more suitable for low

cost applications is discussed in chapter four. Concluding remarks and future prospective

are introduced in chapter five.

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CHAPTER 2

ACTIVE DEVICES USED IN RF TUNING

2.1 INTRODUCTION

Electronic tuning of microstrip antennas is either achieved by the mean of Varactor

diodes and/or PIN diodes. These devices are suitable for applications in microwave

frequencies. Varactors are useful for many RF applications including: frequency tuning

for active and passive circuits, frequency multiplication, frequency conversion, harmonic

generation and parametric amplification. PIN diode is the most important device for

signal control at the microwave range. Signal amplitude and phase can be easily

electronically controlled using PIN diode. In this chapter we will study the main

characteristics of both devices at the frequency of interest (< 2 GHz). The next section

will elaborate the characteristics of varactor diodes and section three is devoted to the

characteristics and the principle of PIN diodes for switching.

2.2 CHARACTERISTICS OF VARACTOR DIODES [24]

The varactor diode is one of the old microwave solid-state-devices. It is also called a

parametric diode. The varactor diode is a nonlinear device and provides voltage-

dependent variable capacitance. Varactors are generally semiconductor p-n junctions,

Schottky-barrier junctions, or point contact diodes made from gallium arsenide or silicon.

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Most varactors are fabricated on n-type semiconductors with p-type diffusion to form

junctions.

The operation of a varactor diode is based on the reverse-biased pn junction. An increase

in the reverse bias widens the depletion region between the p- and n-type substrates, and

the junction capacitance is reduced. The junction capacitance Cj for a reverse-biased pn

junction as a function of the applied reverse dc voltage V is given by

M

bi

joj )

V(1C(V)C = (2.1)

where Cj0 is the junction capacitance at zero bias, bi the contact potential (0.7V for

silicon and 1.3V for GaAs) andMis a coefficient which depends on the junction doping

profile. Both Cj0 and Mare dependent on the doping characteristics of the pn junction.

For the abrupt type varactorM=0.5, but in hyperabrupt type varactorMvaries with the

applied reverse bias between 0.5 and 5. If the range of the applied bias is sufficiently

narrow, the voltage dependency ofMmay be ignored, and it may be replaced by an

average value over that range.

Equation (2.1) is plotted in Fig. 2.1 for an abrupt pn junction (M = 0.5). The equivalent

circuit of this device in its package form is shown in Fig. 2.2. It is modeled by a voltage

dependent junction capacitance Cj (V) and the series resistance R S(V), associated with

the ohmic contact and the finite thickness of the epitaxial layer. Since the depletion

region expands as the bias is increased, the undepleted region becomes smaller which

decreases the resistance of the structure. The series resistance should be as low as

possible in order to keep the losses associated with the diode low. As the operation is

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based on the reverse-biased pn-junction, it can be assumed that the junction resistance

will be very large and may therefore be ignored. In commercial products, the diode is

usually installed in a package. Due to the package material, the geometry and the bonding

wires, the package always has some kind of package inductance Lp and package

capacitance Cp, that have an influence on the performance of the diode. Lp and Cp are

assumed to be constant with the bias voltage. The disadvantage of this model is that it

fails to take into account the non-linearities of the junction. The parameters for this

circuit have to be taken straight from the manufacturers datasheet curves. Usually, the

effect of the package capacitance, Cp on the characteristics of the diode is negligible and

can be ignored and then the impedance of the varactor Zd can be written as

))(

1()(),(

VCLjVRVZ

j

psd

+= (2.2)

Since the operation of a varactor diode is based on the reverse-biased pn junction, the

operation frequency range should be below a particular frequency which is called cutoff

frequency fcoff and determined by the diode series resistance Rs(V) and the junction

capacitance Cj (V) as

)()(2

1)(

VCVRVf

js

cutoff

= (2.3)

Since both Rs and Cj decrease as a function of the increased reverse bias, the cut-off

frequency is usually defined according to the zero bias values of these parameters

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In order to keep the diode impedance, constantly, in the capacitive operating mode, the

series resonance frequency of the diode should, in all circumstances, remain clearly

above the operating frequency range. In the series resonance condition, the reactive part

of the impedance equals zero, and the resonant frequency can be solved from Equation

(2.3) and written as

)(2

1

VCLf

jp

r

= (2.4)

Since the junction capacitance decreases as a function of the applied reverse bias, the

criterion for the capacitive operating mode is met, provided that thefrexceeds the highest

operating frequency at the zero bias level.

In low-loss designs such as antenna applications, fcutoff is usually much higher than the

series resonance frequencyfrof the packed diode (> 10 fr ). This means that in case the

series resonance frequency requirement is fulfilled, the cut-off frequency requirement is

also fulfilled. It must be emphasized that the cut-off frequency mainly defines the energy

dissipation of the varactor, while thefr defines the frequency above which the operation

of the varactor becomes inductive due to the effect of the package inductance.

In mobile phone applications the temperature of the environment can vary within a rather

large range. This can have a significant effect on the properties of the varactor-based

tuning circuit, because the capacitance of the diode tends to increase as the temperature

increases. The capacitance of a hyperabrupt diode is more sensitive to temperature in

comparison to the abrupt diode.

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2.3 CHARACTERISTICS OF PIN DIODES [24]

The p-in diode has its name because of its doping semiconductor profile which consists

of a lightly doped intrinsic region sandwiched between two doped p and n regions. The

semiconductor material is usually silicon, but gallium arsenide can be also used. PIN

diodes are used as switches or attenuators for signals at microwave frequencies. The

sketch of a PIN diode is shown in Fig. 2.3.

The PIN diode is similar to the PN diode but with smaller junction capacitance. Since the

width of the depletion zone is inversely proportional to the resistivity (or doping

concentration) of the p or n region (whichever has the lower impurity doping

concentration, the depletion region of PIN is wider than that in a PN diode. The wider

depletion region corresponds to smaller junction capacitance. The effect is very useful

for a diode used as a microwave switch. This is because the impedance of the diode

under reverse bias gets higher as the capacitance gets lower and the device becomes more

effective as an open circuit. Because of the heavy doping of the P+

or N+

regions, the

depletion does not extend far into them, and the depletion width is essentially equal to the

I region width. The junction capacitance in the reverse bias is determined by this width.

The most important property of the PIN diode is that it can, under certain circumstances,

behave as an almost pure resistance at high frequencies. The value of the resistance is

dependent on the resistivity of the intrinsic region and the applied bias current. The

intrinsic region has a high resistance R0 at zero bias. In forward bias, the junction

resistance depends on the conductivity of the I layer. An increase in the applied forward

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bias increases the injection of carriers from the P+ and N+ regions. This phenomenon

reduces the specific resistance to a level below the one obtained from doping alone, thus

inducing a lower junction resistance. Together with N+

and P+

regions, the I region, forms

a junction capacitance Cj.

The equivalent circuit can be represented as shown in Fig. 2.4. The arrow is connected to

Rj in the forward bias and Cj in the reverse bias. Cj(V) and Rj(V) will depend on the

applied bias as shown in the I-V curve of Fig. 2.5.

The circuit parameters at forward and reverse bias are given as:

Forward bias: the circuit parameters can approximately take the values:

Cj(V) = 1 pF

Rj(V) = 0.5

Zc = -j160 at 1 GHz

The circuit is almost a short circuit.

Reverse bias:

The circuit parameters can be approximated as:

Cj(V) = 0.2 pF

Rj(V) = 20 K

Zc = -j 800 at 1 GHz

Zc is much greater tha 50 and acts as good open circuit

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Fig. 2.1 Varactor characteristics.

VbiVB

Cjo

Cj(V)

Fig. 2.2 Varactor diode equivalent circuit.

Lp Rs(V) Cj(V)

Cp

P + I N +

Fig. 2.3 Physical structure of a PIN diode.

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Fig. 2.4 PIN diode equivalent circuit.

Lp RsCj

Cp

RjForward bias

reverse bias (rb)

I

50 mA

VB (30 100 V)

V1 V

A few A

Fig. 2.5 I-V characteristics of a PIN diode.

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CHAPTER 3

TUNABLE ANTENNA TECHNIQUES AND MOBILE PHONE RF

SYSTEM ARCHITECTURE

3.1 INTRODUCTION

Tunable mobile phone antennas are based on the changing of their resonance properties

by an external reactive loading. In theory, the load can be either capacitive or inductive.

The reactive load can be placed either in series or parallel to the original antenna. The

series connection can be achieved by placing the load between the short-circuit and the

ground, and the parallel connection is achieved by loading the antenna between the

ground plane and the radiating element. Due to the space limitations of the mobile phone

antennas, the parallel inductive or series capacitive loading is not very advisable, because

it shifts the resonant frequency upwards. The reactive tuning makes it possible to enhance

the impedance bandwidth of a narrowband antenna over large frequency range. In

practical mobile phone antenna design, however, the feed point of the antenna is usually

placed on a particular point on the radiating patch, which sets limits on the achievable

tuning range. In addition to the fixed feed point, the tunability of a resonator type

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antenna depends on several other factors, such as the available bias, used tuning

component, coupling of the resonator at zero bias and the characteristics of the antenna

input impedance. The general description of varactor based tuning antenna is introduced

in the next section. Moreover, the concept of switching based technique is described in

section three. In section four, the proposed mobile phone RF sub-system architecture for

dual-band GSM application is presented. Comparison between conventional and the

proposed system is also discussed.

3.2 VARACTOR BASED TUNABLE MICROSTRIP ANTENNASThe use of varactor diodes to tune microstrip antennas has been first introduced in 1982

[16]. In this approach, two varactor diodes are embedded in the patch such that the

symmetry of the patch is retained, which is essential to minimize the cross-polarization

component in the radiation pattern. Fig. 3.1 shows this configuration. A frequency

tuning range of about 30% was achieved depending on the diode characteristics and the

position of the diode in the patch. The structure is half-wavelength which is still too

large for use below 2 GHz. Shorted patch antenna is a good compromise for realizing

high radiation efficiency in a small form factor [17]. Fig. 3.2 illustrate the geometry of a

typical shorted tunable microstrip patch. The antenna length is a quarter wavelength at

the fundamental resonant frequency with the absence of the varactor diode. The

capacitance equivalence of the diode increase the antenna effective length and so reduce

its resonant frequency. Changing the capacitance by changing the bias voltage across the

diode will tune the antenna at the desired frequency. A diode tunable PIFA element that

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is divided into two sections as shown in Fig. 3.3 has been proposed in [18]. Varactor

tuning diodes are located so they connect the two sections electrically together. The

capacitance between the two sections is controlled by tuning the diodes, which

effectively varies the electrical length, and thus the resonant frequency of the top plate.

Fig. 3.1 Electronic tuning of half-wavelength microstrip antenna.

Fig. 3.2 Electronic tuning of quarter-wavelength shorted microstrip antenna.

Fig. 3.3 Diode tunable PIFA [18].

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3.3 SWITCHING BASED TUNABLE MICROSTRIP ANTENNAS

Fig. 3.4 shows some possible tuning techniques based on the capacitive or inductive

loading of a /4-resonator antenna [22]. In case (a), the tuning is accomplished by

adding a switchable capacitive load to the open end of the antenna. In cases (b) and (c)

the resonance properties can also be manipulated by modifying the electrical properties of

the short-circuit. The tuning can be achieved either by inserting additional shorting posts

or by modifying the properties of the original short circuit by a series connected

capacitive circuit. In case (b) the inductance of the short-circuit is reduced by increasing

the number of the shorting posts by using of switches. In case (c), the same effect is

achieved by a capacitive switching method. The most pronounced difference between

these two methods is that when the switch is activated to the low impedance state, the

resonance frequency shifts upwards in case (b) and, on the contrary, downwards in case

(c).

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(a)

Shorting posts

switches

(b)

(c)

(a

capacitor

capacitor

switch

Fig. 3.4 Frequency tuning by switching techniques [22].

(a) Capacitive switching.(a)Tuning by appropriate selection of shorting posts.

(b)Shorting strip with capacitive switch.

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3.4 THE PROPOSED RF SYSTEM ARCHITECTUURE

GSM system is a frequency-duplex multiple access system. Therefore, the required

handset antenna should cover the transmission and receiving frequency bands

simultaneously. As explained in chapter 1, such bandwidth can be obtained at the

expense of size and/or efficiency (see Fig. 1.2). In this case one antenna that covers

transmission and receiving bandwidths can be used. Fig. 3.5 shows the conventional

dual-band, full duplex front end handset system architecture [21]. Tunable antenna

concept is based on tuning the antenna only at the desired band. So, such antenna would

not cover all the bands simultaneously, but provides narrow instantaneous bandwidths

that are dynamically selectable at higher efficiency than conventional antennas. In this

case, two separate antennas, one for the transmission bands and the second for the

receiving bands should be used. The proposed dual-band front end handset GSM system

architecture is shown in Fig. 3.6. In this configuration, two separate antennas are used,

the first one for transmission at both transmission bands (890-915 MHz for GSM) and

(1710-1785 MHz DCS-1800) and the other for receiving at both receiving bands (935-

960 MHz for GSM) and (1805-1880 MHz for DCS-1800). Control unit and biasing

circuits will be needed to tune both antennas at the desired dynamic frequency of

operation. The design and implementation of dual-band (900 1800 MHz) tunable

antenna is described in the next chapter

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Fig. 3.5 The conventional dual-band full duplex RF front end [21].

PA

PA

LNA

LNAImage Rejection BPF

Directional

Coupler

TX/RX

Duplexer

TX/RX

Duplexer

BandSeparating

Duplexer

Image Rejection BPF

Power

Detector

Image Rejection BPFNotch filter

Directional

Coupler Combiner

PA

PATransmitting

antenna

PowerDetector

LNA

LNA

Notch filter Image Rejection BPF

SplitterReceivingantenna

Fig. 3.6 The proposed dual-band full-duplex RF front end based on a tunable antenna pair.

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CHAPTER 4

DUAL BAND TUNABLE ANTENNA FOR CELLULAR PHONE

4.1 INTRODUCTION

Various methods can be used to design compact dual-band antennas. In this chapter we

will introduce a dual-band configuration that can be tuned electronically by the use of

varactor diodes as a variable capacitance device. The proposed structure is small in size,

light in weight, and can be accommodated with active devices and biasing circuits

without the need of excess area. The structure is formed from the integration of a short-

circuited L and inverted-L shape. The resonance property depends on the antenna

geometry and the characteristics of the varactor diode used. The varactor characteristics

and its effect on microstrip circuit are described in the next section. Finally, the analysis

and design of the electronically tunable antenna is presented in the last section.

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4-2 EFFECT OF VARACTOR DIODES IN MICROSTRIP CIRCUITS

The varactor used for this application is the SMTD3001 silicon-based surface mounted

structure that can be easily used with excellent performance up to 3 GHz. The

capacitance voltage characteristic is shown in Fig. 4.1. The diode can operate well in

temperature range from -65o

to 150o. Since, the power supply of the handset mobile

phone is a 3-V battery, we will only be interested in the portion from 0-3 V of the curve.

The junction capacitance values in this range are shown in Table 4.1. The maximum

capacitance can be obtained when zero voltage is applied is 2.2 pF, while the minimum

capacitance is limited by the maximum available voltage, 3-V in mobile handset, is 1.15

pF. Since the diode is usually located at the end of an open circuit line, its effect can be

analyzed as a section of the line with the same width and effective electrical length effas

shown in Fig. 4.2. The effective electrical length effof the diode junction capacitance Cj

can be calculated as a function of the radian frequency and line characteristic

impedance as eff= cot-1

C/Zo

Fig. 4.3 shows the variation of the effective electrical length of the varcator capacitance

against microstrip line width on a Duroid substrate with r = 2.2 and of 1.57 mm

thickness. The curves are calculated for the minimum and maximum capacitances (1.15

and 2.2 pF) at 900 MHz, Fig. 4.3a, and at 1800 MHz, Fig. 4.3b. It is clear from these

curves that, the effective length decreases as the line width increases and the maximum

effective length is obtained at zero bias voltage which corresponds to 2.2 pF. For a line

of width 2 mm, this junction capacitance is equivalent to about 45o, at 900 MHz and more

than 65o

at 1800 MHz. These curves are helpful in the design stage, in order to integrate

the effect of the varactor diode in the initial design.

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Fig. 4.1 Capacitance-Voltage relation of SMTD3001.

Fig. 4.2 Electrical length equivalence of a varactor located at the end of the line.

eff

Table 4.1: change of junction capacitance with bias voltage of the SMTD3100 varactor diode

Reverse bias voltage (V) Junction capacitance (pF)

0 2.2

.3 2

.8 1.6

1 1.5

2.2 1.25

3 1.15

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0

10

20

30

40

50

2 4 6 8 10 12 14 16

Cj = 1.15 pF

Cj = 2.2 pF

Microstrip line width (mm)

eff(degrees)

(a)

Fig. 4.3 The variation of effective electrical length of the varactor diode SMTD3001

against microstrip line width on Duroid substrate with r= 2.2 and 1.57 mm thickness(a)at 900 MHz

(b)at 1800 MHz.

0

10

20

30

40

50

60

70

2 4 6 8 10 12 14 16

Effectiveelectricallength

(degrees)

Cj = 2.2 pF

b

Microstrip line width (mm)

Cj = 1.15 pF

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4.3 COMPACT TUNABLE MICROSTRIP ANTENNA

As shown in Fig. 4.4, the antenna under consideration consists of two resonant elements

an inverted large L-shape element that resonates at the lower frequency (Fig. 4.4a) and

another smaller L-shape element that resonates at the higher frequency (Fig. 4.4b) [25].

The effective electrical lengths of the lines are 1 and 2, where the corresponding

physical lengths are l1, and l2, at the lower frequency and l1, and l2, at the higher

frequency, taking the discontinuities effect into account. Z1 and Z2 are the characteristic

impedances of the microstrip lines of widths W1 or W1 and W2 or W2 respectively. At

resonance

tan 1 tan 2 = K (4.1)

where K is the ratio of the line impedances; K = Z2 / Z1. Equation (4.1) gives the

resonance condition without considering the varactor diode at the ends of l2 and l2 lines.

Now, consider the varactor diodes connected to the patch as shown in Fig. 4.5. Biasing

circuit consists of coupling capacitor 3 pF, and RF chock coils is used with each diode to

provide biasing voltage. The effect of the varactor diode in the circuit can be treated as a

transmission line section, having the same line width (W2), with an effective electrical

length eff as described in Fig. 4.3. In this case the resonance condition defined by

equation (4.1) is modified as

tan 1 tan (2 + eff) = K (4.2)

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(a)

W1

W2

l1

l2

Fig. 4.4 L and inverted L shape antenna [25].

a) L-shaped antenna that operates at the lower frequency.

b) L-shaped antenna that operates at the higher frequency.

c) Integration of (a) and (b).

(b)

l1

l2

W1

W2

(c)

Fig. 4.5 The proposed dual-band tunable microstrip antenna.

V1 V2Probe feed

Shorting post

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Using equation (4.2) (2 + eff) is plotted against 1 for different values of K in Fig. 4.6. It

is observed that, for a certain value of1, (2 + eff) decreases as K decreases resulting

in a reduction of the total antenna size. The total electrical length of the antenna is given

by t = 1 (2 + eff). For K = 1 (uniform resonator), the total electrical length is 90o

and the total length decreases as K decreases .

Fig. 4.6 is a helpful graph for a primary design of each antenna element through judicial

selection of the K factor, and hence the selection of electrical lengths of the antenna arms.

The design and implementation of the dual-band tunable antenna at the GSM/DCS1800

bands is achieved on Duroid dielectric substrate with r = 2.2 and thickness 1.57 mm.

The width of the narrow lines W1 and W1 are chosen to be 4 mm to avoid degradation of

the antenna efficiency and the width of the wider line is selected for a suitable value of

the impedance ratio K to achieve antenna size reduction and concurrently to maintain the

validity of transmission line approximation. We choose W2 = 10 mm and W2 = 7 mm to

provide a suitable radiation aperture at the 900/1800 MHz bands. The characteristic

impedances corresponding to these dimensions are Z1 = Z1 =56.5 , Z2 = 29.5 , and

Z2 = 38.5 , which yield K value of 0.52 at 900 MHz and 0.68 at 1800 MHz. The

effective electrical length of the varactor diode at zero bias in this case is 20o

at 900 MHz

and about 44o

at 1800 MHz. Fig. 4.6 provides good tool to hit a compromise point

between 1 and 2 for both antenna elements. Interpolation between the K - curves can be

used to predict the K = 0.52 curve at 900 MHz and 0.68 at 1800 MHz. The junction

capacitance is very sensitive with the reverse voltage applied zero volts. Therefore, the

frequency change per volt increases in this region that will need a sophisticated voltage

control circuit. To avoid this, the physical size of this structure is optimized to cover the

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GSM bands (890 960 MHz & 1710 1880 MHz) in tuning range between 0.8 V to 3 V.

The antenna physical dimensions after adding the discontinuity effects, are l1= 14.5, l2 =

22.5, l1= 5 and l2 = 8 mm. The discontinuity effects are calculated with the help of IE3D

software; a commercial electromagnetic simulator based on an integral equation method

and the method of moment. The layout of the designed antenna is shown in Fig. 4.7.

Simulations are carried out using IE3D. A series of simulations have been performed for

various values of junction capacitance representing the vractor diodes and biasing circuit.

The return losses (S11) for this antenna for different values of junction capacitance, or

corresponding voltage, at lower band are shown in Fig. 4.8. Variation of resonance

frequency from 826 MHz to 944 MHz is observed for voltage variation from 0 to 3 V.

This means that the, bandwidth the antenna can be tuned within the 3-V battery is about

118 MHz. This is more than the GSM bandwidth requirements at 900 MHz. Good

matching is maintained within this band. At 1800 MHz the simulation results are shown

in Fig. 4.9 for the smaller patch. From 0 to 3 V, the smaller antenna can be tuned from

1440 MHz to 1890 MHz. The tunable bandwidth is 450 MHz. This bandwidth is more

than the DCS-1800 requirements. In this case, we can avoid tuning near the zero voltage

in order to alleviate the frequency sensitivity and provide more stability. The variation of

the resonance frequencies, f1 of the larger patch and f2 for the smaller patch, against

biasing voltage, are shown in Fig. 4.10. The simulated far field radiation patterns at two

different frequencies for each band are shown in Fig. 4.11. Almost no significant

variation for difference biasing conditions is observed. The gain for all cases is about 2.6

dB. These characteristics are adequate for mobile phone requirements.

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1

2

+

eff 4

23

0.8

1

0.4

0.

0.2

Fig. 4.6 1 versus (2 + eff) for different values of K.

a small patch for interconnectionbetween ground/varactor and RF

chock

22.5 mm

10 mm

7 mm

8 mm

4 mm

3 mm

14.5 mm

12 mm

Fig. 4.7 Layout of the dual-band tunable antenna designed for GSM applications.

a small patch for

ground connection

a small patch for

terminal voltage

connection

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Fig. 4.8 Simulation results (S11) of the dual band proposed antenna at the lower

frequency band (GSM-900 MHz) for various values of reverse bias voltage VR.

-25

-20

-15

-10

-5

0

0.8 0.85 0.9 0.95 1

VR= 0.3 Vf1 = 846 MHz

VR= 1 V

f1 = 900 MHz

VR= 2.2 V

f1 = 930 MHzVR= 3 V

f1 = 944 MHz

VR= 4 V

f1 = 964 MHz

VR= 0 V

f1 = 826 MHz

f1 (GHz)

S11 (dB)

-25

-20

-15

-10

-5

0

1.4 1.5 1.6 1.7 1.8 1.9 2

Fig. 4.9 Simulation results (S11) of the dual band proposed antenna at the higher

frequency band (DCS-1800 MHz) for various values of reverse bias voltage VR.

f2 (GHz)

S11 (dB)

VR= 0.3 Vf2 = 1502 MHz

VR= 1 V

f2 = 1698 MHz

VR= 2.2 V

f2 = 1830 MHz

VR= 3 V

f1 = 1890 MHz

VR= 4 V

f1 = 1994 MHz

VR= 0 V

F2 = 1440 MHz

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0.8

1

1.2

1.4

1.6

1.8

2

0 1 2 3

f1

Resonancefrequency(G

Hz)

f2

4

Bias voltage VR (V)

Fig. 4.10 Resonance frequencies of the dual band proposed antenna

against reverse bias voltage VR.

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Fig. 4.11 Far field simulated radiation patterns at different bias conditions

(a) Radiation pattern at 900 MHz (VR= 1 V).(b) Radiation pattern at 930 MHz (VR= 2.2 V).

(c) Radiation pattern at 1502 MHz (VR= .3 V).

(d) Radiation pattern at 1700 MHz (VR= 1 V).

(a)

(d)

(b)

(c)

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4.4 EXPERIMENTAL RESULTS

The tunable antenna designed in the above section is implemented and tested. The layout

of the circuit is shown in Fig. 4.7. The double coated Duroid substrate with r = 2.2 and

1.57 mm thickness is manipulated by the available grooving machine. The chock coils

and varactor diodes are soldered on the same side where the antenna is implemented. An

HP8510A vector network analyzer is used to measure the return loss of the antenna (S11)

at different biasing conditions. The biasing voltage is provided by a dc voltage source.

V1 denotes the biasing voltage for the larger element (resonates at the lower band), and

V2 denotes the biasing voltage for the smaller element (resonates at the higher band).

The measured data are presented in Fig. 4.12. The return losses change from -8 to -15 dB

at different resonances corresponding to voltage changes from 0 to 3 V. The degradation

in return loss as compared to simulation results is attributed to the limitation of the

accuracy of the available fabrication and mounting tools. Frequency shift of about -100

MHz is observed at all biasing voltages. This is attributed to the accuracy of the varactor

diode capacitance. Varactor datasheets indicate that the capacitance accuracy can be

changed in the order of 20%. More than 100 MHz frequency shift results from such

limited accuracy. Usually, carefully designed devices give better results. Also, this

drawback can be overcome by either individually characterizing the varactor diodes or

retune the antenna circuit to compensate for this effect. On this context, another

alternative which yields better results was obtained at the lower band by adding a small

adhesive foil patch to increase the width W1 of the patch. As W1 increases, Z1 decreases

and thus K (Z2/Z1) increases, and so from Fig. 4.6, the corresponding (2 + eff) increases

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that leads to increase of resonance frequency. The modified measurements are shown in

Fig. 4.13.

-20

-15

-10

-5

0

0.7 0.8 0.9 1

-20

-15

-10

-5

0

1.2 1.3 1.4 1.5 1.6 1.7 1.8

Frequency (GHz) Frequency (GHz)

S11(dB)

S11(dB)

Fig. 4.12 Measured return loss of the tunable antenna for reveres bias of 0,

1, and 3 V(a) Lower band tuning.

(b) Higher band tuning.

VR= 0 V

f1 = 740 MHzVR= 1 V

f1 = 780 MHz

VR= 0 Vf2 = 1300 MHz

VR= 3 Vf1 = 840 MHz

VR= 1 V

f2 = 1480 MHzVR= 3 V

f2 = 1630 MHz

(b)(a)

-20

-15

-10

-5

0

0.7 0.8 0.9 10.8 0.9 1.0 1.1

Fig. 4.13 Measured return loss of the tunable antenna for reveres bias of 0,

1, and 3 V for a modified lower band element.

Frequency (GHz)

S11(dB)

VR= 0 Vf1 = 832 MHz

VR= 1 Vf1 = 900 MHz

VR= 3 Vf1 = 930 MHz

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CHAPTER 5

CONCLUSIONS AND RECOMMENDATION

In this project a tunable compact microstrip antenna concept has been studied as a

solution to virtually increase the antenna bandwidth without increasing its size or

reducing its efficiency. In this study, varactor diode has been used as a voltage control

capacitance. The main characteristics of varactor diode have been reviewed. The effect

of loading microstrip lines of different widths by a varactor diode has been analyzed

based on transmission line theory. A design procedure has been developed in chapter 4.

Based on that, a dual-band antenna has been designed for GSM/DCS-1800 bands. The

antenna structure has been selected to integrate the biasing circuit and the varactor diodes

without significant increase the structure area. Duroid dielectric substrate with r = 2.2

and thickness of 1.57 mm has been used. The IE3D simulator has been used to verify the

antenna performance before implementation. It has been shown that, the required

bandwidth can be easily covered using voltage changes from 0 to 3V, available from the

battery of mobile handset. Approximate omni directional similar radiation pattern at

different biasing voltages suitable for handset mobile phones has been observed. The

calculated antenna gain obtained from the simulation was reported as 2.6 dB for all

biasing condtions. The measurements have been performed using HP8510A vector

network analyzer. A frequency shift of about 100 MHz, compared to simulation results,

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has been noticed in the measurements. This shift is attributed to the accuracy of the

capacitance of the low cost varactor diode STMD3100. This frequency shift has been

treated at the lower band by adding a small piece of copper foil made for this purpose.

From the proposed RF system architecture of a mobile handset that can use tunable

antenna, the RF system needs two separate antennas. The first antenna is for the

transmitter and the other for the receiver. Each antenna consists of two elements, two

varactors and two biasing circuit. Since, the antenna used is the short-circuit quarter-

wave branches, two coupling capacitors are needed to isolate biasing voltage from the

ground. In conclusion the basic disadvantage of the tunable antenna is the large number

of components needed for operation which contribute to additional cost. In this regard,

an antenna design with less number of passive and active devices is required and attracts

the attention and effort of researchers in the field of microstrip antennas and circuits.

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