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1
Chapter - 1
Introduction
1.1 Variable Speed Drive:
Rotational industrial loads require operation at any one of a wide
range of operating speeds. Such loads are generally termed as variable
speed drives or adjustable speed drives. The variable speed drive systems
are also an integral part of automation. They help to optimize the process
to reduce the investment costs, energy consumption, and energy cost.
The system efficiency can be increased by the introduction of variable
speed drive operation in place of constant speed operation [1, 2].
There are three basic types of variable speed drive systems: electrical
drives, hydraulic drives and finally mechanical drives. In this thesis, only
electrical drives are focused. Drives employing electric motors are known
as electrical drives. Block diagram of an electric variable speed drive
system is shown in Fig 1.1. It consists of three basic components: the
electric motor, the power electronic converter and the control system.
The electric motor is connected directly or indirectly (through gears) to
the load. The power electronic converter controls the power flow from
power supply to the motor by appropriate control of power
2
semiconductor switches. The recent advances in power semiconductor
devices and converter topologies, electric variable speed drives are
witness a revolution in a wide variety of applications such as machine
tools and robotics drives, fans, pumps, compressors, paper mill, steel
industries, automation, traction applications, ship propulsion and
cement mills.
Fig.1.1. Block diagram of an electric variable speed drive system
1.2 Classification of Variable Speed Drives:
According to the type of electric motor, the electric variable speed
drives can be classified into two categories.
1. DC motor drives
2. AC motor drives
Power Supply
Power Electronic
Converter Gear System
Drive Control Unit
Load
Supervisory Control System
Electric
Motor
3
1.2.1 DC Motor Drives:
Traditionally, separately excited DC machines are the obvious choice
for applications in variable speed drives, where good dynamic response
and steady state performance are required. The control of a separately
excited dc motor is very straightforward, because of the commutator
within the motor. The commutator and brush allow the developed torque
of the motor to be proportional to the armature current if the field
current is held constant. The dc machines also have the excellent
dynamic performance over a wide range of operating conditions due to
inherent decoupling between field flux and armature current.
Applications are used in steel industries, robotic drives, printers,
machine tools, textile and paper industries, etc. On the other hand, dc
machines are inherently bulky, require frequent maintenance, have low
torque-to-weight ratio, in addition to having commutation problems.
Moreover, the mechanical commutator limits the maximum applicable
voltage to about 1500V and the maximum power capacity to a few
hundred kilowatts. The commutator also limits the maximum armature
current and its rate of change.
1.2.2 AC Motor Drives:
AC motors exhibit highly coupled, nonlinear and multi variable
structures as opposed to much simpler decoupled structures of
separately excited DC motors. The AC motors have a number of
advantages: light weight, inexpensive and have low maintenance
4
compared with DC motors. They require control of frequency, voltage and
current for variable speed applications. However, the advantages of AC
drives outweigh the disadvantages. AC drives replace the DC drives in
many domestic and industrial applications. The AC motor drives can be
classified into two categories.
1. Induction motor drives
2. Synchronous motor drives
1.2.2.1 Induction Motor Drives:
The three phase induction motors drives can be classified into two
types, namely
• Squirrel-cage induction motor drives
• Slip-ring induction motor drives
Both the motors are electrically equivalent as long as attention is
confined to the fundamental sine-waves of voltage, current, flux, etc
except the former has rotor-winding terminals permanently shorted
inside the motor. In case of slip-ring induction motor, the terminals of
the rotor three-phase winding are externally available to the user.
1.2.2.1.1 Squirrel-cage Induction Motor Drives:
The Nikola Tesla exhibited a crude type of three-phase induction
motor at the Frankfort exhibition of 1891. An improved construction,
with a distributed stator winding and a cage rotor, was built by Dolivo
Dobrowolsky in conjunction with the Maschinenfabrik Oerlikon and
described in 1893. This motor is the most widely used motor in the
5
industry. Traditionally, it has been used in constant and variable speed
drive applications that do not cater for fast dynamic processes. Because
of recent development of several new control technologies, such as vector
control, sensorless control and direct torque control (DTC), the situation
is changing rapidly. Squirrel-cage induction motors are much cheaper
and more rugged than the dc motor. They require little maintenance.
They can be designed as totally enclosed motors to operate in dirty and
explosive environments. All these features make them attractive for use
in industrial drives. The some of speed control methods are listed below,
which are widely used.
• Scalar control
• Vector control or Field Oriented Control (FOC)
• Sensorless control
• Direct Torque Control (DTC)
1.2.2.1.2 Slip-ring Induction Motor Drives:
The slip-ring induction motors with three rotor slip rings have been
used in adjustable speed drives for many years. In early slip-ring
induction motor drives, adjustable speed is achieved by dissipating the
energy in external resistances, connected to the slip-ring terminals of the
rotor. Modern slip-ring induction motor drives use an inverter to recover
the power from the rotor circuit, feeding it back to the supply system. So,
the speed control methods employed for slip-ring induction motor drives
are
6
• Rotor resistance control
• Slip power recovery schemes (Static Kramer drive & Static
Scherbius drive)
Generally, slip-ring induction motors are used for high power
applications where a small speed range is required.
1.2.2.2 Synchronous Motor Drives:
The speed of synchronous motors with constant rotor excitation is
determined by the stator supply frequency and the number of poles. So,
a variable frequency static inverter can extend its operation as a variable
speed drive. The main applications are gearless rolling mills, mine hoists,
traction, etc.
In this thesis, main attention is given to squirrel-cage induction motor
drives only.
1.3 Control Strategies for Squirrel-cage Induction Motor:
Induction motors are known as workhorses of industry. These are
most widely used motors due to their lower cost, rugged construction
and high power to volume/weight ratio. When operated directly from the
ac line voltage, induction motor operates nearly at constant speed.
However by means of power electronic converters, it is possible to change
the speed of an induction motor. Even though the induction motors are
desirable, their speed control is not as straight forward as that of a dc
motor. Therefore, it was natural for the researchers to think of ways,
which would take the induction motor control closer to that of a dc
7
motor. The various speed control methods, which are used to control the
speed of induction motors discussed in this section.
1.3.1 Volts/Hz
fv Control of Induction Motor:
The volts/Hz control of induction motor is by far the most popular
method of speed control because of its simplicity, and these types of
motors are widely used in industry [2, 3]. In this control, for adjustable
speed applications, supply frequency is varied. However, voltage is
required to be proportional to frequency so that the flux remains
constant, neglecting the stator resistance drop. Hence, in this method,
fv is held constant. In steady state operation, the machine air gap flux
is approximately related tof
v . As the frequency nearly approaches zero,
near zero speed, the magnitude of the stator voltage also tends to zero
and this low voltage is absorbed by the stator resistance. Therefore, at
low speed by injecting the boost voltage, the stator resistance drop is
compensated, so that rated air gap flux and hence the full load torque is
available up to zero speed. At steady state operation, if load torque is
increased, the slip increases within stability limit and a balance will be
maintained between the developed torque and the load torque.
Problems with Volts/Hz Control:
• If the supply voltage to the inverter fluctuates, the air gap flux will
vary.
8
• Also, increase in stator resistance with temperature results in
variation in air gap flux. Hence, in constant f
v control scheme
the air gap flux may drift as a result, torque sensitivity with slip
frequency or stator current will vary. If correct f
v ratio is not
maintained, the flux may be weak or may saturate.
• Torque pulsations are present at low speeds owing to presence of
fifth, seventh and eleventh and higher harmonics.
• Because of the presence of low frequency harmonics, the motor
losses are increased at all speeds causing the derating of the
motor.
These drawbacks can be overcome with the help of vector control
technique where an induction motor is controlled on the same principles
as a separately excited dc motor in which torque component and the flux
component are decoupled.
1.3.2 Vector Control of Induction Motor:
In 1971, F. Blaschke proposed a scheme, which aims at the control of
induction motor like a separately excited dc motor, called Field Oriented
Control (FOC) or vector control. As in the dc machines, torque control in
ac machines is achieved by controlling the motor currents. However, in
contrast to a dc machine, current phasor has to be controlled. This is the
reason for the terminology ‘vector control’. In the vector control, the
induction motor is analyzed in a synchronously rotating reference frame
9
where the sinusoidal variables appear as dc quantities. The torque and
the flux components are identified and controlled independently to
achieve a good dynamic response. Vector controlled techniques
incorporating fast microprocessors and DSPs have made possible the
application of induction motors for high performance applications where
traditionally only dc drives were applied. However, it should be noted
that:
• In dc machines, the armature current and main flux distribution
are fixed in space and where the torque can be established by
independently controlling the excitation flux and armature current.
• Where as in ac machine, it is much more difficult to realize this
principle because these quantities are coupled and are stationary
with respect to the stator and rotor. They also depend on modulus,
frequency and phase angles of stator current.
There are essentially two general methods of vector control [2].
They are:
1. Direct vector control or Feedback method, developed by F.
Blaschke
2. Indirect vector control or Feedforward method, developed by
K. Hasse.
These two methods are differently essential by how the unit vector is
generated for the control.
10
1.3.2.1 Direct Vector Control of Induction Motor:
The direct vector control depends on the generation of unit vector
signals from the stator or rotor flux signals. The air-gap flux signals can
be measured directly or estimated from the stator voltage or current
signals. In these systems, rotor speed is not required for obtaining rotor
field angle information. Here, the actual motor currents are converted to
synchronously rotating frame currents using park transformation. The
resulting dc quantities are compared with the reference d-axis and q-axis
components. The outputs of the controllers are used to generate the
pulsewidth modulated signals for switching the devices in the inverter
bridge feeding the motor. The main disadvantages of this method are:
• Direct method of rotor flux estimation depends on the machine
parameter; the rotor resistance variation, especially, becomes
dominant due to the temperature variation and skin effect.
• The direct method of vector control can be applied typically above
10% of the base speed because of difficulty in accurate flux signal
synthesis at low speeds.
Hence, due to these disadvantages, normally indirect method of vector
control is preferred.
1.3.2.2 Indirect Vector Control of Induction Motor:
In this method, the unit vector signal that transforms the
synchronously rotating stator voltages into stationary frame signals has
been generated from the speed signal and slip signal. The drive can easily
11
be operated from zero speed to constant power field-weakening region. It
is the most popular vector control method in industry. The main
disadvantages of this method are:
• The machine parameter variation affects the slip gain, and
correspondingly, both static and dynamic performances of the
drive are affected.
• The on-line tuning for parameter variation is more difficult.
The control methods discussed so far require a speed sensor for
closed operation. The speed sensor has several disadvantages from
standpoint of drive cost, reliability and noise immunity. The torque is
controlled indirectly.
1.3.3 Sensorless Vector Control of Induction Motor:
Sensorless vector control of an induction motor drive essentially
means vector control without any speed sensor [2, 4]. Here the
terminology ‘sensorless’ refers to only the speed and shaft sensors. It is
possible to estimate the speed signal from machine terminal voltages and
currents. The speed estimation methods can generally be classified as
follows:
• Slip calculation
• Direct synthesis from state equations
• Model referencing adaptive system (MRAS)
• Speed adaptive flux observer
• Extended kalman filter (EKF)
12
• Slot harmonics
Many of the sensorless techniques depend on the machine
parameters, temperature, saturation levels, etc.
1.3.4 Direct Torque Control of Induction Motor:
In the mid 1980s the Direct Torque Control (DTC) principle was
developed by Takahashi and Noguchi for low and medium power
applications and Direct Self Control (DSC) principle was established by
Depenbrock for high power applications. As the name suggests, DTC or
DSC regulates the motor torque and flux directly. In the DTC approach,
the reference torque and reference flux are compared to the estimated
motor torque and the estimated stator flux respectively, both employing
hysteresis controllers. The torque and flux hysteresis controller output
logic signals are evaluated in an optimal switching logic table to generate
the inverter switching device gate signals. The generation of inverter
switching state is made to restrict the stator flux and electromagnetic
torque errors within the respective flux and torque hysteresis bands and
to obtain the fastest torque response and highest efficiency at every
instant. The DTC scheme is found to be very promising and valuable as
compared to FOC. But, DTC has few drawbacks such as more steady
state ripple in flux, torque and current and variable switching frequency
due to hysteresis bands.
13
1.4 Literature Review and State of the Art Assessment:
1.4.1 Introduction:
The one and a half century of progress in the electric machines field,
about three quarters of a century of progress in the power electronics
field, and about half a century of progress in the micro-
electronics/macro-electronics and control fields are inherited in the state
of the art pulsewidth modulated voltage source inverter (PWM-VSI)
drives. Since they involve various disciplines of engineering and there
has always been a strong demand for them in the market, PWM-VSI
drives have continuously drawn the attention of many researchers all
around the world. Among the various PWM-VSI drives, the induction
motor drives with cage type machines have found wide range of
applications and have become the workhorse of industry due to their
simplicity and ruggedness. These motors can be fed from current source
inverters (CSI) or voltage source inverters (VSI), and used as variable
speed drives. Recent advances in semiconductor technology have led to
new generations of fast-acting power semiconductor switches like GTOs,
MOSFETs, IGBTs, and more recently IGCTs. The performance and
characteristics of these switches strongly favor the VSI topology over the
CSI one. This has been a major reason for VSI fed induction motor drives
becoming more popular than CSI fed induction motor drives. Pulsewidth
Modulation (PWM) strategies are required for switching the devices in a
14
VSI appropriately to generate variable voltage, variable frequency, 3-
phase AC required for the variable speed induction motor drive.
Following a brief review of the various control techniques for induction
motor drives and state of the art DTC of induction motor drives will be
described and the fundamental contributions to the area will be
discussed in detail.
1.4.2 Control Techniques for Induction Motor Drives:
The various speed control techniques for three-phase squirrel cage
induction motors are
• Constant Volts per Hertz Control
• Field Oriented Control (FOC) or Vector Control
• Sensorless Vector control
• Direct Torque Control (DTC)
1.4.2.1 Constant Volts per Hertz Control:
The block diagram of volts per hertz control of induction motor is
given in Fig. 1.2. In this method, the inverter output voltage is varied
proportionally to the reference frequency such that constant stator flux is
maintained. In an induction motor drive, this operating mode results in
shunt speed-torque characteristics (linear portion of the torque-speed
curve), yielding low slip frequency and therefore high energy efficiency
and good speed regulation. Therefore, the method gained wide
acceptance in many industrial and residential induction motor drive
15
speed regulation applications as given by B.K. Bose [2], R. Krishnan [3],
D.A. Bradley et al [6] and B. Mokrytzki [7-8].
Fig. 1.2 Open-loop volts per hertz speed control of induction motor
The performance of the fv control is not satisfactory, because the
rate of change of voltage and frequency has to be low. A sudden
acceleration or deceleration of the voltage and frequency can cause a
transient change in the current, which can result in drastic problems.
Moreover, they exhibit limited speed response, poor load torque
disturbance characteristic, and inferior low speed characteristics. Some
efforts were made to improve fv control performance, but none of these
improvements could yield a fv torque controlled drive systems and this
made DC motors a prominent choice for variable speed applications. In
IM
Diode Bridge Rectifier
L
C
Sine Triangle PWM G
+ +
Vo
V*
ω*
AC supply
16
printing press applications, packaging applications, servo applications
with very high resolution position control etc, where precise control is
mandatory, the performance of fv drives is not satisfactory. In such
type of applications, traditionally DC motor drives have been employed
with a shaft encoder. Typical application areas of fv drives are pumps,
ventilation systems, etc. which have passive torque-speed characteristics
and no precise speed regulation requirement.
1.4.2.2 Vector Control:
In the volts per hertz control, the voltage and frequency are the basic
control variables of the induction motor. In a voltage fed drive, both the
torque and air gap flux are functions of voltage and frequency. This
coupling effect is responsible for the sluggish response of the induction
motor and moreover, the system is easily prone to the instability.
However, the continuous progress in induction motor control theory,
power electronics, and digital signal processors yielded the modern
vector controlled induction motor drives [9-15] which can match the
performance and reliability characteristics of dc drives and cost less. The
invention of vector control, which is also known as decoupling,
orthogonal, transvector or field oriented control (FOC) in the beginning of
1970s, and the demonstration that an induction motor can be controlled
like a separately excited dc motor, brought a renaissance in the high-
performance speed control of induction motor drives. Modern vector
controlled squirrel cage induction motor drives meet the demanding
17
performance criteria of most high performance speed control
applications. In the vector control method, an ac machine is controlled
like a separately excited dc machine. This analogy is explained by B.K.
Bose [2] as in Fig. 1.3.
Fig. 1.3 (a) Separately excited dc motor (b) vector controlled induction motor.
In a dc machine, neglecting the armature demagnetization effect and
field saturation, the torque is given by
fate IIkT '= (1.1)
Where aI is the armature or torque component of current and fI is the
field or flux component of current. In a dc machine, the control variables
aI and fI can be considered as orthogonal or decoupled “vectors”. In
normal operation, the field current is set to maintain the rated field flux
aI fI
(a)
IM
Vector
Control
Inverter
*qsi
*dsi
(b)
18
and torque is changed by changing the armature current. Since the
current fI or the corresponding field flux is decoupled from the
armature current, the torque sensitivity remains maximum in both
transient and steady state operations. This mode of control can be
extended to an induction motor also if the machine operation is
considered in a synchronously rotating reference frame where the
sinusoidal variables appear as dc quantities. In Fig. 1.3 the induction
motor with inverter and vector control is shown with two control inputs,
*dsi and *
qsi . The currents *dsi and *
qsi are the direct-axis component and
quadrature-axis component, respectively, of the stator current, where
both are in a synchronously rotating reference frame. In vector control
*dsi is analogous to the field current fI and *
qsi is analogous to the
armature current aI of a dc machine. Therefore, the torque can be
expressed as
*** 'ˆ dsqstqsmte iikikT == ψ (1.2)
Thus the similarity between the production of the electromagnetic
torque in a compensated dc machine and in symmetrical, smooth air gap
induction machine has been established. However, it should be noted
that:
• In dc machines, the armature current and main flux distribution
are fixed in space and where the torque can be established by
independently controlling the excitation flux and armature current.
19
• Where as in an induction machine, it is much more difficult to
realize this principle because these quantities are coupled and are
stationary with respect to the stator and rotor. They also depend
on modulus, frequency and phase angles of stator current.
The search for simple control schemes, similar to those used for dc
machines has led to the development of “vector controlled schemes”.
There are essentially two general methods of vector control. They are:
• Direct vector control or Direct Field Oriented Control (DFOC)
• Indirect vector control or Indirect Field Oriented Control (IFOC)
These methods are differentiated on how the unit vector signals are
generated from stator, rotor or air-gap flux signals. The DFOC method
was presented by F. Blaschke [9] and it employs flux sensors. The IFOC
method was presented by K. Hasse [10] and it employs a shaft encoder to
close the speed loop.
In FOC, the magnetizing flux and torque producing components of the
stator currents are properly and independently distributed both during
steady state and dynamic conditions. As explained by D.W. Novotny and
T.A. Lipo [11] and Rik W. De Doncker and D.W. Novotny [12], by
regulating each component independently with a high performance
current controller, the drive torque can be controlled in the same precise
manner as the DC machine. Since installing flux sensors in the stator or
the air gap of a machine is difficult, and the operation is not reliable, the
DFOC method is practically rarely employed in its original form.
20
Employing flux observers, the DFOC method provides high performance
torque control, in particular in the high speed region where the stator
resistance voltage drop is small compared to the stator EMF and the
stator flux observer is highly accurate. The stator flux oriented DFOC
method is attractive for traction, spindle tool etc, applications which
require operation in a wide field weakening region. However, near zero
speed the stator flux observer estimator error becomes substantial due to
the dominance of the stator resistive voltage component over the nearly
zero EMF and the DFOC method looses performance. In a large number
of applications requiring high performance in the low speed operating
region the rotor flux oriented IFOC method is utilized.
With accurate parameter adaptation, the IFOC based induction
machine drives can provide servo performance in a wide speed region.
Since the torque regulation quality of an FOC induction motor drive is
mainly dependent on the current controller accuracy and bandwidth,
high performance motion control requires high performance current
regulators. The hysteresis type current controllers which have superior
dynamic performance have not gained acceptance in motor drives due to
the difficulty in controlling their switching frequency and significant
waveform distortion. Employing high switching frequency IGBT devices
and high performance digital signal processors or microprocessors, high
performance current controlled drives provide high torque/speed
bandwidth, hence high motion quality. High performance FOC drives
21
have been successfully employed in industrial and servo drive
applications which are summarized by T. Kume and T. Iwakane [13]. The
evolution of FOC drives from concept to industrial products and
successful applications has been summarized by W. Leonard in [14-15]
in detail.
1.4.2.3 Sensorless Vector Control:
Sensorless vector control induction motor drive essentially means
vector control without any shaft encoder or speed sensor. An incremental
shaft mounted speed encoder; usually an optical type is required for
closed loop speed or position control in vector controlled drives. A speed
encoder is undesirable in a drive because it adds cost and poses
reliability problems, besides the need for a shaft extension and mounting
arrangement. To reduce total hardware complexity, cost and to increase
mechanical robustness, it is desirable to eliminate speed and position
sensors in vector-controlled drives. Drives operating in hostile
environments or in high speed drives speed sensors cannot be mounted.
To replace the sensor, the information of the rotor speed is extracted
from measured stator voltages and currents at the motor terminals.
Continuing research has concentrated on the elimination of the speed
sensor at the machine shaft without deteriorating the dynamic
performance of drive control system. Speed estimation is an issue of
particular interest with induction motor drives where the mechanical
speed of the rotor is generally different from the speed of the revolving
22
magnetic field. The advantage of speed sensorless induction motor drives
are reduced hardware complexity, lower cost, reduced size of the drive
machine, elimination of the sensor cable, better noise immunity,
increased reliability and less maintenance requirements.
The pioneering work in the shaft encoderless motor speed control area
was reported by R. Jötten and G. Maeder in 1983 [16]. They employed
the induction motor fundamental model to estimate the slip frequency
and the back emf of the machine and provided a closed loop controller to
regulate the slip such that superior dynamic performance could be
obtained in a wide speed region, including the field weakening region.
Although a large variety of shaft encoderless control methods have been
reported from that time to the present date, only a few found practical
applications, which are given by B.K. Bose [2], Peter Vas [4], Tsugutoshi
Othani et al [17], C. Ilas et al [18] and J. Holtz [19-20].
1.4.2.4 State of the Art DTC of Induction Motor Drives:
In addition to vector control systems, instantaneous torque control
yielding fast torque response can also be obtained by employing Direct
Torque Control (DTC) [21, 23] or Direct Self Control (DSC) [22, 24]. As
the name suggests, the DTC method regulates the motor torque and flux
directly. In the mid 1980s the DTC principle was developed and
discussed by Isao Takahashi and T.Noguchi [21] for low and medium
power applications and DSC principle was established by M. Depenbrock
[22] for high power applications. In this thesis, the attention is mainly
23
focused on the DTC scheme. In the DTC approach, the reference torque
is compared to the estimated motor torque and the reference stator flux
is compared to the estimated stator flux, both employing hysteresis
controllers. The torque and flux hysteresis controller output logic signals
are evaluated in an optimal switching logic table to generate the inverter
switching device gate signals. The generation of inverter switching state
is made to restrict the stator flux linkage and electromagnetic torque
errors within the respective flux and torque hysteresis bands and to
obtain the fastest torque response and the highest efficiency at every
instant. The DTC scheme is found to be very promising and valuable as
compared to FOC. Moreover, using the DTC it is possible to obtain a good
dynamic control of the torque without any speed sensors or position
sensors on the machine shaft. Thus, DTC can be considered as
sensorless type control techniques.
In DTC, the stator flux can be calculated from the motor terminal
voltages and stator resistance. Variations in the stator resistance result
in significant errors in the stator flux, especially at low speeds. This
problem can be overcome by using the slip relation from indirect rotor
flux field orientation to locate the position of the rotor flux. The rotor flux
position is then used to locate the position of the stator flux. Also, the
motor speed and rotor resistance are used to calculate the position of the
stator flux at low speeds. With this, the advantages of DTC scheme are
maintained over the entire speed range as explained by Thomas G.
24
Habetler et al [25]. Moreover, the robust start and improved operation in
the zero speed region can be achieved easily by introducing the
additional carrier signal to the torque comparator input as given by
Kazmierkowski and Kasprowicz [26].
Thus, unlike FOC, DTC operates with closed torque and flux loops but
without current controllers. In spite of its simplicity, DTC allows a good
torque control in steady state and transient operating conditions to be
obtained. Moreover, DTC has simple and robust control structure and is
not sensitive to rotor parameters. A review of recently used DTC
algorithms for VSI fed induction motor drives has been presented and
discussed by Giuseppe S. Buja and Marian P. Kazmierkowski [27]. A
detailed comparison between FOC and DTC, emphasizing advantages
and disadvantages are provided by Domenico Casadei et al [28] and
concluded that DTC might be preferred for high dynamic applications.
Hence, the DTC scheme was introduced in commercial products by Asea
Brown Boveri (ABB) and therefore created wide interest. This is very
significant industrial contribution and it has been stated by ABB that
DTC is the latest ac motor control method and it can be considered to be
next generation motor control technologies. Therefore, DTC has gaining
more industrial applications such as high performance applications,
electric vehicle applications, etc as explained by Peter Vas [4], James N.
Nash [29], Pekka Tiitinen and Surandra [30] and Jawad Fiaz et al [32].
25
Though DTC has high dynamic performance, it has few drawbacks
that can be summarized as high current, torque and flux ripple, variable
switching frequency due to hysteresis controllers and high noise level at
low speeds, etc. The effect of torque and flux hysteresis band amplitudes
on the performance of induction motor drive has been studied by D.
Casadei et al [33] and Jun-Koo Kong et al [34-35]. The amplitude of the
flux hysteresis band mainly affects the motor current distortion in terms
of low order harmonics. Small flux hysteresis bands lead to sinusoidal
current waveforms, while small torque hysteresis bands allow smoothed
torque to be generated. On the other hand, small hysteresis bands
usually determine high switching frequency thereby increasing the
switching losses. Moreover, the switching frequency of the torque
controller has a peak value at medium speed due to the effect of back
emf, while that of the flux controller is proportional to operating speed
[35]. The analytical determination of the relationships between the
applied voltage vector and the corresponding torque and flux variations
is given by D. Casadei et al [36], from which, it has been observed that
the effects produced by a voltage vector are strongly dependent on both
rotor speed and voltage vector direction relative to the rotor flux. The
maximum torque variation is obtained by applying a voltage vector along
the direction perpendicular to the rotor flux vector. Thus, the presence of
torque and flux hysteresis bands in DTC causes ripples in stator current,
stator flux and torque that results in more harmonics in the line current.
26
To increase the dynamic performance of DTC and to decrease the
ripple in torque, various switching control strategies had been proposed
in the literature. The effect of the applied voltage on the torque response
is strongly dependent on rotor angular speed. To tackle the problem of
stator flux drooping at low speeds, to reduce the harmonic contents in
the stator current and to reduce the switching frequency, the method of
“variable switching sectors” for DTC has been proposed by CG Mei et al
[37]. To reduce the ripple in torque further, a series of switching control
strategies have been presented by E. Galvan et al [38] and G. Escobar et
al [39]. In conventional DTC (CDTC), which was proposed by Takahashi,
the selected voltage vector is not always the best one since only the
sector is considered where the flux linkage space vector lies without
considering its accurate location. As the CDTC has a fewer number of
selectable voltage vectors, it causes higher ripples in the flux and torque.
To overcome this problem, a unified flux control (UFC) method for DTC
has been developed by Joon Hyoung Ryu et al [40]. In UFC, a voltage
space vector is calculated for a deadbeat action and a minimum-distance
vector selection scheme replaces the switching vector look-up table to
minimize the flux and torque ripples. Moreover, torque ripple can be
reduced, by applying a suitable voltage vector from the switching table
for the time interval needed by the torque to reach the upper or lower
limit of the band, where the time interval is calculated from a suitable
modeling of the torque dynamics as explained by Vanja Ambrožič et al
27
[41]. This method is also known as band-constrained technique in which,
depending on the inverter voltage vector and the operating conditions,
the time interval may extend over several sampling periods. Therefore,
the inverter switching frequency settles automatically to the minimum
value.
Now a days, the intelligent controllers like fuzzy, neuro and neuro-
fuzzy controllers play a major role in industrial applications. To improve
the dynamic performance of torque and flux, stator resistance
estimation, stator flux estimation, tuning procedure, etc, intelligent
control algorithms given by Sayeed A. Mir et al [42], I.G. Bird and H.
Zelaya De La Parra [43], Fatiha Zidani and Rachid [44], Luis Romeral et
al [45] and Pawel Z. Grabowski et al [46] can be implemented to the DTC
algorithm. As there are no sector borders, there is no current and torque
distortion caused by the sector changes. During the low speed region
also, the performance can be improved by using fuzzy logic or neuro-
fuzzy controllers.
A substantial reduction in torque, flux and current ripples could be
obtained using the discrete space vector modulation (DSVM) algorithm
developed by D. Casadei et al [47] and Xin Wei et al [48]. DSVM uses
prefixed time intervals within a cycle period that results more number of
voltage vectors with respect to those used in conventional DTC. The
increased number of voltage vectors allows the definition of more
accurate switching tables in which the selection of voltage vectors is
28
made according to the rotor speed, the flux error and torque error. In
DSVM, one sampling time period is divided into ‘m’ equal time intervals.
One of the VSI voltage vectors is applied in each time interval. The
number of voltage vectors, which can be generated is directly related to
‘m’. However, a good compromise between the errors compensation and
the complexity of the switching tables is achieved by choosing m = 3 [47-
48]. Using DSVM algorithm with three equal time intervals, 36
synthesized non-zero voltage vectors are obtained. If the stator flux
vector is assumed to be in first sector, then 19 voltage vectors can be
used. Then, different voltage vectors are chosen for different speed
ranges. A fuzzy logic controller can be designed to select synthesized
voltage vectors in DSVM based DTC [48]. Thus, DSVM allows the
performance of DTC scheme in terms of flux and torque ripple and
current distortions to be improved without increasing the complexity of
the power circuit and the inverter switching frequency.
To overcome the problem of variable switching frequency, and torque
ripple, few controllers have been proposed in the literature [49-50]. This
can be done in two ways. In first method, the optimal switching instant is
calculated at each switching cycle to satisfy the ripple minimum
condition based on the instantaneous torque slope equations. In second
method the conventional three-level torque hysteresis comparator is
replaced by a new controller, which consists of two triangular waveform
generators, two comparators and a PI controller. To operate the DTC
29
algorithm with constant switching frequency and to reduce the torque
ripple few methods have been proposed in [51].
In recent years, to overcome the problem of ripples and varying
switching frequency a voltage modulation algorithm, which is known as
Space Vector Pulsewidth Modulation (SVPWM) has been used in the
literature [52-68]. The recently reported SVPWM algorithm [58-67] has
become very popular over the last decade. In this method, the reference
is provided as a voltage space vector, which is sampled once in every
subcycle and an average voltage vector equal to the sampled reference
voltage vector is generated by time-averaging of the different voltage
vectors produced by the inverter. The SVPWM is a superior PWM
technique for three phase inverter drives compared to the traditional
regularly sampled triangular comparison technique. Space vector
approach has the advantages of lower current harmonics and a possible
higher modulation index compared with the three phase sinusoidal
modulation method and ease of digital implementation.
A novel scheme was reported by Thomas G. Habetler et al [52] that
calculate the inverter switching pattern directly in order to control the
torque and flux in a dead beat fashion over a constant switching period.
This is accomplished by calculating the voltage space vector required to
control the torque and flux on a cycle-by-cycle basis using the calculated
flux and torque errors sampled from the previous cycle and estimated
value of the back EMF in the machine.
30
To get constant switching frequency and to increase the inverter
switching frequency for the same sampling frequency, the symmetrical
regular sampled SVM technique was used by Yen-Shin Lai and Jian-Ho
Chen [53] for inverter control of the DTC based drive. A new SFVC based
DTC was reported by D. Casadei et al [54] along with a simple closed loop
flux estimator to improve the drive performance in the very low speed
region, including zero speed. Further, a simplified DTC algorithm based
on SVPWM was reported by Lixin Tang et al [55-56], in which instead of
the switching table and hysteresis controllers, a PI controller and
reference flux vector calculator (RFVC) were used to determine reference
stator flux linkage vector. The RFVC generates the reference flux vector
according to the error in the torque, which is based on the current
estimated flux linkage vector. Moreover, a special SVM pattern has been
used to reduce the switching frequency of the inverter. Further, closed
loop digital control for both flux and torque was implemented by Cristian
Lascu et al [57] in a SVPWM based DTC to improve the transient
performance and steady state ripple and to preserve the robustness. A
sensorless hybrid DTC drive based on SVPWM for high volume low cost
applications was reported by Cristian Lascu and Andrzej M.
Trzynadlowski [57]. In this hybrid method, under the transient operating
conditions, the drive is controlled by using the classical bang-bang DTC
and in the steady state, using linear torque and flux controllers, the
control system generates a reference voltage vector for the inverter.
31
Now a days, the attention is paid to determine the switching losses
of the inverter. The dependency of the switching losses of a bridge leg of a
PWM converter system with a pulse rate was explored by Johann W.
Kolar et al [64]. The modern PWM methods can be separated into two
groups. In the continuous PWM (CPWM) methods, the modulation waves
are always within the triangular peak boundaries and within every
carrier cycle triangle and modulation waves intersect, and, therefore, on
and off switching occur. In the discontinuous PWM (DPWM) methods, the
modulation wave of a phase has at least one segment which is clamped
to the positive or negative dc rail for at most a total of 120o, therefore,
within such intervals the corresponding inverter leg discontinues
modulation. Since no modulation implies no switching losses, the
switching loss characteristics of CPWM and DPWM methods are
different. Hence, in recent years the attention is paid on the DPWM
methods. Several DPWM methods have been reported in the literature
[64-68]. A few carrier based DPWM methods are reported by Ahmet
Hava et al [65-67]. Among these, Depenbrock’s DPWM1 and Ogasawara’s
DPWM methods have gained recognition due to their low harmonic
distortion at high voltage utilization and the controllability of the
switching losses. Simple and powerful analytical and graphical carrier
based PWM tools have been presented in [67]. Also, expressions for
harmonic distortion and switching loss factor (SLF) are given. Moreover,
the performance characteristics of various PWM methods have been
32
compared. The switching loss and waveform quality indicate SVPWM at
low modulation and DPWM methods at the high modulation range have
superior performance. Based on this comparison, a high-performance
generalized DPWM (GDPWM) method, with superior high modulation
operating range performance characteristics was presented in [66]. The
GDPWM algorithm is suitable for most high performance PWM-VSI drive
applications. Also, an algorithm combining GDPWM and conventional
SVPWM to maximize the drive performance in the whole modulation
range is developed. Also, the relationship between zero sequence signal
and space vectors and the relationship between the distribution of zero
vectors and different carrier based PWM modulators are systematically
investigated without dependence on the load. Then the expressions for
the continuous modulating signals for the switching devices of the VSI
required to generate unbalanced three phase voltages are given by
Olorunfemi Ojo [68]. Then, for the generation of balanced and
unbalanced phase voltages, a GDPWM scheme, in which the modulation
signals of the switching devices are inherently discontinuous, is reported
by Olorunfemi Ojo [68]. In this method, by varying a parameter various
discontinuous modulating signals can be generated.
So far, a number of PWM techniques have been discussed for VSI fed
induction motor drives. The techniques for the generation of PWM
waveforms can be broadly divided into:
• Offline PWM generation techniques
33
• Online PWM generation techniques
Offline PWM techniques are those where the switching instants of the
inverter are stored in the form of lookup tables, which are previously
calculated and used. The online PWM techniques are more common
where the fundamental cycle is divided into many subcycles in each of
which the volt-second balance is maintained. The online PWM
techniques can be further subdivided into two categories on the basis of
approach, namely the triangle comparison (TC) approach and the space
vector (SV) approach. In the TC approach, three-phase modulating waves
are compared against a common triangular carrier to determine the
switching instants of the three phases. The most common and popular
modulating waves are sinusoidal waves. Any triplen frequency
component can be added as zero sequence components to the 3-phase
sinusoidal waves. The choice of these triplen frequency components is a
degree of freedom in this approach. In the SV approach, the voltage
reference is provided in terms of a revolving space vector. The magnitude
and the frequency of the fundamental component are specified by the
magnitude and frequency respectively of the reference vector. The
reference vector is sampled once in every subcycle. The inverter is
maintained in different states for appropriate durations such that an
average voltage vector equal to the sampled reference vector is generated
over the given subcycle. The inverter states used are the two zero voltage
vectors, and the two active voltage vectors, whose voltage vectors are the
34
closest to the commanded voltage vector. The division of the zero voltage
vector duration between the two zero states is a degree of freedom in the
space vector approach. This division of zero vector time in a subcycle is
equivalent to adding a common-mode component to the 3-phase average
pole voltages. The same PWM waveform can be generated based on both
the approaches as explained by G. Narayanan and V. T. Ranganathan
[63].
The ripple in torque can also be decreased by using the multilevel
inverters. An increase in the number of levels improves the torque quality
reducing the ripple amplitude. Therefore, by using the multilevel
inverters, the torque performance of direct torque control of induction
motor in high power and medium power applications can be improved as
explained by Kyo-Beum Lee et al [69], A. Damiano et al [70], Zhuohui
Tan et al [71] and José Rodríguez [72]. But, in the multi level concept,
though the torque performance is improved, the cost and complexity will
be increased.
Though the look-up table based 3-level inverter fed DTC drives give
good performance when compared with the look-up table based 2-level
inverter fed DTC drives, it gives varying switching frequency operation of
the inverter and gives more harmonic distortion. To obtain constant
switching frequency operation and to achieve superior waveform quality,
various PWM algorithms have been proposed in the literature. Nabae,
et.al. proposed a PWM algorithm for neutral point clamped (NPC) 3-level
35
inverter in [73]. Nowadays, the multilevel inverter fed drives are
becoming popular in many industrial and electrical vehicles applications
especially for medium and high power applications [74-77]. A detailed
survey on the multilevel inverters and various topologies of the multilevel
inverters are discussed in [77]. The waveform quality can be increased by
increasing the number of levels. But, as the number of level increases,
the complexity involved in the PWM algorithm and power circuit also
increases. To decrease the complexity involved in the PWM algorithms for
a multilevel inverter, several simplified PWM algorithms have been
proposed in the literature. A simplified SVPWM algorithm has been
proposed for a three-level inverter by suing the concept of SVPWM
algorithm for a two-level inverter in [78-79]. In this algorithm, the
switching times can be calculated similar to a two-level inverter.
However, this algorithm requires angle and sector calculations, which
increases the complexity of the PWM algorithm as the number of levels
increases.
To decrease the complexity of the SVPWM algorithm, it is
necessary to avoid the angle and sector calculations. A simplified
approach for SVPWM algorithm is proposed in [80], which uses
instantaneous phase voltages only for the calculation of gating times of
the inverter. The same approach is extended to the various
discontinuous PWM algorithms along with the SVPWM algorithm in [81].
However, these approaches are proposed for two-level inverters only. The
36
same approach is extended to a n-level inverter in [82]. This algorithm
also uses instantaneous phase voltages only. By using the concept of
effective time, the algorithm is extended for multilevel inverters under
both linear and over modulation regions.
Nowadays, many researchers have been focused their interest on
open-end winding induction motor drives in medium power applications.
The open-end winding induction motor drives offer many advantages
when compared with the normal drives. The open-end winding induction
motor drives fed by two inverters on either ends. By using the two 2-level
inverters on both sides of the winding, the phase voltages can be
obtained similar to the three-level inverter. To control these two 2-level
inverters, various PWM approaches are presented in the literature [83-
92].
Among the various PWM algorithms, decoupled and nearest sub-
hexagonal centre PWM (NSHCPWM) algorithm are popular approaches
for open-end winding induction motor drives. In both the approaches,
the two inverters will be operated with 180 degrees phase shift. Though
the implementation of decoupled PWM algorithm is simple, it gives more
harmonic distortion in line currents and voltages [91]. Hence, nowadays
the research interests have been focused on NSHCPWM algorithm. In
[83], a look-up table based NSHCPWM algorithm has been presented.
However, this approach will generate large common mode voltage
variations. To overcome the drawbacks of NSHCPWM algorithm, which is
37
presented in [83], various approaches have been proposed in the
literature [86-92].
Thus, though the DTC offers good dynamic performance, it has few
drawbacks such as steady state ripple in torque, flux and current,
varying switching frequency and sensitive to load torque disturbances.
Hence alternatives must be explored to reduce the steady state ripple in
torque, flux and current and memory size and to get constant switching
frequency. Mainly, this research is focused on the various PWM
algorithms to overcome the problems of steady state ripple, switching
frequency variations and memory size. Moreover, various simplified PWM
algorithms have been presented for multilevel inverters and open-end
winding induction motor drives.
1.5 Summary:
Recently, the DTC is gaining popularity in the high-performance
applications due to its numerous advantages. Moreover, the PWM
algorithms also attracting many researchers nowadays due to the
advantages of PWM algorithms. A detailed literature survey and the state
of art of DTC drives and various PWM algorithms have been presented in
this chapter.